WO2010010710A1 - Power converter - Google Patents
Power converter Download PDFInfo
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- WO2010010710A1 WO2010010710A1 PCT/JP2009/003471 JP2009003471W WO2010010710A1 WO 2010010710 A1 WO2010010710 A1 WO 2010010710A1 JP 2009003471 W JP2009003471 W JP 2009003471W WO 2010010710 A1 WO2010010710 A1 WO 2010010710A1
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- voltage
- power
- output
- circuit
- inverter
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- G—PHYSICS
- G05—CONTROLLING; REGULATING
- G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
- G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
- G05F1/66—Regulating electric power
- G05F1/67—Regulating electric power to the maximum power available from a generator, e.g. from solar cell
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
- H02J7/34—Parallel operation in networks using both storage and other dc sources, e.g. providing buffering
- H02J7/35—Parallel operation in networks using both storage and other dc sources, e.g. providing buffering with light sensitive cells
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
- H02J7/34—Parallel operation in networks using both storage and other dc sources, e.g. providing buffering
- H02J7/345—Parallel operation in networks using both storage and other dc sources, e.g. providing buffering using capacitors as storage or buffering devices
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/06—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider
- H02M3/07—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider using capacitors charged and discharged alternately by semiconductor devices with control electrode, e.g. charge pumps
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
- Y02E10/00—Energy generation through renewable energy sources
- Y02E10/50—Photovoltaic [PV] energy
- Y02E10/56—Power conversion systems, e.g. maximum power point trackers
Definitions
- the present invention relates to a power converter for converting DC power of a DC power source into DC power having a different voltage.
- a booster circuit as a conventional power converter is configured to include a switch element, an inductor, a diode, and a smoothing electrolytic capacitor on the output side. Then, the input voltage from the solar cell smoothed by the smoothing electrolytic capacitor on the input side is boosted by turning on and off the switch element of this booster circuit, or is passed through without being boosted by the booster circuit. (See, for example, Patent Document 1).
- Such a power conversion device requires a large-capacity reactor along with an increase in output power capacity, resulting in problems such as an increase in size and weight of the device. Further, when the switching element is switched at a high frequency in order to avoid this problem, a great loss and noise are generated.
- the present invention has been made to solve the above problems, and in a power conversion device that performs DC / DC conversion, the power loss and noise are reduced, and the device configuration is reduced in size and weight.
- the purpose is to promote.
- the power converter according to the present invention is configured by connecting in series the AC side of one or more single-phase inverters each having a semiconductor switch element and a DC voltage source, and connecting the AC side in series with the output of the DC power source.
- a shorting switch having one end connected to the inverter circuit and the other end connected to the negative electrode of the smoothing capacitor. Then, DC / DC conversion is performed using charging / discharging of DC power in the inverter circuit.
- the DC / DC conversion is performed using the charging / discharging of the DC power in the inverter circuit, a large capacity reactor is not required. Further, the short-circuit switch and the semiconductor switch element in the inverter circuit do not require high-frequency switching, and the voltage handled by switching of the inverter circuit can be made relatively small. For this reason, the power converter device with which reduction of the power loss and noise, and reduction in size and weight of the device configuration are promoted can be realized.
- Embodiment 7 of this invention It is a main circuit block diagram of the power converter device by Embodiment 7 of this invention. It is a figure which shows the voltage level which the total output voltage of the inverter circuit by Embodiment 7 of this invention can take. It is a figure which shows the voltage level which the total output voltage of the inverter circuit by Embodiment 7 of this invention can take. It is a figure which shows the relationship of each capacitor voltage with respect to the DC power supply voltage of the power converter device by Embodiment 7 of this invention. It is a figure which shows the relationship of each capacitor
- FIG. 1 is a main circuit configuration diagram of a power conversion device according to Embodiment 1 of the present invention.
- the AC side of an inverter circuit 20 is connected in series to the output of a DC power source 1 made of a solar cell or the like.
- the inverter circuit 20 is configured by connecting the AC sides of the first and second single-phase inverters 20a and 20b in series, and the DC power source 1 uses the sum of the outputs of the single-phase inverters 20a and 20b as the output of the inverter circuit 20.
- the first and second single-phase inverters 20a and 20b constituting the inverter circuit 20 are composed of semiconductor switch elements 21 to 24 and 31 to 34 and first and second capacitors 25 and 35 as DC voltage sources.
- the semiconductor switch elements 21 to 24 and 31 to 34 are IGBTs (Insulated Gate Bipolar Transistors) in which diodes are connected in antiparallel, and MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) in which diodes are built in between source and drain. Etc. are used.
- a short-circuit switch 4 and a rectifier diode 5 as an element for switching between conduction / non-conduction are connected to the subsequent stage of the inverter circuit 20, and the cathode side of the rectification diode 5 is connected to the positive electrode of the smoothing capacitor 6 in the output stage.
- the connection point between the shorting switch 4 and the anode of the rectifier diode 5 is connected to the AC output line at the rear stage of the inverter circuit 20, and the other end of the shorting switch 4 is connected to the negative electrode of the smoothing capacitor 6.
- the shorting switch 4 may be a semiconductor switch element such as IGBT or MOSFET, or a mechanical switch.
- FIG. 6 shows the case where the voltage of the DC power supply 1 is 180 V
- FIGS. FIG. 10 shows a list of these operations.
- S is a short-circuit switch 4
- PV is a DC power source 1
- C 1 and C 2 are first and second capacitors 25 and 35
- Co is a smoothing capacitor 6
- a voltage discharged from C 1 and C 2 is a positive voltage.
- C1 and C2 are charged with negative voltages.
- the set voltage of the first capacitor 25 in the first single-phase inverter 20a is 60V
- the set voltage of the second capacitor 35 in the second single-phase inverter 20b is 120V.
- Case B1 when the voltage of the DC power supply 1 is 120 V, in Case B1, the shorting switch 4 is turned on to bypass the smoothing capacitor 6, and the inverter circuit 20 charges DC power.
- the second single-phase inverter 20b is controlled so as to charge the second capacitor 35, and the first single-phase inverter 20a has an output of zero.
- the short circuit switch 4 is turned off, and the inverter circuit 20 discharges the DC power.
- the second single-phase inverter 20 b is controlled so as to discharge the second capacitor 35, the sum of the voltages of the DC power supply 1 and the second capacitor 35 becomes 240 V, and the smoothing capacitor 6 is connected via the rectifier diode 5. Is increased to 240V.
- CaseC3 shown in FIG. 9 may be used instead of the above CaseC2. That is, when the semiconductor switch elements 21, 24, 32, and 33 in the inverter circuit 20 are turned on with the shorting switch 4 turned off, the current flowing from the DC power supply 1 flows through the following path, The smoothing capacitor 6 is charged with a DC voltage of 240 V with a voltage obtained by subtracting the voltage of the first capacitor 25 from the sum of the voltage of the capacitor 2 and the voltage of the capacitor 35, and the boosting operation is completed.
- DC power source 1 ⁇ semiconductor switch element 21 ⁇ first capacitor 25 ⁇ semiconductor switch element 24 ⁇ semiconductor switch element 32 ⁇ second capacitor 35 ⁇ semiconductor switch element 33 ⁇ rectifier diode 5 ⁇ smoothing capacitor 6
- the shorting switch 4 is turned on to bypass the smoothing capacitor 6, and the inverter circuit 20 charges DC power.
- the first and second single-phase inverters 20a and 20b are controlled so that the first and second capacitors 25 and 35 are charged.
- the short circuit switch 4 is turned off, and the inverter circuit 20 discharges the DC power.
- the first single-phase inverter 20a is controlled so as to discharge the first capacitor 25, and the second single-phase inverter 20b has an output of zero.
- the second single-phase inverter 20b is controlled so as to discharge the second capacitor 35. In this case, the first single-phase inverter 20a charges the first capacitor 25. Discharge DC power.
- the inverter circuit 20 is output-controlled according to the voltage of the DC power supply 1 so that the output voltages of the first and second single-phase inverters 20a and 20b are superimposed on the voltage of the DC power supply 1, and the smoothing capacitor 6 has a desired value. Output voltage.
- the inverter circuit 20 is controlled so as to switch the charging / discharging operation of the DC voltage.
- the shorting switch 4 when the shorting switch 4 is in the ON state, the DC voltage of the inverter circuit 20 can be charged by bypassing the smoothing capacitor 6, and the charged energy is transferred to the smoothing capacitor 6 when the shorting switch 4 is in the OFF state. Can be used for discharging. Therefore, the short-circuit switch 4 and the semiconductor switch element in the inverter circuit 20 do not require high-frequency switching, and the inverter circuit 20 can make the voltage handled by switching lower than the set voltage of the smoothing capacitor 6. As described above, by using low-frequency switching and making the voltages of the capacitors 25 and 35 lower than the set voltage of the smoothing capacitor 6, reduction of power loss and noise and reduction in size and weight of the device configuration were promoted. A power converter can be realized.
- the inverter circuit 20 showed what was comprised by the two single phase inverters 20a and 20b, as shown in FIG. 11, only one single phase inverter 20a is an inverter circuit.
- the inverter circuit 20 may be configured by connecting in series the AC side of three or more single-phase inverters 20a to 20c as shown in FIG.
- reference numeral 20c denotes a third single-phase inverter, which includes semiconductor switch elements 41 to 44 and a third capacitor 45 as a DC voltage source.
- the same effect as described above can be obtained regardless of the number of single-phase inverters in the inverter circuit 20.
- the output voltage of the inverter circuit 20 is divided into a plurality of parts. The voltage handled by the single-phase inverter can be lowered, and the switching loss is further reduced.
- first and second single-phase inverters constituting the inverter circuit 20 include semiconductor switches 22, 23, 32, 33 and diodes 26, 27, 36, 37, and first, second, as shown in FIG.
- the first and second single-phase inverters 30a and 30b configured by full-bridge inverters including the capacitors 25 and 35 may be used, and similar effects can be obtained.
- the cathode side of the rectifier diode 5 is connected to the positive electrode of the smoothing capacitor 6 at the output stage.
- the rectifier diode 5 is connected to the negative electrode side of the smoothing capacitor 6 and the negative electrode is connected to the rectifier diode 5. It may be arranged so as to be connected to the anode side, and the same operation as in the above embodiment can be obtained.
- the inverter circuit 20 is connected to the positive electrode of the smoothing capacitor 6 at the output stage.
- the inverter circuit 20 is connected to the negative electrode side of the smoothing capacitor 6 as shown in FIG. You may arrange as follows. Furthermore, as shown in FIG. 15, using two inverter circuits 20X and 20Y, the inverter circuit 20X is connected to the positive side of the smoothing capacitor 6 at the output stage, and the inverter circuit 20Y is connected to the negative side of the smoothing capacitor 6 at the output stage. It may be arranged as described.
- a switch 9 may be used in place of the rectifier diode 5 as an element for switching between conduction and non-conduction.
- This switch 9 is controlled so that the on / off state of the shorting switch 4 is reversed. That is, when the shorting switch 4 is on, the switch 9 is off, bypassing the smoothing capacitor 6 and charging the DC voltage of the inverter circuit 20. The charged energy is used for discharging the smoothing capacitor 6 with the switch 9 turned on when the shorting switch 4 is turned off.
- the switch 9 may be a semiconductor switch element such as IGBT or MOSFET, or a mechanical switch.
- FIG. 1 one end of the shorting switch 4 is connected to the AC output line of the inverter circuit 20, but in this second embodiment, one end of the shorting switch 4a is connected to the inverter circuit as shown in FIG. 20 is connected to the negative electrode of the second capacitor 35 of the half-bridge single-phase inverter 30 that is the last-stage single-phase inverter (in this case, the second single-phase inverter).
- the other end of the short-circuit switch 4a is connected to the negative electrode of the smoothing capacitor 6 as in the first embodiment.
- the rectifier diode 5 has an anode connected to the positive electrode of the second capacitor 35 and a cathode connected to the positive electrode of the smoothing capacitor 6.
- the control of the inverter circuit 20 and the shorting switch 4a is the same as in the first embodiment, but the number of elements through which current passes is the same when the shorting switch 4a is on / off. Can be reduced, conduction loss can be reduced, and the conversion efficiency of the entire power conversion device can be improved.
- the single-phase inverter at the last stage can be configured by the half-bridge single-phase inverter 30, and the circuit configuration can be simplified.
- the last-stage single-phase inverter may be constituted by a half-bridge single-phase inverter 40 including a semiconductor switch 32, a diode 36, and a second capacitor 35, and the same effect can be obtained.
- the first single-phase inverter 30 a is configured by a full-bridge inverter including semiconductor switches 22 and 23, diodes 26 and 27, and a first capacitor 25.
- the single-phase inverter at the last stage is composed of the half-bridge single-phase inverters 30 and 40.
- it may be a full-bridge single-phase inverter 30b.
- the short-circuit switch 4a by connecting one end of the short-circuit switch 4a to the negative electrode of the second capacitor 35, when the short-circuit switch 4a is turned on, the number of elements through which current passes can be reduced, and the effect of reducing conduction loss can be obtained. It is done.
- the main circuit of the power conversion device has the same configuration as that shown in FIG. 1 of the first embodiment.
- the input voltage detector 10 that detects the voltage Vin obtained from the DC power supply 1 and the first and second voltages that detect the voltages of the first and second capacitors 25 and 35 in the inverter circuit 20.
- capacitor voltage detectors 11 and 12 an output voltage detector 13 that detects the voltage Vo of the output-side smoothing capacitor 6, and a control unit 14.
- the control unit 14 receives the output signals of the voltage detectors 10 to 13 and controls the switching elements and the short-circuit switch 4 in the inverter circuit 20.
- the control unit 14 is constituted by, for example, a microcomputer or a digital signal processor.
- the control unit 14 monitors the input voltage Vin obtained from the DC power source 1 (S1), and determines whether or not the input voltage Vin exceeds a predetermined input voltage set value V1 (S2). When the input voltage Vin exceeds the voltage V1, the main circuit is activated. Next, the capacitor voltage Vc in the inverter circuit 20 is monitored (S3), and it is determined whether or not the capacitor voltage Vc exceeds the capacitor voltage set value V2 (S4).
- the capacitor voltage Vc is the voltages Vc1 and Vc2 of the first and second capacitors 25 and 35, respectively, and the capacitor voltage setting value V2 is also set for the first and second capacitors 25 and 35, respectively. Two voltage values V21 and V22.
- the respective voltages Vc1, Vc2 of the first and second capacitors 25, 35 are compared with the respective voltage setting values V21, V22, and until both voltages Vc1, Vc2 exceed the respective voltage setting values V21, V22.
- the charging mode is determined (S5), a drive signal for turning on the shorting switch 4 is output, and a drive signal is output to each switching element in the inverter circuit 20 in accordance with the input voltage Vin from the DC power source 1. Then, the DC power in the inverter circuit 20 is charged (S6).
- step S4 the voltages Vc1 and Vc2 of the first and second capacitors 25 and 35 are compared with the voltage setting values V21 and V22, and both voltages Vc1 and Vc2 are compared with the voltage setting values V21 and V22. Is exceeded, that is, when the charging of the first and second capacitors 25 and 35 is completed, the voltage Vo of the smoothing capacitor 6 on the output side is monitored (S7). Then, it is determined whether or not the voltage Vo of the smoothing capacitor exceeds the output voltage set value V3 that is the target voltage value (S8). When the smoothing capacitor voltage Vo is equal to or less than the output voltage set value V3, it is determined that the discharge mode is set (S9).
- the battery is discharged (S10) and charged until the voltage Vo of the smoothing capacitor 6 reaches the output voltage set value V3.
- the control is performed so that the capacitors 25 and 35 in the inverter circuit 20 are fully charged and then discharged, even if the DC power source 1 has a large voltage fluctuation like a solar cell.
- the voltage of the smoothing capacitor 6 can be easily and reliably controlled.
- the main circuit configuration shown in FIG. 1 of the first embodiment is used.
- any of the various main circuit configurations shown in the first and second embodiments is used. Can be controlled similarly, and the same effect can be obtained.
- Embodiment 4 FIG.
- the short-circuit switch 4 when charging of the capacitors 25 and 35 in the inverter circuit 20 is completed, the short-circuit switch 4 is turned off to switch the charging / discharging of the DC power in the inverter circuit 20.
- charging / discharging is switched at the following timing.
- the shorting switch 4 is turned off from on under the condition that the voltage Vin from the DC power source 1 and the output voltage of the inverter circuit 20 (the sum of the output voltages of the single-phase inverters 20a and 20b) are opposite in polarity and substantially equal in magnitude.
- charging / discharging of DC power in the inverter circuit 20 is switched.
- the charging voltage during charging of the inverter circuit 20 as a whole is substantially equal to the voltage Vin from the DC power supply 1, and the sum of the output voltages of the single-phase inverters 20a and 20b, which is the output voltage of the inverter circuit 20, is superimposed on the voltage Vin. Then, when the voltage becomes approximately zero, the shorting switch 4 is turned off from on. At this switching timing, almost no current flows, so that zero current switching is possible and switching loss and noise can be reduced. For this reason, highly efficient and reliable control is realizable.
- FIG. 22 is a main circuit configuration diagram of a power conversion device according to Embodiment 5 of the present invention. Since this main circuit configuration is the same as the main circuit configuration of the first embodiment, description thereof is omitted.
- FIG. 24 shows the case where the voltage Vin is 60V ⁇ Vin ⁇ 80V
- FIG. 25 shows the case where the voltage Vin of the DC power supply 1 is 80V ⁇ Vin ⁇ 120V
- FIG. 25 shows the case where the voltage Vin of the DC power supply 1 is 120V ⁇ Vin ⁇ 180V. 26.
- FIG. 27 shows a list of these operations. 27 shows the case where the voltage Vin is 50V, 60V, 70V, 80V, 90V, 105V, 120V, 135V, 150V, 165V, and 180V for convenience.
- Vbit1 is the output voltage of the first single-phase inverter 20a
- Vbit2 is the output voltage of the second single-phase inverter 20b
- Va is the first
- 2 is a voltage obtained by superimposing the output voltages Vbit1 and Vbit2 of the single-phase inverters 20a and 20b, that is, Vin + Vbit1 + Vbit2.
- S (ON / OFF) is a signal indicating the on / off state of the short-circuit switch 4.
- a plurality of control modes including a combination of output control of the first and second single-phase inverters 20a and 20b and on / off control of the shorting switch 4 are set in advance, and the voltage Vin of the DC power source 1 is set.
- the control mode is selected and switched accordingly.
- control modes A to D are used.
- charging / discharging of the DC power of the inverter circuit 20 by a predetermined control operation of the power conversion device is repeated at a constant cycle.
- a plurality of control operations by different control operations are performed.
- a period hereinafter referred to as a section
- the power conversion device When 40V ⁇ Vin ⁇ 60V shown in FIG. 23, the power conversion device operates in the control mode A.
- this control mode A one cycle of the DC power charging / discharging operation of the inverter circuit 20 is divided into first to fourth sections, and in the first to third sections, the short-circuit switch 4 is turned on and smoothed. The capacitor 6 is bypassed, and the inverter circuit 20 charges DC power.
- the fourth section the short-circuit switch 4 is turned off, the inverter circuit 20 discharges DC power, and the smoothing capacitor 6 is charged via the voltage rectifier diode 5 with a voltage sum superimposed on the voltage of the DC power supply 1.
- the voltage Vo of the smoothing capacitor 6 is an output voltage.
- the step-up ratio is 4.
- the first capacitor 25 is charged to 60 V in the first period, and the sum of the voltage Vin of the DC power source 1 and the voltage of the first capacitor 25 is calculated in the second period.
- the second capacitor 35 is 120V.
- the first capacitor 25 is charged again to 60V.
- the shorting switch 4 is turned off, and the sum of the voltage Vin of the DC power source 1, the voltage of the first capacitor 25, and the voltage of the second capacitor 35 is passed through the rectifier diode 5.
- the smoothing capacitor 6 is charged with a DC voltage of 240V.
- the power conversion device When 60V ⁇ Vin ⁇ 80V shown in FIG. 24, the power conversion device operates in the control mode B.
- this control mode B one cycle in the DC power charging / discharging operation of the inverter circuit 20 is divided into the first to third sections, and in the first and second sections, the short-circuit switch 4 is turned on and smoothed. The capacitor 6 is bypassed, and the inverter circuit 20 charges DC power.
- the third section the short circuit switch 4 is turned off and the inverter circuit 20 discharges DC power, and charges the smoothing capacitor 6 via the rectifier diode 5 with a voltage sum superimposed on the voltage of the DC power supply 1.
- the boost ratio is 3.
- the first capacitor 25 is charged up to 80V in the first period, and the sum of the voltage Vin of the DC power supply 1 and the voltage of the first capacitor 25 in the second period.
- the second capacitor 35 to 160V.
- the short-circuit switch 4 is turned off, and the DC voltage 240 V is applied to the smoothing capacitor 6 via the rectifier diode 5 by the sum of the voltage Vin of the DC power supply 1 and the voltage of the second capacitor 35. Charge.
- the power converter operates in the control mode C.
- this control mode C one cycle in the DC power charging / discharging operation of the inverter circuit 20 is divided into first and second sections.
- the shorting switch 4 is turned on and the smoothing capacitor 6 is turned on. Bypassing, the inverter circuit 20 charges DC power.
- the short-circuit switch 4 is turned off, and the inverter circuit 20 discharges DC power, and charges the smoothing capacitor 6 via the rectifier diode 5 with a voltage sum superimposed on the voltage of the DC power supply 1.
- the boost ratio is 2.
- the first capacitor 25 is charged to 40V and the second capacitor 35 is charged to 80V in the first section.
- the shorting switch 4 is turned off, and the sum of the voltage Vin of the DC power source 1, the voltage of the first capacitor 25, and the voltage of the second capacitor 35 is passed through the rectifier diode 5. 6 is charged with a DC voltage of 240V.
- the power conversion device When 120V ⁇ Vin ⁇ 180V shown in FIG. 26, the power conversion device operates in the control mode D.
- this control mode D one cycle in the DC power charging / discharging operation of the inverter circuit 20 is divided into first to fourth sections, and in the first section, the shorting switch 4 is turned on and the smoothing capacitor 6 is turned on. Bypassing, the inverter circuit 20 charges DC power.
- the short-circuit switch 4 is turned off and the inverter circuit 20 discharges the DC power, and the smoothing capacitor 6 is charged via the rectifier diode 5 with the voltage sum superimposed on the voltage of the DC power supply 1.
- the boost ratio is 1.3.
- the shorting switch 4 is turned off.
- the DC voltage 240V is charged to the smoothing capacitor 6 through the rectifier diode 5 by the sum of the voltage Vin of the DC power supply 1 and the voltage of the first capacitor 25.
- the first capacitor 25 is recharged to 60 V by the sum of the voltage Vin of the DC power source 1 and the voltage of the second capacitor 35, and the DC voltage 240 V is applied to the smoothing capacitor 6 via the rectifier diode 5.
- the smoothing capacitor 6 is charged with the DC voltage 240 V via the rectifier diode 5 by the sum of the voltage Vin of the DC power source 1 and the voltage of the first capacitor 25.
- the control modes A to D are switched and used in accordance with the voltage Vin of the DC power supply 1, thereby boosting the voltage using charging / discharging of DC power in the inverter circuit 20.
- the step-up ratio is determined for each control mode, and the control modes A to D are selected such that the step-up ratio increases as the voltage Vin decreases, and the output voltage Vo is boosted to a range of 160V ⁇ Vo ⁇ 240V. Further, the control modes A to D are selected so that the fluctuation of the output voltage Vo is suppressed even if the voltage Vin of the DC power supply 1 fluctuates. In this way, the boost ratio is selected by determining and switching the control modes A to D according to the voltage Vin of the DC power supply 1.
- each control mode A to D the output of the first and second single-phase inverters 20a and 20b is controlled so that the first and second capacitors 25 and 35 balance the power transfer by the charge / discharge operation.
- one cycle in each control mode includes a plurality of sections, but each single-phase inverter is included in one cycle except when each single-phase inverter 20a, 20b does not generate a voltage through one cycle.
- 20a and 20b include both sections in which a positive voltage is output and sections in which a negative voltage is output. Further, as shown in FIGS.
- the section length is equal between the sum of the sections in which each single-phase inverter 20a, 20b outputs a positive voltage and the sum of the sections in which a negative voltage is output within one cycle. For this reason, in the first and second capacitors 25 and 35, the power transfer by the charge / discharge operation is reliably balanced within one cycle.
- the inverter circuit 20 since the DC / DC conversion is performed using the charging / discharging of the DC power in the inverter circuit 20 as described above, a large capacity reactor is not required.
- the shorting switch 4 when the shorting switch 4 is switched on / off, the inverter circuit 20 is controlled so as to switch the charging / discharging operation of the DC voltage. For this reason, the short-circuit switch 4 and the semiconductor switch elements 21 to 24 and 31 to 34 in the inverter circuit 20 do not require high-frequency switching, and the inverter circuit 20 sets the voltage handled by switching higher than the set voltage of the smoothing capacitor 6. Can be lowered. Therefore, it is possible to realize a power conversion device in which reduction of power loss and noise and reduction in size and weight of the device configuration are promoted.
- control modes A to D each of which is composed of a combination of output control of the first and second single-phase inverters 20a and 20b and on / off control of the short-circuit switch 4, each having a different step-up ratio, are set in advance.
- the control modes A to D are selected and switched according to the voltage Vin of the power source 1. Then, the respective output voltages of the first and second single-phase inverters 20 a and 20 b are superimposed on the voltage Vin of the DC power supply 1, and a desired voltage is output to the smoothing capacitor 6.
- the plurality of control modes A to D are set by a combination of output control of the plurality of single-phase inverters 20a and 20b and on / off control of the shorting switch 4, and a wide range of boost ratios determined for each control mode is set. can do. In this case, four step-up ratios from 1.3 to 4 are set. For this reason, the step-up ratio can be selected over a wide range, voltage fluctuation of the output voltage Vo can be suppressed with respect to a wide range of input voltage (voltage Vin), and a desired output voltage Vo can be obtained.
- first and second single-phase inverters 20a and 20b are output-controlled so that the first and second capacitors 25 and 35 balance the power transfer by the charge / discharge operation. It is not necessary to supply or control the capacitors 25 and 35 from the outside, and it is not necessary to install a DC / DC converter.
- the solar cell has a voltage range that varies greatly depending on the number of direct-current power supplies that can be installed outdoors, in addition to conditions such as the amount of solar radiation and temperature. For this reason, the voltage Vin input to the power converter varies widely. .
- this embodiment since a desired output voltage Vo can be obtained with respect to a wide range of input voltages (voltage Vin), this embodiment is particularly effective when a solar cell is used for the DC power supply 1.
- FIG. 29 is a diagram showing output characteristics of the solar cell.
- MPPT Maximum Power Point Tracking
- control is generally used as a method of making maximum use of the electric power obtained from the solar cell. In this case, it is necessary to maintain the voltage at the maximum output point Vpmax. As shown in FIG. 29, the voltage at the maximum output point Vpmax fluctuates, but in this embodiment in which a desired output voltage Vo is obtained for a wide range of input voltages, this is used in combination with the solar cell MPPT control. Furthermore, the effective use of electric power can be promoted.
- the duty ratio of the positive / negative voltage output of the first and second single-phase inverters 20a and 20b can be adjusted without changing the on / off control of the short-circuit switch 4 that determines charging / discharging of the entire inverter circuit 20. Also good.
- the first and second single-phase inverters 20a and 20b can adjust the voltage ratio due to the loss component and dead time included in the circuit. Even when the voltage ratio between the first and second capacitors 25 and 35 deviates from the set value, adjustment is possible. For example, in the control by the control mode A shown in FIG.
- the positive voltage output period of the first single-phase inverter 20a and the negative voltage output period of the second single-phase inverter 20b are included in the charging period of the inverter circuit 20, respectively.
- the voltage of the first capacitor 25 can be adjusted low (high), and the voltage of the second capacitor 35 can be adjusted high (low) to adjust the voltage ratio.
- the ratio between the set voltage Vc1 of the first capacitor 25 in the first single-phase inverter 20a and the set voltage Vc2 of the second capacitor 35 in the second single-phase inverter 20b is 1: 2. Although fixed, it is variable in the sixth embodiment.
- the main circuit configuration is the same as that of the fifth embodiment.
- the power conversion apparatus is preset with a plurality of control modes including a combination of output control of the first and second single-phase inverters 20a and 20b and on / off control of the short-circuit switch 4. Then, the control mode is selected and switched according to the voltage Vin of the DC power supply 1.
- control modes A to E are used.
- the same control modes A to D as in the fifth embodiment and the control mode E described later are used.
- the control mode E is used and the voltage of the DC power supply 1 is set.
- the control mode D is used when Vin is 160V ⁇ Vin ⁇ 180V.
- Control modes A to C are used in the same manner as in the fifth embodiment.
- control mode E when 120V ⁇ Vin ⁇ 160V will be described with reference to FIG.
- this control mode E one cycle in the DC power charging / discharging operation of the inverter circuit 20 is divided into first to third three sections.
- the shorting switch 4 is turned on and the smoothing capacitor 6 is turned on. Bypassing, the inverter circuit 20 charges DC power.
- the short-circuit switch 4 is turned off and the inverter circuit 20 discharges the DC power, and charges the smoothing capacitor 6 via the rectifier diode 5 with the voltage sum superimposed on the voltage of the DC power supply 1. To do.
- the ratio between the set voltage Vc1 of the first capacitor 25 and the set voltage Vc2 of the second capacitor 35 is 1: 1, and the boost ratio is 1.5.
- the boost ratio is 1.5.
- the first and second capacitors 25 and 35 are charged up to 80V in the first period.
- the shorting switch 4 is turned off.
- the DC voltage 240V is charged to the smoothing capacitor 6 through the rectifier diode 5 by the sum of the voltage Vin of the DC power supply 1 and the voltage of the first capacitor 25.
- the DC voltage 240V is charged to the smoothing capacitor 6 through the rectifier diode 5 by the sum of the voltage Vin of the DC power supply 1 and the voltage of the second capacitor 35. Also in this case, the output of the first and second single-phase inverters 20a and 20b is controlled so that the first and second capacitors 25 and 35 balance the power transfer by the charge / discharge operation.
- FIG. 31 is a diagram illustrating an operation in the control mode D when 160V ⁇ Vin ⁇ 180V. Except for the voltage range of the DC power source 1, it is the same as that of the fifth embodiment, and the ratio of the set voltage Vc 1 of the first capacitor 25 to the set voltage Vc 2 of the second capacitor 35 is 1: 2, and the boost ratio is 1 .3. A list of these operations is shown in FIG. For convenience, FIG. 32 shows the case where the voltage Vin is 50V, 60V, 70V, 80V, 90V, 105V, 120V, 130V, 140V, 150V, 160V, 165V, 180V.
- FIG. 33 shows the relationship between the voltage Vc1 and Vc2 of the first and second capacitors 25 and 35 and the voltage Vo (output voltage) of the smoothing capacitor 6 on the output side with respect to the voltage Vin of the DC power supply 1.
- the ratio shown in the figure is the voltage ratio of the first and second capacitors 25 and 35, and indicates that the ratio is 1: 1 only when 120V ⁇ Vin ⁇ 160V, and 1: 2 otherwise.
- the voltage fluctuation rate of the output voltage Vo falls within a smaller range than that in the fifth embodiment.
- the voltage is boosted using charging / discharging of DC power in the inverter circuit 20.
- the step-up ratio is determined for each control mode, and the control is performed so that the step-up ratio increases as the voltage Vin decreases, and the fluctuation of the output voltage Vo is suppressed even if the voltage Vin of the DC power supply 1 fluctuates.
- Modes A to E are selected and the output voltage Vo is boosted to a range of 160V ⁇ Vo ⁇ 240V. That is, the boost ratio is selected according to the voltage Vin, and the control modes A to E are determined based on the selected boost ratio to control the power converter.
- the same effect as in the fifth embodiment is obtained, and the voltage ratio of the first and second capacitors 25 and 35 is made variable, so that the number of control modes and boost ratios that can be set can be increased.
- the voltage ratios of the first and second capacitors 25 and 35 are two types of 1: 1 and 1: 2, and five step-up ratios are set by the control modes A to E. For this reason, it becomes possible to select a larger step-up ratio, and the voltage fluctuation of the output voltage Vo can be further suppressed.
- the rectifier diode 5 as a semiconductor element for determining conduction / non-passage is connected to the subsequent stage of the inverter circuit 20, but as shown in FIG. 16 of the first embodiment.
- a semiconductor switch 9 may be used instead of the rectifier diode 5.
- the semiconductor switch 9 is controlled so that the on / off state of the shorting switch 4 is reversed. That is, when the shorting switch 4 is on, the semiconductor switch 9 is off, and the DC voltage of the inverter circuit 20 is charged by bypassing the smoothing capacitor 6.
- the charged energy is used for discharging the smoothing capacitor 6 with the semiconductor switch 9 turned on when the shorting switch 4 is turned off.
- the first and second single-phase inverters constituting the inverter circuit 20 may be used by replacing the semiconductor switch elements 21, 24, 31, and 34 with diodes. The effect is obtained. Also, the cathode side of the rectifier diode 5 is connected to the positive electrode of the smoothing capacitor 6 at the output stage, but the rectifier diode 5 is connected to the negative electrode side of the smoothing capacitor 6 and the negative electrode is connected to the anode side of the rectifier diode 5. You may arrange so that.
- one end of the shorting switch 4 is connected to the AC output line of the inverter circuit 20, but as shown in the second embodiment, one end of the shorting switch 4a is You may connect to the negative electrode of the 2nd capacitor
- the second single-phase inverter 30b in the last stage may be a half-bridge single-phase inverter, and the other end of the short-circuit switch 4a is connected to the negative electrode of the smoothing capacitor 6 as in the fifth and sixth embodiments. Is done.
- the rectifier diode 5 has an anode connected to the positive electrode of the second capacitor 35 and a cathode connected to the positive electrode of the smoothing capacitor 6.
- the single-phase inverter 30b at the last stage can be configured with a half-bridge single-phase inverter, and the circuit configuration can be simplified.
- the inverter circuit 20 is configured by two single-phase inverters. However, the inverter circuit 20 may be configured by connecting the AC sides of three or more single-phase inverters in series.
- FIG. 34 is a main circuit configuration diagram of the power conversion device according to the seventh embodiment of the present invention. As shown in FIG. 34, the AC side of the inverter circuit 20 is connected in series to the output of the DC power source 1 composed of a solar cell or the like. The inverter circuit 20 is configured by connecting the AC sides of the first to third single-phase inverters 31a to 31c in series.
- the sum of the outputs of the single-phase inverters 31a to 31c is used as the output of the inverter circuit 20 for the DC power source 1. Is superimposed on the DC voltage from The first and second single-phase inverters 31a and 31b constituting the inverter circuit 20 include semiconductor switch elements 22, 23, 32 and 33, diodes 26, 27, 36 and 37, and first and second DC voltage sources. Capacitors 25 and 35.
- the third single-phase inverter 31 c at the last stage is a half-bridge single-phase inverter composed of a semiconductor switch element 42, a diode 46, and a third capacitor 45.
- the smoothing capacitor 6 is connected to the subsequent stage of the inverter circuit 20 via the rectifier diode 5, and the short-circuit switch 4 a is connected between the negative electrode of the third capacitor 45 and the negative electrode of the smoothing capacitor 6.
- the power conversion device has a plurality of control modes including a combination of output control of the first to third single-phase inverters 31a to 31c and on / off control of the shorting switch 4a.
- the control mode is set in advance and switched according to the voltage Vin of the DC power supply 1.
- the maximum voltage Vc (N) and the minimum voltage Vc (1) satisfy the following relational expression.
- Vc (N) ⁇ ( ⁇ k 1 to N ⁇ 1 Vc (k)) + Vc (1)
- the voltage ratio of the two capacitors 25 and 35 is 1: 2 or 1: 1, and both satisfy the above relational expression.
- the voltage ratio of the first to third capacitors 25, 35, and 45 is made variable, and a control mode is used in which two voltage ratios of “1: 1: 2” and “1: 1: 3” are used.
- FIG. 38 shows the relationship between the voltage Vc1, Vc2, Vc3 of each of the capacitors 25, 35, 45 and the voltage Vo (output voltage) of the smoothing capacitor 6 on the output side with respect to the voltage Vin of the DC power source 1 in this case.
- the ratio shown in the figure is the voltage ratio of each capacitor 25, 35, 45. As shown in FIG.
- step-up ratios are selected according to the voltage Vin and controlled by the corresponding seven control modes, and the voltage fluctuation rate of the output voltage Vo falls within a range of ⁇ 14%. Also in this case, by increasing the number of single-phase inverters 31a to 31c in series, more boost ratios can be selected and the voltage fluctuation rate can be reduced. Further, since the voltage ratio of the first to third capacitors 25, 35 and 45 is variable, the number of control modes and boost ratios that can be set can be increased. In this case, it is possible to select a larger number of step-up ratios than when fixed at either “1: 1: 2” or “1: 1: 3”, and the voltage fluctuation of the output voltage Vo can be further suppressed. .
- Embodiment 8 FIG. Next, a power conversion device according to embodiment 8 of the present invention will be described with reference to FIG.
- the main circuit of the power conversion device has the same configuration as that shown in FIG. 18 of the second embodiment.
- the input voltage detector 10 that detects the voltage Vin obtained from the DC power supply 1 and the hysteresis comparator that compares the voltage Vin with voltage thresholds Vth (Vth1, Vth2) as preset voltage determination values.
- Vth voltage thresholds
- Vth voltage thresholds
- the control unit 18 controls the semiconductor switch elements and the short-circuit switches 4a of the single-phase inverters 30a and 30b in the inverter circuit 20 in the control mode selected by the output signal of the mode selection unit 17.
- the mode selection unit 17 and the control unit 18 are composed of, for example, a microcomputer or a digital signal processor.
- Vth1 Vth2.
- Vth2 the control mode ⁇
- the control mode ⁇ is switched to decrease the boost ratio of the power converter.
- the operating point A2 in the figure shifts to B2.
- the control in the control mode ⁇ continues, and when the voltage decreases to the voltage threshold Vth1, the control mode ⁇ is switched to increase the boost ratio of the power converter.
- the operating point B1 in the figure shifts to A1.
- the voltage Vin input from the DC power source 1 fluctuates greatly
- the control mode is switched twice, but in the case of FIG. 42 showing the comparative example in which the hysteresis width is not provided, the control mode is switched four times for the same voltage Vin.
- the switching between the control mode C and the control mode E is performed when the input voltage Vin is 120 V (see FIG. 33), and Vth1 and Vth2 are provided with hysteresis widths of 5 to 10 V.
- the output can be stabilized by setting.
- the voltage threshold (Vth1, Vth2) for switching between the two control modes is shown.
- the voltage threshold for switching is provided according to the number of control modes.
- the hysteresis comparator may be configured as shown in FIG. As shown in FIG. 43, the comparator 19a, the voltage dividing resistors 19b to 19d, and the transistor 19e constitute a hysteresis comparator 19. In this case, the transistor 19e is turned on / off by the output signal of the comparator 19a, and the ratio of the voltage dividing resistors 19b to 19d for detecting the input voltage Vin is changed using the transistor 19e, thereby realizing the hysteresis characteristic. Yes.
- the comparator 19a can detect the voltage Vin using the voltage threshold Vth1 when the voltage Vin decreases and the voltage threshold Vth2 (> Vth1) when the voltage Vin increases. Also in this case, by providing the hysteresis width (Vth2-Vth1), the output of the power converter can be stabilized.
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Abstract
Description
以下、この発明の実施の形態1による電力変換装置について説明する。図1はこの発明の実施の形態1による電力変換装置の主回路構成図である。
図1に示すように、太陽電池等から成る直流電源1の出力に、インバータ回路20の交流側が直列接続される。インバータ回路20は、第1、第2の単相インバータ20a、20bの交流側を直列接続して構成され、各単相インバータ20a、20bの出力の総和を、インバータ回路20の出力として直流電源1からの直流電圧に重畳する。インバータ回路20を構成する第1、第2の単相インバータ20a、20bは、半導体スイッチ素子21~24、31~34および直流電圧源としての第1、第2のコンデンサ25、35から構成される。ここで、半導体スイッチ素子21~24、31~34は、ダイオードが逆並列に接続されたIGBT(Insulated Gate Bipolar Transistor)やソース・ドレイン間にダイオードが内蔵されたMOSFET(Metal Oxide Semiconductor Field Effect Transistor)などを用いる。
Hereinafter, a power converter according to
As shown in FIG. 1, the AC side of an
まず、図2に示すCaseA1のように、短絡用スイッチ4をオン状態で、インバータ回路20内の半導体スイッチ素子21、24、32、34をオンすると、直流電源1から流れ込む電流は以下の経路を流れ、第1のコンデンサ25を60Vまで充電する。
直流電源1→半導体スイッチ素子21→第1のコンデンサ25→半導体スイッチ素子24→半導体スイッチ素子32→半導体スイッチ素子34→短絡用スイッチ4→直流電源1 First, an operation when the voltage of the
First, when the
直流電源1→半導体スイッチ素子22→第1のコンデンサ25→半導体スイッチ素子23→半導体スイッチ素子31→第2のコンデンサ35→半導体スイッチ素子34→短絡用スイッチ4→直流電源1 Subsequently, as in Case A2 shown in FIG. 3, when the
直流電源1→半導体スイッチ素子22→第1のコンデンサ25→半導体スイッチ素子23→半導体スイッチ素子32→第2のコンデンサ35→半導体スイッチ素子33→整流ダイオード5→平滑コンデンサ6 Next, when the
この後、CaseA3において、短絡用スイッチ4をオフ状態として、インバータ回路20は直流電力を放電する。このとき、第1、第2のコンデンサ25、35を放電させるように第1、第2の単相インバータ20a、20bを制御すると、直流電源1および第1、第2のコンデンサ25、35の電圧の和は240Vとなり、整流ダイオード5を介して平滑コンデンサ6の電圧を240Vまで昇圧する。 Thus, when the voltage of the
Thereafter, in Case A3, the shorting
まず、図5に示すCaseB1のように、短絡用スイッチ4をオン状態で、インバータ回路20内の半導体スイッチ素子22、24、31、34をオンすると、直流電源1から流れ込む電流は以下の経路を流れ、第2のコンデンサ35を120Vまで充電する。
直流電源1→半導体スイッチ素子22→半導体スイッチ素子24→半導体スイッチ素子31→第2のコンデンサ35→半導体スイッチ素子34→短絡用スイッチ4→直流電源1 Second, the operation when the voltage of the
First, when the
直流電源1→半導体スイッチ素子22→半導体スイッチ素子24→半導体スイッチ素子32→第2のコンデンサ35→半導体スイッチ素子33→整流ダイオード5→平滑コンデンサ6 Next, when the
まず、図7に示すCaseC1のように、短絡用スイッチ4をオン状態で、インバータ回路20内の半導体スイッチ素子21、24、31、34をオンすると、直流電源1から流れ込む電流は以下の経路を流れ、第1のコンデンサ25を60Vまで、第2のコンデンサ35を120Vまで、それぞれ充電する。
直流電源1→半導体スイッチ素子21→第1のコンデンサ25→半導体スイッチ素子24→半導体スイッチ素子31→第2のコンデンサ35→半導体スイッチ素子34→短絡用スイッチ4→直流電源1 Third, the operation when the voltage of the
First, as in CaseC1 shown in FIG. 7, when the
直流電源1→半導体スイッチ素子22→第1のコンデンサ25→半導体スイッチ素子23→半導体スイッチ素子32→半導体スイッチ素子34→整流ダイオード5→平滑コンデンサ6 Next, when the
直流電源1→半導体スイッチ素子21→第1のコンデンサ25→半導体スイッチ素子24→半導体スイッチ素子32→第2のコンデンサ35→半導体スイッチ素子33→整流ダイオード5→平滑コンデンサ6 Further, CaseC3 shown in FIG. 9 may be used instead of the above CaseC2. That is, when the
そして、CaseC2、C3において、短絡用スイッチ4をオフ状態として、インバータ回路20は直流電力を放電する。CaseC2では第1のコンデンサ25を放電させるように第1の単相インバータ20aを制御し、第2の単相インバータ20bは出力0とする。CaseC3では第2のコンデンサ35を放電させるように第2の単相インバータ20bを制御し、この場合、第1の単相インバータ20aは第1のコンデンサ25を充電させるが、インバータ回路20全体としては直流電力を放電する。 Thus, when the voltage of the
In Cases C2 and C3, the
また、短絡用スイッチ4のオン/オフ切り換え時に、インバータ回路20は、直流電圧の充電/放電動作を切り替えるように制御される。即ち、短絡用スイッチ4がオン状態の時は、平滑コンデンサ6をバイパスしてインバータ回路20の直流電圧を充電でき、充電されたエネルギを、短絡用スイッチ4がオフ状態の時に平滑コンデンサ6への放電に使える。このため、短絡用スイッチ4およびインバータ回路20内の半導体スイッチ素子は、高周波スイッチングが不要であり、インバータ回路20は、スイッチングで扱う電圧を平滑コンデンサ6の設定電圧よりも低くできる。このように、低周波スイッチングを用いると共に、各コンデンサ25、35の電圧を平滑コンデンサ6の設定電圧より低くすることにより、電力損失およびノイズの低減化と装置構成の小型軽量化とが促進された電力変換装置が実現できる。 In this embodiment, since the DC / DC conversion is performed using the charging / discharging of the DC power in the
In addition, when the shorting
上記実施の形態1では短絡用スイッチ4の一端は、インバータ回路20の交流出力線に接続したが、この実施の形態2では、図17に示すように、短絡用スイッチ4aの一端は、インバータ回路20を構成する最後段の単相インバータ(この場合、第2の単相インバータ)であるハーフブリッジ単相インバータ30の第2のコンデンサ35の負極に接続する。短絡用スイッチ4aの他端は、上記実施の形態1と同様に、平滑コンデンサ6の負極に接続される。また、整流ダイオード5はアノードを第2のコンデンサ35の正極に接続し、カソードを平滑コンデンサ6の正極に接続する。
In the first embodiment, one end of the shorting
次に、この発明の実施の形態3による電力変換装置について図20に基づいて説明する。なお、電力変換装置の主回路は、上記実施の形態1の図1で示した同様の構成である。
図20に示すように、直流電源1から得られる電圧Vinを検出する入力電圧検出器10と、インバータ回路20内の第1、第2のコンデンサ25、35の各電圧を検出する第1、第2のコンデンサ電圧検出器11、12と、出力側の平滑コンデンサ6の電圧Voを検出する出力電圧検出器13と、制御部14とを備える。制御部14は、各電圧検出器10~13の各出力信号を入力としてインバータ回路20内のスイッチング素子および短絡用スイッチ4を制御する。なお、制御部14は例えばマイクロコンピュータやデジタルシグナルプロセッサなどで構成される。
Next, a power converter according to
As shown in FIG. 20, the
制御部14では直流電源1から得られた入力電圧Vinを監視し(S1)、入力電圧Vinが所定の入力電圧設定値V1を超えたかどうかを判断する(S2)。入力電圧Vinが電圧V1を超えると主回路を起動する。
次に、インバータ回路20内のコンデンサ電圧Vcを監視し(S3)、コンデンサ電圧Vcがコンデンサ電圧設定値V2を超えたかどうかを判断する(S4)。ここで、コンデンサ電圧Vcは、第1、第2のコンデンサ25、35の各電圧Vc1、Vc2であり、コンデンサ電圧設定値V2も第1、第2のコンデンサ25、35に対してそれぞれ設定される2つの電圧値V21、V22である。 The operation of the
The
Next, the capacitor voltage Vc in the
上記実施の形態3では、インバータ回路20内の各コンデンサ25、35の充電が完了すると、短絡用スイッチ4をオンからオフにしてインバータ回路20における直流電力の充放電を切り替えたが、この実施の形態では、以下のタイミングで充放電を切り替える。
直流電源1からの電圧Vinとインバータ回路20の出力電圧(各単相インバータ20a、20bの出力電圧の総和)とが、逆極性で大きさがほぼ等しい条件で、短絡用スイッチ4をオンからオフにしてインバータ回路20における直流電力の充放電を切り替える。
すなわち、インバータ回路20全体として充電時での充電電圧が直流電源1からの電圧Vinとほぼ等しく、インバータ回路20の出力電圧である各単相インバータ20a、20bの出力電圧の総和を電圧Vinに重畳すると電圧が概0となるとき、短絡用スイッチ4をオンからオフにする。この切り替えのタイミングでは、電流がほぼ流れていないため、零電流スイッチングが可能となりスイッチング損失およびノイズを低減できる。このため、高効率で信頼性の高い制御が実現できる。
In the third embodiment, when charging of the
The shorting
That is, the charging voltage during charging of the
次に、この発明の実施の形態5による電力変換装置について説明する。図22はこの発明の実施の形態5による電力変換装置の主回路構成図である。なお、この主回路構成は、上記実施の形態1の主回路構成と同様であるため、説明を省略する。
Next, a power conversion device according to
なお、第1の単相インバータ20aにおける第1のコンデンサ25の設定電圧Vc1と第2の単相インバータ20bにおける第2のコンデンサ35の設定電圧Vc2との比は(Vc1:Vc2)=(1:2)とする。 In FIG. 23 to FIG. 26, Vbit1 is the output voltage of the first single-
The ratio of the set voltage Vc1 of the
例えば、直流電源1の電圧Vinが60Vのとき、第1区間では第1のコンデンサ25を60Vまで充電し、第2区間では、直流電源1の電圧Vinと第1のコンデンサ25の電圧との和で第2のコンデンサ35を120Vまで充電する。第3区間では第1のコンデンサ25を60Vまで再度充電する。そして、第4区間では、短絡用スイッチ4をオフ状態にして、直流電源1の電圧Vinと第1のコンデンサ25の電圧および第2のコンデンサ35の電圧との和で、整流ダイオード5を介して平滑コンデンサ6に直流電圧240Vを充電する。 When 40V <
For example, when the voltage Vin of the
例えば、直流電源1の電圧Vinが80Vのとき、第1区間では第1のコンデンサ25を80Vまで充電し、第2区間では、直流電源1の電圧Vinと第1のコンデンサ25の電圧との和で第2のコンデンサ35を160Vまで充電する。そして、第3区間では、短絡用スイッチ4をオフ状態にして、直流電源1の電圧Vinと第2のコンデンサ35の電圧との和で、整流ダイオード5を介して平滑コンデンサ6に直流電圧240Vを充電する。 When 60V <
For example, when the voltage Vin of the
例えば、直流電源1の電圧Vinが120Vのとき、第1区間では第1のコンデンサ25を40Vまで充電すると共に、第2のコンデンサ35を80Vまで充電する。第2区間では、短絡用スイッチ4をオフ状態にして、直流電源1の電圧Vinと第1のコンデンサ25の電圧および第2のコンデンサ35の電圧との和で、整流ダイオード5を介して平滑コンデンサ6に直流電圧240Vを充電する。 In the case of 80V <Vin ≦ 120V shown in FIG. 25, the power converter operates in the control mode C. In this control mode C, one cycle in the DC power charging / discharging operation of the
For example, when the voltage Vin of the
例えば、直流電源1の電圧Vinが180Vのとき、第1区間では第1のコンデンサ25を60Vまで充電すると共に、第2のコンデンサ35を120Vまで充電する。第2~第4区間では、短絡用スイッチ4をオフ状態にする。まず、第2区間では、直流電源1の電圧Vinと第1のコンデンサ25の電圧との和で、整流ダイオード5を介して平滑コンデンサ6に直流電圧240Vを充電する。第3区間では、直流電源1の電圧Vinと第2のコンデンサ35の電圧との和で、第1のコンデンサ25を60Vまで再度充電すると共に、整流ダイオード5を介して平滑コンデンサ6に直流電圧240Vを充電する。そして、第4区間では、直流電源1の電圧Vinと第1のコンデンサ25の電圧との和で、整流ダイオード5を介して平滑コンデンサ6に直流電圧240Vを充電する。 When 120V <Vin ≦ 180V shown in FIG. 26, the power conversion device operates in the control mode D. In this control mode D, one cycle in the DC power charging / discharging operation of the
For example, when the voltage Vin of the
このように、直流電源1の電圧Vinに応じて制御モードA~Dを決定して切り替えることにより昇圧比を選択する。
直流電源1の電圧Vinに対する第1、第2のコンデンサ25、35の電圧Vc1、Vc2、および出力側の平滑コンデンサ6の電圧Vo(出力電圧)の関係を図28に示す。図28に示すように、出力電圧Voの電圧変動率は±20%の範囲に収まる。 As described above, the control modes A to D are switched and used in accordance with the voltage Vin of the
In this way, the boost ratio is selected by determining and switching the control modes A to D according to the voltage Vin of the
FIG. 28 shows the relationship between the voltage Vc1 and Vc2 of the first and
上述したように、各制御モードにおける1周期は複数の区間が含まれるが、各単相インバータ20a、20bが1周期内を通して電圧を発生させない場合を除いて、1周期内に、各単相インバータ20a、20bが正電圧を出力させる区間と負電圧を出力させる区間との双方の区間が含まれる。さらに、図23~26に示されるように、1周期内で、各単相インバータ20a、20bが正電圧を出力させる区間の総和と負電圧を出力させる区間の総和とでは、区間長が等しい。このため、第1、第2のコンデンサ25、35は、充放電動作による電力授受が1周期内で確実にバランスされる。 In each control mode A to D, the output of the first and second single-
As described above, one cycle in each control mode includes a plurality of sections, but each single-phase inverter is included in one cycle except when each single-
また、短絡用スイッチ4のオン/オフ切り換え時に、インバータ回路20は、直流電圧の充電/放電動作を切り替えるように制御される。このため、短絡用スイッチ4およびインバータ回路20内の半導体スイッチ素子21~24、31~34は、高周波スイッチングが不要であり、インバータ回路20は、スイッチングで扱う電圧を平滑コンデンサ6の設定電圧よりも低くできる。従って、電力損失およびノイズの低減化と装置構成の小型軽量化とが促進された電力変換装置が実現できる。 In this embodiment, since the DC / DC conversion is performed using the charging / discharging of the DC power in the
In addition, when the shorting
複数の制御モードA~Dは、複数の単相インバータ20a、20bの出力制御および短絡用スイッチ4のオン/オフ制御の組み合わせから設定されるもので、制御モード毎に決まる昇圧比を広範囲に設定することができる。この場合、1.3~4までの4段階の昇圧比が設定されている。このため、昇圧比が広範囲に選択でき、広範囲の入力電圧(電圧Vin)に対して出力電圧Voの電圧変動を抑制でき、所望の出力電圧Voが得られる。 Further, a plurality of control modes A to D, each of which is composed of a combination of output control of the first and second single-
The plurality of control modes A to D are set by a combination of output control of the plurality of single-
図29は、太陽電池の出力特性を示す図である。太陽電池から得られる電力を最大限利用する方法として、MPPT(Maximum Power Point Tracking)制御が一般的に用いられ、その場合、電圧を最大出力点Vpmaxに維持する必要がある。図29に示すように、最大出力点Vpmaxとなる電圧は変動するが、広範囲の入力電圧に対して所望の出力電圧Voが得られるこの実施の形態では、太陽電池のMPPT制御と併用することで、さらに電力の有効利用が促進できる。 In addition, the solar cell has a voltage range that varies greatly depending on the number of direct-current power supplies that can be installed outdoors, in addition to conditions such as the amount of solar radiation and temperature. For this reason, the voltage Vin input to the power converter varies widely. . In this embodiment, since a desired output voltage Vo can be obtained with respect to a wide range of input voltages (voltage Vin), this embodiment is particularly effective when a solar cell is used for the
FIG. 29 is a diagram showing output characteristics of the solar cell. MPPT (Maximum Power Point Tracking) control is generally used as a method of making maximum use of the electric power obtained from the solar cell. In this case, it is necessary to maintain the voltage at the maximum output point Vpmax. As shown in FIG. 29, the voltage at the maximum output point Vpmax fluctuates, but in this embodiment in which a desired output voltage Vo is obtained for a wide range of input voltages, this is used in combination with the solar cell MPPT control. Furthermore, the effective use of electric power can be promoted.
次に、この発明の実施の形態6について説明する。
上記実施の形態5では、第1の単相インバータ20aにおける第1のコンデンサ25の設定電圧Vc1と第2の単相インバータ20bにおける第2のコンデンサ35の設定電圧Vc2との比は1:2に固定したが、この実施の形態6では可変にする。なお、主回路構成は上記実施の形態5と同様である。
上記実施の形態5と同様に、電力変換装置は、第1、第2の単相インバータ20a、20bの出力制御および短絡用スイッチ4のオン/オフ制御の組み合わせから成る複数の制御モードが予め設定され、直流電源1の電圧Vinに応じて制御モードを選択して切り替える。この場合、制御モードA~Eが用いられる。上記実施の形態5と同様の制御モードA~Dと、後述する制御モードEとを用い、直流電源1の電圧Vinが120V<Vin≦160Vの場合に制御モードEを用い、直流電源1の電圧Vinが160V<Vin≦180Vの場合に制御モードDを用いる。制御モードA~Cについては、上記実施の形態5と同様に用いる。
Next, a sixth embodiment of the present invention will be described.
In the fifth embodiment, the ratio between the set voltage Vc1 of the
As in the fifth embodiment, the power conversion apparatus is preset with a plurality of control modes including a combination of output control of the first and second single-
例えば、直流電源1の電圧Vinが160Vのとき、第1区間では第1、第2のコンデンサ25、35をそれぞれ80Vまで充電する。第2、第3区間では、短絡用スイッチ4をオフ状態にする。まず、第2区間では、直流電源1の電圧Vinと第1のコンデンサ25の電圧との和で、整流ダイオード5を介して平滑コンデンサ6に直流電圧240Vを充電する。第3区間では、直流電源1の電圧Vinと第2のコンデンサ35の電圧との和で、整流ダイオード5を介して平滑コンデンサ6に直流電圧240Vを充電する。
この場合も、第1、第2の単相インバータ20a、20bは、第1、第2のコンデンサ25、35が充放電動作による電力授受をバランスさせるように出力制御される。 The operation in the control mode E when 120V <
For example, when the voltage Vin of the
Also in this case, the output of the first and second single-
また、これらの動作の一覧を表にしたものを図32に示した。なお、図32では便宜上、電圧Vinが50V、60V、70V、80V、90V、105V、120V、130V、140V、150V、160V、165V、180Vの場合を示した。 FIG. 31 is a diagram illustrating an operation in the control mode D when 160V <
A list of these operations is shown in FIG. For convenience, FIG. 32 shows the case where the voltage Vin is 50V, 60V, 70V, 80V, 90V, 105V, 120V, 130V, 140V, 150V, 160V, 165V, 180V.
この実施の形態では、上記実施の形態5と同様の効果を得ると共に、第1、第2のコンデンサ25、35の電圧比を可変としたため、設定可能な制御モードおよび昇圧比の数を多くできる。この場合、第1、第2のコンデンサ25、35の電圧比は、1:1と1:2との2種で、制御モードA~Eにより5段階の昇圧比が設定されている。このため、より多くの昇圧比を選択可能となり、出力電圧Voの電圧変動をより抑制することができる。 As described above, by switching between the control modes A to E according to the voltage Vin of the
In this embodiment, the same effect as in the fifth embodiment is obtained, and the voltage ratio of the first and
上記実施の形態5、6では、インバータ回路20は、2台の単相インバータで構成されたものを示したが、3台以上の単相インバータの交流側を直列接続して構成しても良い。
図34は、この発明の実施の形態7による電力変換装置の主回路構成図である。
図34に示すように、太陽電池等から成る直流電源1の出力に、インバータ回路20の交流側が直列接続される。インバータ回路20は、第1~第3の単相インバータ31a~31cの交流側を直列接続して構成され、各単相インバータ31a~31cの出力の総和を、インバータ回路20の出力として直流電源1からの直流電圧に重畳する。インバータ回路20を構成する第1、第2の単相インバータ31a、31bは、半導体スイッチ素子22、23、32、33、ダイオード26、27、36、37および直流電圧源としての第1、第2のコンデンサ25、35から構成される。また、最後段の第3の単相インバータ31cは、半導体スイッチ素子42とダイオード46と第3のコンデンサ45とから成るハーフブリッジ単相インバータで構成される。
またインバータ回路20の後段には整流ダイオード5を介して平滑コンデンサ6が接続され、第3のコンデンサ45の負極と平滑コンデンサ6の負極との間に短絡用スイッチ4aが接続される。
In the fifth and sixth embodiments, the
FIG. 34 is a main circuit configuration diagram of the power conversion device according to the seventh embodiment of the present invention.
As shown in FIG. 34, the AC side of the
Further, the smoothing
そして、複数(この場合3個)のコンデンサ25、35、45の電圧値Vc(k)のうち、最大となる電圧Vc(N)と最小となる電圧Vc(1)が次の関係式を満たすように制御する。
Vc(N)≦(Σk=1~N-1Vc(k))+Vc(1)
なお、上記実施の形態5、6においても、2個のコンデンサ25、35の電圧比は1:2あるいは1:1であり、いずれも上記関係式を満たしている。 Similar to the fifth and sixth embodiments, the power conversion device has a plurality of control modes including a combination of output control of the first to third single-
Of the voltage values Vc (k) of a plurality (three in this case) of
Vc (N) ≦ (Σ k = 1 to N−1 Vc (k)) + Vc (1)
In the fifth and sixth embodiments, the voltage ratio of the two
なお、各コンデンサ25、35、45の電圧比は、制御モードに応じて決まるため、制御モードを用いて電力変換装置を制御することにより各コンデンサ25、35、45の電圧比も制御できる。 When the voltages Vc1, Vc2, and Vc3 of the first to
In addition, since the voltage ratio of each capacitor |
このように、単相インバータ31a~31cの直列数を増やすことで、より多くの昇圧比を選択でき、電圧変動率を低減することができる。 When the voltage ratio of the first to
Thus, by increasing the number of single-
この場合も、単相インバータ31a~31cの直列数を増やすことで、より多くの昇圧比を選択でき、電圧変動率を低減することができる。また、第1~第3のコンデンサ25、35、45の電圧比を可変としたため、設定可能な制御モードおよび昇圧比の数を多くできる。この場合、「1:1:2」「1:1:3」のいずれか一方に固定した場合よりも、多くの昇圧比を選択可能となり、出力電圧Voの電圧変動をより抑制することができる。 Next, the voltage ratio of the first to
Also in this case, by increasing the number of single-
次に、この発明の実施の形態8による電力変換装置について図39に基づいて説明する。なお、電力変換装置の主回路は、上記実施の形態2の図18で示した同様の構成である。
図39に示すように、直流電源1から得られる電圧Vinを検出する入力電圧検出器10と、電圧Vinと予め設定した電圧判定値としての電圧閾値Vth(Vth1、Vth2)とを比較するヒステリシスコンパレータ16と、制御モードを切り替えるモード選択部17と、制御部18とを備える。制御部18は、モード選択部17の出力信号により選択された制御モードにて、インバータ回路20内の各単相インバータ30a、30bの半導体スイッチ素子および短絡用スイッチ4aを制御する。なお、モード選択部17と制御部18は例えばマイクロコンピュータやデジタルシグナルプロセッサなどで構成される。
Next, a power conversion device according to
As shown in FIG. 39, the
まず、電力変換装置が制御モードαで制御されて出力していたとする。入力される電圧Vinの増加に応じて出力電圧Voも上昇するが、電圧Vinが電圧閾値Vth2に達すると、電力変換装置の昇圧比を下げるように制御モードβに切り替える。これにより、図中の動作点A2からB2へと移行する。この後、電圧Vinが低下しても、制御モードβでの制御は継続し、電圧閾値Vth1まで電圧が低下すると、電力変換装置の昇圧比を上げるように制御モードαに切り替える。これにより、図中の動作点B1からA1へと移行する。 The control mode switching operation will be described below with reference to FIG. Note that Vth1 <Vth2.
First, it is assumed that the power converter is controlled and output in the control mode α. As the input voltage Vin increases, the output voltage Vo also increases. However, when the voltage Vin reaches the voltage threshold Vth2, the control mode β is switched to decrease the boost ratio of the power converter. As a result, the operating point A2 in the figure shifts to B2. Thereafter, even if the voltage Vin decreases, the control in the control mode β continues, and when the voltage decreases to the voltage threshold Vth1, the control mode α is switched to increase the boost ratio of the power converter. As a result, the operating point B1 in the figure shifts to A1.
例えば、上記実施の形態6では、制御モードCと制御モードEとの切り替えは、入力電圧Vinが120Vで切り替えるものとしたが(図33参照)、5~10Vのヒステリシス幅を設けてVth1、Vth2を設定することで出力が安定化できる。 Although the voltage Vin input from the
For example, in the sixth embodiment, the switching between the control mode C and the control mode E is performed when the input voltage Vin is 120 V (see FIG. 33), and Vth1 and Vth2 are provided with hysteresis widths of 5 to 10 V. The output can be stabilized by setting.
図43に示すように、コンパレータ19a、分圧抵抗19b~19dおよびトランジスタ19eでヒステリシスコンパレータ19を構成する。この場合、コンパレータ19aの出力信号でトランジスタ19eをオン/オフし、入力された電圧Vinを検出する分圧抵抗19b~19dの比をトランジスタ19eを用いて変化させることで、ヒステリシス特性を実現している。コンパレータ19aは、電圧Vinの下降時に電圧閾値Vth1、上昇時に電圧閾値Vth2(>Vth1)を用いて電圧Vinを検出することができる。この場合も、ヒステリシス幅(Vth2-Vth1)を設けることにより、電力変換装置の出力を安定化できる。 Further, the hysteresis comparator may be configured as shown in FIG.
As shown in FIG. 43, the
Claims (27)
- 半導体スイッチ素子と直流電圧源とをそれぞれ有した1以上の単相インバータの交流側を直列接続して構成され、該交流側を直流電源の出力に直列接続して上記各単相インバータの出力の総和を上記直流電源の出力に重畳するインバータ回路と、
該インバータ回路の後段に導通/非導通が切り替わる素子を介して接続され、該インバータ回路からの出力を平滑する平滑コンデンサと、
上記インバータ回路に一端が接続され、他端が上記平滑コンデンサの負極に接続された短絡用スイッチとを備え、
上記インバータ回路における直流電力の充放電を利用して直流/直流変換を行うことを特徴とする電力変換装置。 The AC side of one or more single-phase inverters each having a semiconductor switch element and a DC voltage source are connected in series, and the AC side is connected in series to the output of the DC power supply to output the output of each single-phase inverter. An inverter circuit that superimposes the sum on the output of the DC power supply,
A smoothing capacitor connected to a subsequent stage of the inverter circuit via an element for switching between conduction / non-conduction and smoothing the output from the inverter circuit;
A short-circuit switch having one end connected to the inverter circuit and the other end connected to the negative electrode of the smoothing capacitor;
A power conversion apparatus that performs DC / DC conversion by using charge / discharge of DC power in the inverter circuit. - 上記導通/非導通が切り替わる素子は整流ダイオードであることを特徴とする請求項1に記載の電力変換装置。 The power conversion device according to claim 1, wherein the element for switching between conduction and non-conduction is a rectifier diode.
- 上記各単相インバータは、上記半導体スイッチ素子にダイオードを直列接続したブリッジ回路と上記直流電圧源とで構成されることを特徴とする請求項1または2に記載の電力変換装置。 The power converter according to claim 1 or 2, wherein each single-phase inverter includes a bridge circuit in which a diode is connected in series to the semiconductor switch element and the DC voltage source.
- 上記短絡用スイッチの一端は、上記インバータ回路を構成する1以上の上記単相インバータの内、最後段に接続された単相インバータにおける上記直流電圧源の負極に接続されることを特徴とする請求項1または2に記載の電力変換装置。 One end of the short-circuit switch is connected to a negative electrode of the DC voltage source in a single-phase inverter connected to the last stage among the one or more single-phase inverters constituting the inverter circuit. Item 3. The power conversion device according to Item 1 or 2.
- 上記各単相インバータは、上記最後段に接続された単相インバータのみハーフブリッジインバータで構成されることを特徴とする請求項4に記載の電力変換装置。 The power converter according to claim 4, wherein each single-phase inverter includes only a single-phase inverter connected to the last stage as a half-bridge inverter.
- 上記直流電源の電圧を検出する手段を備え、上記インバータ回路は、上記直流電源の電圧に応じて出力制御されることを特徴とする請求項1または2に記載の電力変換装置。 3. The power converter according to claim 1, further comprising means for detecting a voltage of the DC power supply, wherein the inverter circuit is output-controlled according to the voltage of the DC power supply.
- 上記各単相インバータの上記直流電圧源の電圧を検出する手段を備え、該各直流電圧源の電圧が全て、それぞれ設定された電圧値を超えるまで上記短絡用スイッチをオン状態とし、その後上記短絡用スイッチをオフ状態とすることを特徴とする請求項6に記載の電力変換装置。 Means for detecting the voltage of the DC voltage source of each single-phase inverter, and turning on the short-circuit switch until the voltages of the DC voltage sources all exceed the set voltage values, and then the short circuit The power switch according to claim 6, wherein the power switch is turned off.
- 上記短絡用スイッチのオン/オフ切り替え時に、上記インバータ回路は、直流電力の充電/放電を切り替えるように制御されることを特徴とする請求項6に記載の電力変換装置。 The power converter according to claim 6, wherein the inverter circuit is controlled to switch between charging / discharging of DC power when the shorting switch is switched on / off.
- 上記各単相インバータの直流電圧は、上記平滑コンデンサの設定電圧以下とすることを特徴とする請求項1または2に記載の電力変換装置。 3. The power converter according to claim 1 or 2, wherein the DC voltage of each single-phase inverter is set to be equal to or lower than a set voltage of the smoothing capacitor.
- 上記直流電源は、太陽電池であることを特徴とする請求項1または2に記載の電力変換装置。 The power converter according to claim 1 or 2, wherein the DC power source is a solar battery.
- 上記各単相インバータの上記直流電圧源は、コンデンサにより構成されることを特徴とする請求項1または2に記載の電力変換装置。 The power converter according to claim 1 or 2, wherein the DC voltage source of each single-phase inverter is constituted by a capacitor.
- 上記インバータ回路は上記単相インバータを複数個有し、該各単相インバータの出力制御および上記短絡用スイッチのオンオフ制御の組み合わせから成る複数の制御モードを、上記直流電源の電圧に応じて切り替えて用いることにより、上記インバータ回路における直流電力の充放電を利用して直流/直流変換を行うことを特徴とする請求項1に記載の電力変換装置。 The inverter circuit includes a plurality of the single-phase inverters, and switches a plurality of control modes including a combination of output control of each single-phase inverter and on / off control of the short-circuit switch according to the voltage of the DC power supply. The power conversion device according to claim 1, wherein the power conversion device performs DC / DC conversion by using charge / discharge of DC power in the inverter circuit.
- 上記複数の制御モードは、各制御モード毎に直流/直流変換の昇圧比を有し、
上記直流電源の電圧に応じて上記制御モードを決定して切り替えることにより、上記昇圧比を選択することを特徴とする請求項12に記載の電力変換装置。 The plurality of control modes have a DC / DC conversion step-up ratio for each control mode,
The power conversion apparatus according to claim 12, wherein the step-up ratio is selected by determining and switching the control mode according to a voltage of the DC power supply. - 上記制御モードの切り替えは、上記直流電源の電圧変動による上記電力変換装置の出力電圧の変動を小さくするように上記制御モードを選択して切り替えることを特徴とする請求項13に記載の電力変換装置。 14. The power converter according to claim 13, wherein the control mode is switched by selecting the control mode so as to reduce fluctuations in the output voltage of the power converter due to voltage fluctuations of the DC power supply. .
- 上記各単相インバータの上記各直流電圧源の電圧比を可変とし、上記各制御モードに応じて決まる上記電圧比が複数種となるように、上記複数の制御モードが設定されることを特徴とする請求項13に記載の電力変換装置。 The voltage ratio of each DC voltage source of each single-phase inverter is variable, and the plurality of control modes are set such that the voltage ratio determined according to each control mode is plural. The power conversion device according to claim 13.
- 上記複数(N個)の直流電圧源の電圧値Vc(k)は、電圧比が2の累乗(1:2:4・・:2N-1)であることを特徴とする請求項12~14のいずれか1項に記載の電力変換装置。 15. The voltage value Vc (k) of the plurality (N) of DC voltage sources has a voltage ratio that is a power of 2 (1: 2: 4...: 2N-1). The power converter device according to any one of the above.
- 上記複数(N個)の直流電圧源の電圧値Vc(k)のうち、最大となる電圧Vc(N)と最小となる電圧Vc(1)が次の関係式、
Vc(N)≦(Σk=1~N-1Vc(k))+Vc(1)
を満たすことを特徴とする請求項12~15のいずれか1項に記載の電力変換装置。 Among the voltage values Vc (k) of the plurality of (N) DC voltage sources, the maximum voltage Vc (N) and the minimum voltage Vc (1) are expressed by the following relational expression:
Vc (N) ≦ (Σk = 1 to N−1Vc (k)) + Vc (1)
The power conversion device according to any one of claims 12 to 15, wherein: - 上記複数の単相インバータのうち、2以上の単相インバータの上記直流電圧源の電圧が同じであることを特徴とする請求項12~15のいずれか1項に記載の電力変換装置。 The power converter according to any one of claims 12 to 15, wherein among the plurality of single-phase inverters, the voltages of the DC voltage sources of two or more single-phase inverters are the same.
- 上記各単相インバータの出力による上記各直流電圧源の電力授受がバランスするように、上記複数の制御モードが設定されることを特徴とする請求項12~15のいずれか1項に記載の電力変換装置。 The power according to any one of claims 12 to 15, wherein the plurality of control modes are set so that power exchange of each DC voltage source by an output of each single-phase inverter is balanced. Conversion device.
- 上記制御モードを用いた制御において、上記短絡用スイッチの制御を変化させずに、上記各単相インバータの正負の電圧出力のデューティ比を調整することにより、該各単相インバータの上記各直流電圧源の電圧比を調整することを特徴とする請求項19に記載の電力変換装置。 In the control using the control mode, the DC voltage of each single-phase inverter is adjusted by adjusting the duty ratio of the positive / negative voltage output of each single-phase inverter without changing the control of the short-circuit switch. The power converter according to claim 19, wherein the voltage ratio of the source is adjusted.
- 上記複数の制御モードの切り替えは、予め設定された電圧判定値と上記直流電源の電圧とを比較して行い、上記電圧判定値にヒステリシス幅を設けたことを特徴とする請求項12~15のいずれか1項に記載の電力変換装置。 16. The switching between the plurality of control modes is performed by comparing a preset voltage determination value with a voltage of the DC power supply, and a hysteresis width is provided in the voltage determination value. The power converter of any one of Claims.
- 上記短絡用スイッチの一端は、上記インバータ回路の後段の交流側出力線に接続されることを特徴とする請求項12~15のいずれか1項に記載の電力変換装置。 The power conversion device according to any one of claims 12 to 15, wherein one end of the shorting switch is connected to an AC output line downstream of the inverter circuit.
- 上記複数の単相インバータの出力制御および上記短絡スイッチのオンオフ制御の組み合わせから成る上記各制御モードによる該電力変換装置の制御動作は、所定の制御動作が一定の周期で繰り返されるものであることを特徴とする請求項12~15のいずれか1項に記載の電力変換装置。 The control operation of the power converter according to each control mode including a combination of the output control of the plurality of single-phase inverters and the on / off control of the short-circuit switch is such that a predetermined control operation is repeated at a constant cycle. The power conversion device according to any one of claims 12 to 15, characterized in that:
- 上記所定の制御動作による1周期内で、上記各単相インバータの出力による上記各直流電圧源の電力授受がバランスすることを特徴とする請求項23に記載の電力変換装置。 24. The power conversion device according to claim 23, wherein power exchange of each DC voltage source by the output of each single-phase inverter balances within one cycle by the predetermined control operation.
- 上記所定の制御動作による1周期は、少なくとも2種の異なる制御動作期間によって構成されていることを特徴とする請求項23に記載の電力変換装置。 24. The power conversion device according to claim 23, wherein one cycle of the predetermined control operation is composed of at least two different control operation periods.
- 上記各単相インバータが上記1周期内を通して電圧を発生させない場合を除いて、該1周期内に、上記各単相インバータが正電圧を出力させる制御動作期間と負電圧を出力させる制御動作期間とが含まれることを特徴とする請求項25に記載の電力変換装置。 A control operation period in which each single-phase inverter outputs a positive voltage and a control operation period in which a negative voltage is output within the one cycle, except for the case where each single-phase inverter does not generate a voltage through the one cycle. The power conversion device according to claim 25, wherein:
- 上記1周期内で、上記各単相インバータが正電圧を出力させる制御動作期間の総和と負電圧を出力させる制御動作期間の総和とは期間の長さが等しいことを特徴とする請求項26に記載の電力変換装置。 27. The total length of control operation periods in which each single-phase inverter outputs a positive voltage and the total of control operation periods in which a negative voltage is output within the one cycle are equal to each other in length. The power converter described.
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US13/055,153 US8754549B2 (en) | 2008-07-24 | 2009-07-23 | Power conversion device |
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Also Published As
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CN103038990A (en) | 2013-04-10 |
CN103038990B (en) | 2016-06-22 |
DE112009001793T5 (en) | 2011-07-21 |
US20110121661A1 (en) | 2011-05-26 |
JP5028525B2 (en) | 2012-09-19 |
US8754549B2 (en) | 2014-06-17 |
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JPWO2010010710A1 (en) | 2012-01-05 |
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