JPS6098830A - Reactive power compensating device - Google Patents

Abstract

1. Reactive-power compensator for compensating a reactive-current component in an alternating-voltage system, comprising a) a current converter bridge circuit (1 - 4 ; 27, 28) having an alternating-voltage input (19) which is to be connected to alternating-voltage system (18) and a direct-current output (20), b) forced-commutation current converter rectifiers (1 - 4) in all branches of the current converter bridge circuit, which rectifiers are arranged in such a manner that a defined current direction is produced at the direct-current output (10), c) a capacitor (9) which is connected in parallel with the alternating-voltage input (19), and d) a smoothing choke (10) as termination at the direct-current output (20), e) a drive unit (11) for driving the forced-commutation current converter rectifiers (1 - 4) and f) a pulse-width modulator (12) which is connected to this unit and which g) outputs, with a put se frequency (f) which is greater than the frequency of the alternating-voltage system (18), h) width-modulated, approximately rectangular current pulses (I, I1 , I2 ) for controlling the commutation process as determined by a control voltage (Us ) dependent on the magnitude and phase of the reactive-current component (IB ) to the drive unit (11) in such a manner that the fundamental oscillation of the compensation current (I) is identical in frequency and amplitude with respect to the reactive-current component (IB ) but shifted in phase by 180 degrees, characterized in that i) for the purpose of generating a sinusoidal control voltage (US ), a first multiplier (13) is provided which multiplies a reference voltage (UR ), which is phase-shifted by an angle of 90 degrees with respect to the system voltage (UN ) of the alternating-voltage sytem (18), or amplified reference voltage (a. UR ) by a multiplication factor (M), the magnitude of which is proportional to the magnitude of the reactive-current component (IB ) and which changes its sign when the reactive-current component (IB ) changes between the inductive and the capacitive range.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention provides AC voltage %: AC reactive current component at the source;
(a) An inverter bridge circuit having an AC voltage input side and a DC output side to be connected to an AC voltage power supply; (b) all bridge branches of the inverter bridge circuit (C) A forced commutated inverter valve is arranged so that a fixed current direction can be obtained on the DC output side, and (C) a parallel-connected capacitor (d) is installed on the AC voltage input side. )v■δIC output side is provided with a smooth choke as a termination part 1;51''1-.

1 This J. U na invalid 'rp; fluent it+i {:↓'7
The device is a standard from known technology. (L. Abraham and M. Hiuolor: Reactive Current Compensation via Semiconductor Switches or Inverters, VDID Expert Conference, Hannover 1969, pp. 100-114) (L. Abraham. M. Hiuolor:
Blindstromkompene-at±on
iider Halbleit;or8c+lalte
r oderUm r l Qh ter lVDK
1”8Ch tagu njj=”10k jr O
nik.

Hannover. 1969,”,1no-
1 1 4).

As the use of % air machinery and equipment with large reactive f and current consumption increases =+~, the reactive '71: l power demand and demand jFk on the CC secondary power supply load side is increasing.

As a result, on the one hand, the power supply is subject to additional loads, and on the other hand, lightning and pressure fluctuations at the source are difficult to control.

In order to reduce such undesirable effects, or at least to a tolerable degree, there is an increasing trend towards compensating the effects of market currents at the point of origin, i.e. at the load itself. . The reactive component complement 'rj' is the dispersion i'14 of device 6. According to the form, there are many no JAJJ1 deer compensators as follows? In other words, it has a configuration that is advantageous in terms of cost I, in particular, in terms of control time, as much as possible, and in addition, it has a stepless series of generated inductive or capacitive reactive currents j'i'. l: Target++! Reactive power that enables i ifr? Requires Ji tit equipment.

Today, a switched capacitor battery (TSC: thyristor switched capacitor) in combination with a controllable inductance (TCR: thyristor controlled reactance) is used as a reactive power compensator 1. In that case, the maximum inductive charge occurring in the charge load of the capacitor battery I is shifted according to the load. Since capacitor batteries can only be connected and connected in stages, the reactive power range that exists between each stage is taken care of by a controllable inductance. (Federal Republic of Germany Patent Application No. 1962272). In the case of this type of reactive power compensation, it must be expected that the cost of the capacitor batteries to be installed increases with increasing reactive power of the load, so that in the case of high reactive power, the compensation device becomes significantly more expensive.

In contrast, the conference mentioned at the outset proposed the use of reactive current inverters, which essentially consist of an inverter bridge circuit with forcedly commutated inverter valves, for reactive power compensation. The DC output side of the bridge is terminated by a smoothing coil. A capacitor is connected in parallel on the AC voltage input side.
The inverter valve is protected from overvoltage during forced commutation with an AC voltage power supply with a reactance.

Forced commutation in the above-mentioned reactive current inverter referred to as "Type A" is controlled from the time course as follows:
That is, the control is such that rectangular current blocks of polarity alternating with the frequency of the alternating voltage power supply are produced. When compensating for inductive reactive currents, C is approximately 90' with respect to the power supply voltage.
'', and a delay of approximately 90 degrees occurs when compensating for the capacitive reactive 111 flow.

Although operation of such reactive current inverters is possible without the large and expensive capacitor batteries required in other compensation schemes, it presents different problems. On the other hand, the amplitude of the reactive current sent out, applied, and supplied by the inverter is determined by the phase shift of the reactive current with respect to the power supply voltage.
It can only be adjusted by the magnitude of the difference angle value α that deviates from the ideal value of 0. As a result, in addition to the desired reactive power, alternating heights of active power result. On the other hand, the transition between the inductive and the visual operating regions of the reactive current inverter takes place only discontinuously, ie with a 180° phase jump, resulting in large gaps in the control region. Furthermore, the reactive current delivered or applied by the DC block contains a significant overoscillation component, which is filtered out by expensive input filters to avoid disturbing power supply reactions.
must be suppressed.

Therefore, the objective of the present invention is to achieve inductivity at a small cost? The object of the present invention is to provide a reactive power compensator in the form of a reactive current inverter that can be continuously controlled throughout the +i:"j+'+,' compensation region and at the same time has a clearly reduced over-oscillation component in the reactive current.

According to the invention, this problem is solved by the features set forth in claim 1 in a reactive power compensator of the type mentioned at the outset.

The gist of the invention is that in the reactive current inverter, the control elements for the inverter valves forced to be commutated with 'C' are selected as follows: the reactive current impulse is supplied with 'C' or 'Jk' applied. The reactive current is selected to consist of a series of sinusoidally length-modulated current blocks or current pulses, the pulse frequency being greater than the frequency of the mains alternating voltage. The sine function used for length modulation has the same frequency as the reactive current component to be compensated, but is out of phase by 180 degrees with respect to these components.

The first advantage obtained by the above-mentioned selection and configuration of the control elements for forced commutation inverter valves is that due to the sinusoidal length modulation of the current pulses, a high fundamental oscillation component and only a slight overoscillation component in the compensation current can be obtained. (which decreases with increasing pulse frequency).

Another advantage is that the amplitude of the fundamental wave can be varied continuously by changing the pulse length without changing the phase of the fundamental wave with respect to the power supply voltage or the reactive current component of the power supply.

As a sixth advantage, this results in a continuous transition between the inductive and sutureable compensation areas.

For example, if the reactive current component to be compensated is
When switching to the neutral region, in the reactive current inverter,
First, the capacitive compensation current is controlled to be reduced by decreasing the pulse width.

/''\ ('t:j 'The widening of the Urn pulse causes the weir to be affected to the following degree, i.e., to the extent that the now capacitive reactive current component of the power supply is compensated for.Pulse widening and phase Such a combination or combination with switching creates a substantially gap-free control region of the flow compensator according to the invention.

Next, the present invention will be explained using the illustrated embodiments.

The main application area of the reactive current compensator of the present invention when compensating reactive currents is a three-phase power supply, and therefore a three-phase inverter bridge is used inside the compensator, but the following description is related to one embodiment. For the sake of clarity, C is made for the simplest case of a single-phase AC voltage power supply. The implementation of the invention can be easily adapted by a person skilled in the art in the case of a three-phase power supply.

FIG. 1 shows a block diagram of an advantageous embodiment of the reactive current compensator according to the invention. A load 5 is connected to an alternating current voltage power supply 18 having a sinusoidal power supply voltage UN, through which a load current, also sinusoidal, flows. At this time, it is assumed that the load current is not in phase with the power supply voltage UN.

A current measuring device 6 is connected in the lead line to the load 5 for supplying the powder and UN, and a voltage measuring device 7 is provided in parallel with the load. The measuring device is of the type normally used for monitoring the supply fL source, and is a measuring transducer known per se and familiar to those skilled in the art.

Similarly, in parallel with the load 5, the inverter valve 1 is forcedly commutated.
．． . . , 4, an inverter bridge circuit having an AC voltage input side 19 and a DC current output side 20 are connected. Forced current inverter valve 1. ..., 4 are arranged in the individual bridge branches in such a way that a fixed current direction is obtained on the DC output side 20 for the flow 1b flowing therethrough. . Therefore, the glaze inverter bridge circuit is also called an inverter.

The AC voltage input side 19 is connected to the AC voltage power supply 18 at °C.
while the DC output side 20 is terminated by a smooth choke 10. Additional 1+ in parallel to the AC voltage input side 19
:, 'j”ii +1j11 1MTh O”)
ICs', (') Ia y'-Yuben 1,...
4 +/,) 1i'l! A capacitor 9 is provided for (.

Impark valve with forced commutation 1. ..., 4 is an independent element, and C1 or 1.1 is connected to the bridge by an appropriate additional circuit connection (commutating circuit) so that C1 is not approached or blocked regardless of the input pressure that governs each time. Particularly advantageous in this case is the use of a cut-off thyristor (()To : /r"-1' turn-off, since its thyristor is negligible since no additional circuit connection is required. This is because the simplified configuration of the current compensator is positive.

Inverter valve with forced commutation1. . . , 4 is connected to the book fitil unit 11 through its control input side, and this control unit 1i1i1 generates an appropriate pulse for shutting off the valve δ°v1 according to a predetermined time sequence or time relationship. There is one sending out. The above-mentioned predetermined sequence or time relationship is determined by the sequence of logic pulses, and this logic pulse train is connected to the pulse length modulator 12 on the pulse input side AXB of the control unit 11, and the combination of U8〉0 is the output thereof. Tsukuda■ Sends an equal signal corresponding to A-[1 to logging 1'', and in the case of U8 to O, sends a signal corresponding to logic "0".

The L pressure TJs is at the same time the modulation voltage for the pulse length modulator 12, the pulse frequency f of which is equal to the pulse period I skin (set by the pulse generator 17). 0 As the pulse length modulator 12, a circuit known in the field of pulse length modulation technology can be used, and in this known circuit, a sawtooth frequency signal of pulse frequency J3f generated by the pulse frequency generator 1γ is applied to the modulating voltage signal. The sum signal 'tl' is fed to a one-image dazzle detector. At the output of this detector, a pulse modulated pulse with l'f <'j' set to an appropriate value is supplied. appear.

The control Li LEUS is the product of a multiplication process, and in this process, in the first multiplier 13, the amplified base bypass voltage a−U
R is multiplied by a multiplication factor M. That Vx that visiting clerk is a load 1 (i that takes a positive or negative value according to the phase state of machining.
In this other multiplication and θ step, the second multiplier 1G is inputted to the second multiplier 1G. The product is multiplied, and the product is processed in the low-pass filter 1/ of the follower connection, and the direct current filt EF hv is extracted.

(increased) base voltage a UR and inverted base j bald voltage -b-ITRG, .: one common base voltage U,
The signal is derived from the amplifier 021 through the inverting amplifier 15. The above 諫7φ'i(? IJ' T, IR is F(
9 from the output signal of the i pressure measuring device 7 at the phase shifter 8.
0 Caused by phase shift. With amplifiers 15 and 21, the above standard 1! j' pressure, -b-1, and -b-1 can be adjusted or set separately to six values suitable for further processing of the signal. However, the amplifier 21 may be omitted.

The reactive power supplement A11, B'; I8 operation is clarified from the pressure waveform and time course. In the above, it is assumed that the dielectric reactive force is taken out from the AC voltage galvanic corrosion 18 by the load, that is, the load current ■ has a phase angle of 0 with respect to the power supply voltage UN, and lags ψ by 90°. The current and voltage vectors are shown in the vector diagram of Fig. 2.The vector of the loaded electric machining is rotated clockwise by a phase angle ψ with respect to the vector of the power supply voltage UN.The vector of the loaded electric machining is , active current component M, and reactive current component Ig
(This should be compensated for by a reactive power compensator).

This example of delayed load electric machining is reactive current component machining.

runs parallel to the vector of the sinusoidal reference voltage UR (which is delayed by 90 degrees with respect to the voltage UN by the phase shifter 8). On the other hand, in the case of a leading load current,
That is, in the case of capacitive reactive power, the vector of the reactive 'i flow component' runs parallel to the inverted reference, 1-b-UR.

The time course of voltage and current is shown in FIGS. 3a and 3b. Figure 3a shows the relationship between the power supply voltage UN)L and 'V gap t, the load electrical processing delayed by ψ, and the reference voltage UR delayed by 90°.
Show the button. On the same scale, FIG. 6b again shows the load application together with its reactive current component and the control voltage U8 required for the pulse length modulator 12. The control tfl pressure U8 is always 180° out of phase with respect to the reactive current component,
Therefore, it is in phase with the compensation current Δ(which is to be compensated by the reactive power compensator), and the reactive current component is exactly compensated.

The reference voltage UR and the amplified reference voltage a-UR have a constant amplitude 1194 over time and are delayed by 90 degrees in phase with the power supply voltage UN, and their amplitude and phase are generally larger than the reactive current component. Since the control voltage US is switched between inductive (+90) and inductive (-90), a corresponding follow-up control of the control voltage US takes place as follows: the amplified reference voltage a-
UR is multiplied by a multiplication factor M, which is variable in magnitude and polarity. In the case of inductive reactive current component IB, the control voltage U due to the negative polarity of M
A required phase shift of 180° relative to a-URl of 8 is produced.

In the case of capacitive reactive current components, the negative polarity of M does not change the phase, since the required phase shift already exists between rice and a-UR.

The magnitude of M affects the amplitude of U8 and, via the pulse length modulator 12, also affects the amplitude of the composite compensation current Δ. Its magnitude therefore depends on the amplitude of the ineffective 'turbine flow component' 8 and is controlled by the 'C' order, ie, .DELTA. For example, form (1) where the power supply voltage UH depends on time
When UN=UN8in ω, where UN is the amplitude and ω is the angular frequency, the following equation holds true for URL and 8 as shown in FIG.

(2) UR = URslH("t90) = -UR
CoBωt(3)■=engine'5in(ωt±ψ) Known as the voltage measuring device 7 in the second multiplier 1G.

By multiplying the respective associated signal quantities from the flow measuring device 6, the following equation is obtained.

(4) -'b-UR・Work=bUW Work0Sin(ωt4
4+) coe ωt After the colossal filter 14 suppresses the component that oscillates at the angular frequency ω, the following DC signal component remains for the multiplication coefficient M.

(5,7bIU$f 0.5in(-+9')" +
2 bUR work -e1nψHowever, the amplitude rice - invalid 11te flow component work 8's work 0.1llin 9+.

Of (5), the multiplication coefficient M proportional to the DC component has precisely the desired characteristics. This multiplication factor is negative for the inductive reactive current component (−ψ) and is negative for the capacitive reactive current component (−ψ).
-ψ), it is a pressure and is proportional to the amplitude factor d of the reactive current component factor 3. In that case, the number of ratio measurements can be adjusted appropriately by the transfer coefficients of the current measuring device 6 and the inverting amplifier 15.

According to the control voltage U8 determined by M and a-UR, the inverter bridge circuit generates a compensated 1F current Δ in the form of sinusoidally length-modulated square wave pulses as shown in diagram m30.
cause en. Forced commutation inverter 1. ...
, 4 is shown in FIG. 6d. In this case, the hatched areas indicate the period during which the respective inverter valve is electrically connected. At time t2, for example, the inverter valve 1.4 is simultaneously switched on, so that from that point on, according to the design of FIG. flowing. At time t'2, inverter valve 1 is switched off and inverter valve 2 is switched on instead, while inverter valve 4 remains in the conducting state. The compensation current Δ is zero until the inverter valve 1 is connected again. The conductively connected inverter valves 2 and 4 form a freewheeling branch to a smooth choke 10 in that phase of the compensation current Δ, which causes the freewheeling branch to have an unchanging magnitude on the basis of the energy obtained by said choke. DC current is driven. Such alternating cycles of Δ epulses and freewheel times are repeated with pulse frequency f and with on-off ratios alternating within 13-12 in time intervals in which the compensation current Δ epulses are positive. The agent of the inverter valve 1.2 is replaced in the adjacent interval 12-11 with a negative complement IrL current Δ current. The negative current pulse enters the conduction time of the inverter 2, while the inverter valve 1.3 together forms a freewheeling branch.

6a to 6d show the control sequence for the rust-conducting reactive current component. For capacitive reactive current component IB, U9
, I are 180° out of phase.

The Δ knee pulse of the pressure then falls within the time interval 12-11, so that the control times of the inverter valves are correspondingly interchanged.

Strong 7B according to Figure 3d! 1 commutated inverter valve 1
,..., 4, appropriate logic pulses from the pulse length modulator 12 and the frame detector 22 are required at the control unit's pulse input 41 (11A, B). The pulses are shown in the two upper rows of FIG. 4. The logic pulse train delivered by the polarity detector 22 causes the control voltage U to rise at times t1, t2, and t3.
Each time the polarity of S is switched, it switches from logic "1" to logic "0" and vice versa. Pulse length modulator 12
The train emitted from the circuit consists of positive sinusoidal length modulated logic pulses. This can be done, for example, in the following way: the control voltage US is first applied to the pulse length modulator 12 in °C to a full-wave rectifier and only subsequently used for the actual pulse length modulation. The goal is to make sure that you are able to do so.

The combination of logic pulses on the pulse inputs A, B takes place in the control unit 11 according to the truth table of FIG. For example, both pulse input sides A and B are logic “0”
'', the logic ``1'' is delivered to the output associated with the inverter valve 1.3, while the output for the inverter valve 2.3 becomes the logic ``0''. Four different output pulse trains are thus produced from the A and B input pulses, which are shown in the bottom four columns of FIG. Each of the above columns is associated with one inverter valve. One of those output pulse trains in FIG.
The respective time intervals in the two (in which time intervals the signal is placed at logic "'1") correspond to the conductive connection time intervals shown hatched in FIG. 3d.

A suitable circuit configuration of the control unit 11 that allows the necessary logic combinations is shown in FIG. The logic pulses on the pulse input side A and B are
; the output of this grate is connected on the one hand directly via a pulse converter 23 to the control input of the inverter valve 2 and on the other hand to the first inverter 2.
4 and the control input side of the inverter valve 1 via the same pulse converter 23.

From the pulse input side A, two further lines reach the C inverter valves 3 and 4 via a second inverter 25 and a pulse converter 23 (or directly via a pulse converter).

The pulse converter 23 is constructed in the same way for the output side of all C and forms the appropriate control pulses for the forcedly commutated inverter valves 1° . . . , 4 from the logic pulses, for example by means of differentiation. .

The control pulse has the following characteristics. That is, it is characterized in such a way that during a transition from logic "0" to "1" the associated inverter valve is electrically connected, while during a jump from logic "1" to "0"' the inverter valve is switched off. ing.

The technical configuration of the pulse converter 230 circuit depends on the firing and shutting characteristics of the inverter valve used and is known to those skilled in the art.

Overall, the reactive current compensator obtained according to the invention allows a rapid, continuous and flexible control of the reactive current components occurring in an alternating voltage power supply with little constructional outlay.

In this case, the sizes of the smoothing coil 10 and the capacitor 9 can be selected to be smaller as the pulse frequency f becomes higher. Therefore, the pulse frequency is determined by the power supply voltage UN
It is advantageous to adjust the frequency above 10 KHz, in particular above 10 KHz.

In this way, for example in the case of cut-off thyristors, with the possible switching (circuit) frequencies, the costs in the passive components are kept low.

A further reduction of the over-oscillation component in the compensation current circuit can be achieved as follows: As shown in FIG. Let it be controlled. In this case, AC Tb and piezoelectric power source are used as three-phase power source and C
This is an indication. The inverter valves of each inverter bridge circuit 27 and 28 are connected to 6
A phase bridge is provided in the entire circuit.

For redundancy reasons, the lateral inverter bridge circuits 27, 28 are then constructed in the same manner as in the control section of FIG. It should be taken into account that instead of the single-phase supply voltage of FIG. 1, the three line voltages of the six-phase supply are now viewed in their own right and are used for control purposes.

Each inverter bridge circuit 27 , 28 supplies an associated compensation current Δkl to Δk to the alternating voltage power supply 18 .

At that time, both compensation currents ΔΔ1. By superimposing the Δfactors 2, the 'C composite compensation effect: the flow Δfunctions is obtained.

Corresponding to FIG. 1, the C double inverter bridge circuit 27.
One pulse frequency generator 17 , 17 ′ is assigned to each of 28 . Both pulse frequency generators 17, 17'
preferably generate the same pulse frequency f. All other voltages used for control in both inverter bridge circuits 27, 28 are in phase, but between the pulse trains of the pulse frequency generator 17, 17' there is a deviation angle β associated with the pulse frequency f. A phase difference of the form occurs (Figure 8).

Is this achieved by &:L synchronization line 29 and both pulse frequencies are generated by this line? ( ) 17 and 17' are not connected and are synchronized taking into account the angle β.

Compensation of both inverter bridge circuits 27.28 '4 style JT
-z, 7! Each pulse train of II2 is similarly shifted by a shift angle β, for example, 66°, as shown in FIG. The deviation angle β is advantageously selected such that a certain, particularly strong λ, overoscillation frequency in the composite compensation current is highly attenuated due to mutual cancellation (in the case of β - 36°) (over-vibration at 5 times the fundamental frequency).

In the same way, by adding another inverter bridge circuit controlled by the shift 'C clock and appropriately setting different shift angles, multiple over-vibrations of different frequencies can be prevented from affecting each other to a certain degree. can give. In that case, the deviation angle is selected by Fourier analysis of the compensation current pulse.
It can be easily carried out by one skilled in the art.

Effects of the Invention The first advantage obtained by the above-mentioned selection of the control element for the forced commutated impark valve and the 47g configuration is that, based on the sinusoidal length modulation of the current pulse, there is a high fundamental vibration component in the compensation current and a high fundamental vibration component in the compensation current. A further advantage is the presence of a slight overoscillation component (which decreases with increasing pulse frequency) at The amplitude of the fundamental wave can be varied continuously by changing the pulse length without changing the pulse length.

As a third advantage, this results in a continuous transition between the inductive and capacitive compensation regions.

For example, when the reactive current component to be compensated switches from the inductive to the capacitive region, in the reactive current inverter, the capacitive compensation current is first controlled to be reduced by a seven-wave width reduction.

[Brief explanation of the drawing]

FIG. 1 is a block connection diagram of one embodiment of the reactive current compensator of the present invention, FIG. 2 is a vector diagram showing the relative positional relationship of current and voltage generated in the circuit of FIG. 1, FIG. 3
Figure b is a waveform diagram showing the time course of the current and voltage in Figure 2;
FIG. 3C is a waveform diagram showing the time course of the length-modulated compensation (ffIL flow); FIG. The figures are waveform diagrams showing the time course of the logic pulses in the control unit of Figure 1 with respect to the compensation current of Figure 3C, Figure 5 is a diagram showing the truth table related to the logic combination of the logic pulses of Figure 4, and Figure 6 The figure is a connection diagram showing an embodiment of the control unit in Figure 1, Figure 7 is a layout configuration diagram of two inverter bridges that perform staggered clock control in a three-phase AC power supply, and Figure 8 is a diagram showing the deviation of the control unit in Figure 7. It is a waveform diagram showing the pulse phase shift of the compensation current during clock control. 5... Load, 6... Current measuring device, 7... Voltage measuring device, 8... Phase shifter, 11 ...control unit, 1
2... Pulse length modulator, 13... First multiplier, 16
...Second multiplier, 17...Pulse frequency generator, 2
2...Polarity detector.

Claims (1)

[Scope of Claims] 1. A force compensator for compensating for reactive current components in an AC power supply, comprising (a) an AC voltage input side (19) to be connected to an AC hook pressure power supply (18); ) and a DC output side (20))-11, (b) inverter valves (1...4) with forced commutation in all bridge branches of the inverter bridge circuit;
), the inverter valve has a DC output (IItl(20)
(C) A capacitor (9) connected in parallel on the AC voltage input side (19) (d) 1μ current output side (20) (Q) The inverter bridge circuit has a pulse frequency fil! on its AC voltage input side (19) which is greater than the frequency V of the AC voltage source (18). (f) receiving a compensating current (Δ) in the form of an approximately rectangular current pulse; (f) said current pulse being modulated according to a sinusoid in its polarity and length; The length change is caused by the compensation current (Δ
This reactive power compensator is characterized in that the fundamental vibration of the motor is the same in terms of frequency and amplitude as compared to the reactive current component, but is 180° out of phase with respect to the reactive current component. . 2. The compensating device according to claim 1, wherein the forcedly commutated inverter valves (1° . . . , 4) have a local risk. B. Forced commutation inverter valves (1,..., 4)
The means for controlling fljli m1'+unit (
11) and a pulse length modulator 5 (12) connected thereto, which modulator uses a pulse frequency (f) to generate a control voltage (UB
11. A compensating device according to claim 1, wherein the compensating device is adapted to send a signal to a jj control unit) (11) for controlling a pulsed commutation process which is pulsed by K and C length modulated. 4. The first multiplier (13
), and the multiplier generates a reference voltage (UR) or an amplified reference voltage (a-UR) that is phase-shifted by an angle of 90 degrees with respect to the power supply voltage (UN') of the AC voltage power supply (18).
The magnitude of the multiplication coefficient is proportional to the magnitude of the reactive current component (XB'), and the multiplication coefficient is such that the reactive current component (IB) is inductive and capacitive. 7. The compensating device according to claim 6, wherein the sign changes when switching between the two regions. 5. Reference (('i pressure (UR) is 2!s output from the power supply voltage (UN) by the voltmeter measuring device (7) and the post-connected phase shift clock (8), and the power supply voltage (UN) A compensating device according to claim 4, wherein the compensation device has a delay of X. 6. A second multiplier (16) for forming a multiplication coefficient (M).
& design, one input side of the second multiplier is an inverting amplifier (1
5) Connected to the output 111+ of the phase shifter (8), the other input side is connected to the current measuring device (6), which is connected to the alternating current voltage signal (18).
Measure the load current (1) flowing through the load (5) and send it to the device (16), which is an alternating voltage signal corresponding to the load current (1) in terms of magnitude and phase. The compensation device according to claim 5 Rc. 2. The compensator according to claim 1, wherein the pulse frequency (f) is much higher than the frequency of the power supply voltage (UN), which is higher than 10 KH2. 8. (a) Same type of impark bridge circuit (27゜2
(b) The inverter bridge circuits (27, 28) are connected to the AC voltage power supply (18) in the front row, (C) Each inverter bridge (27, 28) has a compensation current (Δ ．．
2) reception, in which the current pulses of the compensation current have the same frequency (f), but appear offset from each other by a deviation angle (β); Supplementary equipment as described. 9. A claim that provides a means for determining the deviation angle (β), in which the over-oscillation component takes a minimum value when the compensation tlj flow (Δkl, Δk,) is superimposed