JP2021190825A - Noise filter and power conversion device - Google Patents

Abstract

To provide a noise filter capable of attenuating a transition component from a common mode component into a differential mode component, and to provide a power conversion device comprising the noise filter.SOLUTION: In a power conversion device, a noise filter 18B has: a common mode choke coil Lc, a plurality of input capacitors Cx1-1 to CX1-3 star-connected to an input side of the common mode choke coil; a plurality of output capacitors CX2-1 to CX2-3 star-connected to an output side of the common mode choke coil; and a ground capacitor CY connected between a neutral point of the plurality of output capacitors and the ground. The power conversion device has the noise filter, and an inverter connected with the noise filter.SELECTED DRAWING: Figure 3

Description

The present disclosure relates to noise filters and power converters.

Power conversion devices such as inverters and pulse width modulation (PWM) rectifiers use semiconductor elements such as MOSFETs (Metal Oxide Semiconductor Field Effect Transistor) and IGBTs (Insulated Gate Bipolar Transistor) as switching elements. It is composed. Power converters are being applied in various fields due to the merit of being able to convert electricity into a desired form.

However, electromagnetic noise (conduction noise or radiation noise) generated by a switching operation in which the switching element is repeatedly turned on and off at a high speed of several kHz to several 100 kHz may propagate from the power conversion device main body to space and spread. The electromagnetic noise may propagate and spread through a cable connecting the power conversion device and the grid or a cable connecting the power conversion device and the load. The electromagnetic noise spread in this way may cause problems such as damage and malfunction of peripheral devices and noise mixing of wireless devices. Therefore, a limit value is set by the EMC (Electro Magnetic Compatibility) standard for the electromagnetic noise generated by an electric / electronic device such as a power converter, and it is required to sufficiently reduce the electromagnetic noise.

As an EMC filter for reducing such electromagnetic noise, a noise filter including a common mode choke coil is known (see, for example, Patent Documents 1 and 2).

Japanese Unexamined Patent Publication No. 7-22886 Japanese Unexamined Patent Publication No. 2014-216997

Conduction noise is roughly classified into a common mode component and a differential mode component according to the propagation path, and it is necessary to appropriately set the noise attenuation performance of the EMC filter in order to satisfy the standard.

However, it may be required not only to reduce the common mode component and the differential mode component, but also to reduce the mode conversion component between the common mode component and the differential mode component.

The present disclosure provides a noise filter capable of attenuating a conversion component from a common mode component to a differential mode component, and a power conversion device including the noise filter.

This disclosure is
Common mode choke coil and
A plurality of input capacitors delta-connected to the input side of the common mode choke coil, and
A plurality of output capacitors star-connected to the output side of the common mode choke coil, and
Provided is a noise filter including a ground capacitor connected between the neutral point of the plurality of output capacitors and the ground. The present disclosure also provides a power conversion device including the noise filter.

According to the present disclosure, the conversion component from the common mode component to the differential mode component can be attenuated.

It is a figure which shows the configuration example of the motor drive system which includes the power conversion device in one Embodiment. It is a figure which shows the structural example of the noise filter in the 1st comparative form. It is a figure which shows the structural example of the noise filter in the 2nd comparative form. It is a figure which shows an example of the measurement result of the conduction noise of the inverter to which the noise filter shown in FIG. 3 is applied. It is a figure which shows an example of the parallel resonance of the noise filter shown in FIG. It is a figure which shows an example of the one-circle path of the parallel resonance circuit of the noise filter shown in FIG. It is a figure which shows the connection state at the time of measuring the inductance component of a parallel resonance circuit. It is a figure which shows an example of the measurement result of the conduction noise of the inverter to which the noise filter shown in FIG. 3 is applied. It is a figure which shows the structural example of the noise filter in one Embodiment. It is a figure which shows an example of the one-circle path of the parallel resonance circuit of the noise filter shown in FIG. It is an example of the simple simulation result of the noise terminal voltage in each case of FIG. 4 (reference), FIG. 8 (moving to 150 kHz or less) and FIG. 9 (system side Δ connection). It is a figure which shows an example of the simulation result which changed the ratio of the capacitance of the line capacitor of the system side and the inverter side in the state which kept the resonance frequency substantially constant from the condition of the reference of FIG. It is an enlarged view of a part of the frequency range of FIG. It is a figure which shows an example of the simulation result at the time of increasing the ratio by increasing each capacitance of the line capacitor on the inverter side without changing the capacitance of the line capacitor on the system side from the condition of the reference of FIG. .. FIG. 14 is an enlarged view of a part of the frequency range of FIG. It is a figure which shows an example of the simulation result at the time of increasing the ratio by reducing each capacitance of the line capacitor of a system side without changing the capacitance of the line capacitor of an inverter side from the reference condition of FIG. FIG. 16 is an enlarged view of a part of the frequency range of FIG. It is a figure which shows an example of the simulation result at the time of increasing the ratio by reducing each capacitance of the line capacitor on the system side based on the condition which fixed the capacitance of all line capacitors to 0.3 [μF]. FIG. 18 is an enlarged view of a part of the frequency range of FIG.

Hereinafter, embodiments according to the present disclosure will be described with reference to the drawings.

FIG. 1 is a diagram showing a configuration example of a motor drive system including a power conversion device according to an embodiment. The motor drive system 100 shown in FIG. 1 is supplied by a power conversion device 300 that converts AC power supplied from an AC power source 90 that supplies three-phase AC power into AC power having different frequencies, and a power conversion device 300. It is equipped with a motor 94 driven by AC power.

The motor drive system 100 includes an input cable 91 for connecting an AC power source 90 such as a commercial power source and a power conversion device 300. Note that FIG. 1 shows a form in which a LISN (Line Impedance Stabilization Network) for measuring conduction noise of a variable speed motor drive system 100 is inserted between an AC power supply 90 and an input cable 91. In normal use, where it is not necessary to measure conduction noise, the LISN is not inserted.

The input cable 91 is supplied from the cable 91r to which the R-phase AC voltage supplied from the AC power supply 90 is supplied, the cable 91s to which the S-phase AC voltage supplied from the AC power supply 90 is supplied, and the AC power supply 90. It has a cable 91t to which an AC voltage of the T phase to be supplied is supplied, and a grounding cable 91a. The cables 91r, 91s, and 91t are connected to the input terminals R, S, and T provided in the power conversion device 300, respectively. The grounding cable 91a is connected to the input grounding terminal E provided in the power conversion device 300. The power conversion device 300 is grounded to a reference ground surface 93 such as the ground or the floor surface via a ground cable 91a.

The power conversion device 300 includes a noise filter 18 connected to the input terminals R, S, T and the input ground terminal E, and an inverter 200 connected to the noise filter 18.

The noise filter 18 is a device for reducing electromagnetic noise (particularly conduction noise), and has a function of reducing conduction noise flowing out from the inverter 200 of the power conversion device 300 to the AC power supply 90. The noise filter 18 may have a function of reducing radiated noise radiated from the inverter 200 of the power conversion device 300 to the external space. The noise filter 18 has a configuration including an LC filter, and a detailed configuration thereof will be described later.

The inverter 200 converts the three-phase alternating current supplied from the alternating current power supply 90 into direct current by diode rectification in the converter section 19 and outputs the alternating current of a desired voltage and frequency generated by the inverter section 20 to output the motor. Drives 94. The inverter 200 has a converter unit 19 connected to the noise filter 18, decoupling capacitors C _dc and C _s connected to the converter unit 19, and an inverter unit 20 connected to the decoupling capacitors C dc and Cs. ing.

The converter unit 19 is a rectifier circuit that converts three-phase alternating current input from intermediate terminals R', S', and T'to direct current via a noise filter 18. The converter unit 19 has six diodes 19a, 19b, 19c, 19d, 19e, 19f to which each of the AC voltages input from the intermediate terminals R', S', and T'is applied via the noise filter 18. is doing.

The diode 19a and the diode 19d are connected in series. The AC voltage of the R phase of the AC power supply 90 is applied to the intermediate point which is the connection point between the diode 19a and the diode 19d via the power supply line of the R phase of the noise filter 18. The cathode of the diode 19a is connected to the positive electrode bus 21 of the inverter 200, and the anode of the diode 19d is connected to the negative electrode bus 22 of the inverter 200.

The diode 19b and the diode 19e are connected in series. The AC voltage of the S phase of the AC power supply 90 is applied to the intermediate point which is the connection point between the diode 19b and the diode 19e via the power supply line of the S phase of the noise filter 18. The cathode of the diode 19b is connected to the positive electrode bus 21 of the inverter 200, and the anode of the diode 19e is connected to the negative electrode bus 22 of the inverter 200.

The diode 19c and the diode 19f are connected in series. The AC voltage of the T phase of the AC power supply 90 is applied to the intermediate point which is the connection point between the diode 19c and the diode 19f via the power supply line of the T phase of the noise filter 18. The cathode of the diode 19c is connected to the positive electrode bus 21 of the inverter 200, and the anode of the diode 19f is connected to the negative electrode bus 22 of the inverter 200.

The decoupling capacitors C _dc and C _s are connected in parallel between the positive electrode bus 21 and the negative electrode bus 22. The decoupling capacitor C _dc has a larger capacitance than the decoupling capacitor C _s. The decoupling capacitor C _dc reduces the ripple of the DC voltage between the positive electrode bus 21 and the negative electrode bus 22. Decoupling capacitor C _s reduces the high frequency noise than the ripple of the DC voltage.

The inverter unit 20 is a circuit that converts a direct current between the positive electrode bus 21 and the negative electrode bus 22 into a three-phase alternating current. The inverter unit 20 has six switching elements S ₁ , S ₂ , S ₃ , S ₄ , S ₅ , and S ₆ . The switching elements S ₁ , S ₂ , S ₃ , S ₄ , S ₅ , and S ₆ are semiconductor elements such as, for example, IGBTs.

The connection point between the switching element S ₁ and the switching element S ₄ is connected to the output terminal U. The connection points between the switching unit S ₂ and the switching unit S ₅ are connected to the output terminal V. The connection points between the switching unit S ₃ and the switching unit S ₆ are connected to the output terminal W.

The motor drive system 100 includes an output cable 92 that connects the power conversion device 300 and the motor 94. The output cable 92 supplies a cable 92u for supplying a U-phase AC voltage to the motor 94, a cable 92v for supplying a V-phase AC voltage to the motor 94, and a W-phase AC voltage to the motor 94. It has a cable 92w for the purpose and a grounding cable 92a. The cables 92u, 92v, and 92w are connected to the output terminals U, V, and W provided in the power conversion device 300, respectively. The grounding cable 92a is connected to the output grounding terminal E'' provided in the power conversion device 300.

The motor 94 is grounded to the grounding portion 12 provided in the power conversion device 300 via the grounding cable 92a and the output grounding terminal E''. The grounding portion 12 may be a heat sink for the inverter 20. The grounding portion 12 is grounded to the reference ground surface 93 via the intermediate grounding terminal E', the ground wire of the noise filter 18, the input grounding terminal E, and the grounding cable 91a. Therefore, the motor 94 is also grounded to the reference ground surface 93 via the power conversion device 300 and the ground cable 91a.

The measuring device (not shown) measures and evaluates the voltage that reaches the LISN via the input cable 91 as conduction noise. Here, a plurality of forms can be considered as the noise filter 18 for satisfying the noise regulation defined by CISPR or the like.

FIG. 2 is a diagram showing a configuration example of a noise filter in the first comparative mode. FIG. 3 is a diagram showing a configuration example of the noise filter in the second comparative mode.

The noise filter 18A shown in FIG. 2 has a _{plurality of line capacitors C X1-1} to C _X1-3 and C _X2-1 to C _X2-3 connected by Δ connection (delta connection) on both sides of _{the common mode choke coil L C. And a plurality of grounding capacitors C Y1} to C _Y3 connected between the power line and the ground line of each phase on the inverter side. On the other hand, the noise filter 18B shown in FIG. 3, the capacitor between a plurality of lines Y-connection (star connection) on both sides of the common choke coil _{L C C X1-1 ~ C X1-3,} C X2-1 ~ C X2- ₃ and a plurality of line-to-line capacitors C _X2-1 to C _X2-3 on the Y-connected side of the inverter are provided with _{one grounding capacitor C Y} connected between the neutral point and the ground wire.

The configuration of FIG. 2 has an advantage that the equivalent line capacitor capacity can be increased as compared with the configuration of FIG. 3 by making a Δ connection of the line capacitors. On the other hand, the configuration of FIG. 3 has an advantage that a capacitor having a lower withstand voltage than that of the configuration of FIG. 2 can be applied by Y-connecting the line-to-line capacitor. Further, since the configuration of FIG. 3 has only one grounded capacitor, there is an advantage that the mounting area can be reduced in consideration of the insulation distance and the like. In view of these characteristics, the configuration of FIG. 2 is preferably applied to equipment of a relatively low voltage system (for example, AC 200V system), and the configuration of FIG. 3 is preferably applied to a higher voltage system (for example, AC 200V system). It is preferable to apply it to AC 400V system) equipment.

The noise filters 18A and 18B shown in FIGS. The inductance and the line-to-line capacitor reduce the conduction noise of the differential mode component flowing in the power line of each phase.

However, the conduction noise reduced by the noise filter is not only the common mode component and the differential mode component, but also the component that converts the mode between the respective components (from the common mode component to the differential mode component, or from the differential mode component to the common mode). Ingredients to ingredients) may also need to be reduced.

Next, the problem of mode conversion from the common mode component to the differential mode component, which occurs when the noise filter 18B shown in FIG. 3 is applied, will be described.

FIG. 4 is a diagram showing an example of a measurement result of conduction noise of an inverter to which the noise filter shown in FIG. 3 is applied. Unnecessary peaks around 250kHz exceed the standard. The peak exceeding this standard is caused by parallel resonance as shown in FIG.

_{Consider the case where the high-frequency leakage current i L} flowing through the ground wire in a non-conducting state of one phase of the three-phase diode rectifier circuit (T phase in FIG. 5) bypasses the ground capacitor C _Y and returns to the inverter side. .. In this case, in the R phase and S phase, the high frequency leakage current i _{L returns} to the conduction phase (R phase and S phase) of the diode rectifier circuit via _{the line capacitors C X2-2} and C X2-3 on the inverter side. Is secured. On the other hand, in the T phase, the high frequency leakage current i _L is passed through the common mode choke coil L _C , the line capacitors C _X1-1 to C _X1-3 on the system side, and the power lines of the R and S phases. It will return to the S-phase diode. At this time, a parallel resonance path (parallel resonance circuit) indicated by the solid arrow in FIG. 5 is formed, and a large current flows at the resonance frequency. A large conduction noise is observed when a part of this resonance current flows out to the system side (AC power supply side). That is, the common mode component flowing through the ground wire is _{converted into the interphase voltage via the line capacitors C X1-1} to C _X1-3 on the system side, that is, the differential mode component.

The resonance frequency of this parallel resonance circuit coincides with the resonance frequency of the circuit path A shown in FIG. Here, the inductance component of the parallel resonance circuit is equivalent to the leakage inductance of the common mode choke coil L _C, obtained by measuring and connected as shown in FIG.

For example, when the capacitances of the line capacitors C _X1-1 to C _X1-3 and C _X2-1 to C _X2-3 are 0.15 [μF], and the leakage inductance measured by the connection shown in Fig. 7 is 8 [μH]. The resonance frequency of is 252 [kHz], which is almost the same as the resonance frequency shown in FIG. As a countermeasure method at this time, there is a method of lowering the resonance frequency to the outside of the standard target frequency range. Specifically, when the capacitances of the line capacitors C _X1-1 to C _X1-3 and C _X2-1 to C _X2-3 are increased to 1.0 [μF], the resonance frequency becomes 97 [kHz]. As a result, as shown in FIG. 8, the resonance frequency moves to the standard lower limit frequency 150 [kHz] or less as intended, so that the standard can be satisfied.

In general, the size and exciting inductance of the common mode choke coil are often determined according to the countermeasures against the common mode component, and it is difficult to intentionally adjust the leakage inductance. As a result, in order to adjust the resonance frequency low, the capacitance of the line capacitor is increased.

However, as described above, in order to adjust the resonance frequency only by the capacitance of the line capacitor, it is often necessary to change to a very large capacitance, and there is a problem that the noise filter becomes large.

The noise filter in one embodiment according to the present disclosure reduces the deterioration of conduction noise due to resonance of the path converted from the common mode component to the differential mode component. Further, according to the noise filter in one embodiment according to the present disclosure, it is possible to suppress the increase in size of the noise filter by setting the capacitance of each line capacitor to an appropriate value.

FIG. 9 is a diagram showing a configuration example of a noise filter in one embodiment. The noise filter 18C shown in FIG. 9 is different from the noise filter 18B shown in FIG. 3 in that the line capacitors C _X1-1 to C _X1-3 on the system side are connected by Δ.

Noise filter 18C includes a common mode choke coil L _C, a plurality of line-to-line capacitor C _X1-1 ~ C _X1-3 are Δ-connected to the input side of the common mode choke coil L _C, the common mode choke coil L _{C Between} the neutral point 16 and the ground wire 17 of the _{plurality of line capacitors C X2-1} to C _X2-3 and the plurality of line capacitors C _X2-1 to C X2-3 that are Y-connected to the input side. It is equipped with one grounding capacitor C _{Y to be connected.} The input terminals R, S, T and the input ground terminal E are the terminals on the input side of the noise filter 18C, and the intermediate terminals R', S', T'and the intermediate ground terminal E'are on the output side of the noise filter 18C. It is a terminal.

The common mode choke coil L _C has a structure in which a power line for three phases is wound around one core, and the end of each input side of the power line wound around the core is connected to the input terminals R, S, T. And the end of each output side is connected to the intermediate terminals R', S', T'.

The plurality of line-to-line capacitors C _X1-1 to C _X1-3 are examples of a plurality of input capacitors, and are delta-connected to the input terminals R, S, and T sides. The line capacitor C _X1-1 is connected between the R phase power line and the S phase power line, and the line capacitor C _X1-2 is connected between the R phase power line and the T phase power line. The line capacitor C _X1-3 is connected between the S-phase power line and the T-phase power line.

The plurality of line-to-line capacitors C _X2-1 to C _X2-3 are examples of a plurality of output capacitors, and are star-connected to the intermediate terminals R', S', and T'. The line capacitor C _X2-1 is connected between the R phase power line and the neutral point 16, and the line capacitor C _X2-2 is connected between the R phase power line and the neutral point 16. The line capacitor C _X2-3 is connected between the T-phase power line and the neutral point 16.

One grounding capacitor C _Y is connected between the neutral point 16 and the ground wire 17. The ground wire 17 is an example of a ground, and connects between the input ground terminal E and the intermediate ground terminal E'. The number of ground terminals of the noise filter may be one or a plurality, and in this example, the case of two (input ground terminal E and intermediate ground terminal E') is shown.

The parallel resonance of the above-mentioned mode-converted component of the noise filter 18C of FIG. 9 is determined by the path B of the cycle of the solid arrow shown in FIG. FIG. 10 is a diagram showing an example of a circuit path of the parallel resonant circuit of the noise filter 18C shown in FIG. Parallel resonance circuit is formed by the common mode choke coil L _C and a plurality of lines between the capacitors C _X1-1 ~ C _X1-3 and a plurality of lines between the capacitors C _X2-1 ~ C _X2-3 and grounding capacitor C _Y To.

As shown in FIG. 10, the path B passes through _{two line capacitors C X1-1} and C _{X1-2 connected in parallel on the system side.} Therefore, the combined capacitance of the resonance path on the system side is larger than that in the form in which _{the line capacitors C X1-1} to C _X1-3 on the system side are connected by Y. Specifically, for example, each capacitance of a plurality of line capacitors (Δ connection C _X1-1 to C _X1-3 , Y connection C _X2-1 to C _X2-3 ) is 0.15 [μF], as shown in FIG. The resonance frequency when the leakage inductance measured by the connection is 8 [μH] is 205 [kHz], and the resonance frequency can be significantly lowered as compared with the reference form (FIG. 3) of 252 [kHz] in which all Y connections are made.

The standard can be satisfied by setting the resonance frequency of the parallel resonant circuit having the one-round path B determined in FIG. 10 to the standard lower limit frequency 150 [kHz] or less.

FIG. 11 is an example of a simple simulation result of the noise terminal voltage in each case of FIG. 4 (reference mode (FIG. 3)), FIG. 8 (moving to 150 kHz or less), and FIG. 9 (system side Δ connection). According to FIG. 11, simulation results in which the tendencies match are obtained at about 1 [MHz] or less, and as described above, it is possible to confirm how the resonance frequency changes.

In addition, under the standard mode and the conditions for system side delta connection, the measured value greatly exceeds the quasi-peak value regulation value (solid line) of the conduction noise regulation value (IEC61800-3) of the variable motor speed drive system (PDS). do. On the other hand, under the condition that the resonance frequency is moved to 150 kHz or less, it can be predicted that all the measured values can clear the mean value regulation value (dotted line). In other words, it can be confirmed from the simulation results that the behavior of the observed conduction noise can be roughly reproduced, although there is some error in the absolute value of the peak value of the resonance frequency.

From the above, the resonance frequency of the component that converts from the common mode component to the differential mode component by making a Δ connection only to the line capacitor on the system side as in the noise filter 18C shown in FIG. 9 is set to the line capacitor with the same capacitance. Can be lower than Y connection.

As a result, the noise filter 18C can be made smaller because the capacity of the line capacitor required to lower the resonance frequency until the standard can be satisfied can be made smaller. Since the noise filter 18C has a Y-connected line capacitor on the inverter side, the problem of insulation distance from the ground potential is unlikely to occur.

Further, as described above, when the line-to-line capacitor is connected by Δ, a component having a higher withstand voltage than that of Y connection must be applied. When a filter board having the same capacity of 200V system and 400V system is shared, it is often necessary to make a Y connection in the past. However, in the case of the noise filter in one embodiment according to the present disclosure, it is preferable to prepare a mounting pattern on the same board that allows the line capacitor on the system side to be connected to either Y connection or Delta connection on the filter board. .. In other words, the noise filter configuration can be rearranged relatively easily by making a delta connection only when applying to a 200V system model.

The resonance frequency of the component that converts from the common mode component to the differential mode component is determined by the one-round path B shown in FIG. Here, even for the same resonance frequency, than the current through the line between the capacitor C _X1-1 -C _X1-3 the system side, the line capacitor C _X2-1 the inverter side, the C _X2-2 If the current returned to the intermediate terminals R'and S'is increased, the conduction noise flowing out to the system side can be reduced. That may be smaller than the plurality of lines between the capacitances of the capacitors C _X2-1 ~ C _X2-3 of each capacitance inverter side of the plurality of lines between the capacitor C _X1-1 -C _X1-3 the mains.

In particular, the ratio of the capacitance of each capacitance and the inverter side of the plurality of lines between the capacitors C _X2-1 ~ C _X2-3 number of lines between the capacitor C _X1-1 -C _X1-3 the mains 1: 3 or more Is more effective. In other words, each capacitance of the plurality of lines between the capacitors C _X2-1 ~ C _X2-3 is, if it is more than three times the respective capacitances of the plurality of lines between the capacitor C _X1-1 -C _X1-3, more effective Is. More preferably, the capacitances of the plurality of lines between the capacitors C _X2-1 ~ C _X2-3 is, if it is less than 10 times 5 times or more of the capacitances of a plurality of lines between the capacitor C _X1-1 -C _X1-3 , Even more effective.

Next, the difference in the reduction effect due to the difference in the circuit constant will be described based on the simulation result.

FIG. 12 is a diagram showing an example of a simulation result in which the ratio of the capacitance of the line capacitor on the system side and the inverter side is changed while the resonance frequency is kept substantially constant from the reference condition of FIG. FIG. 13 is an enlarged view of a part of the frequency range of FIG. When the resonance peak near 250 [kHz] is confirmed, it can be confirmed that the resonance peak decreases as the ratio increases. Further, when the ratio is 1: 3 or more, a reduction effect of about 10 [dB] or more can be obtained, and it can be confirmed that the peak can be reduced more effectively.

_{FIG. 14 shows the capacitances of the line capacitors C X2-1} to C _X 2-3 on the inverter side without changing the capacitance of the line capacitor on the system side from the reference conditions of FIG. 11 (0.15 [μF]). It is a figure which shows an example of the simulation result at the time of increasing the ratio by increasing. FIG. 15 is an enlarged view of a part of the frequency range of FIG. The resonance frequency at the resonance peak in FIG. 15 moves in the lower direction as the ratio of the line capacitor on the system side and the inverter side increases. Further, the amount of reduction of the resonance peak is also larger than that in FIG. It is considered that this is because _{the current flowing into the line capacitors C X2-1} to C _X2-3 on the inverter side became large, and as a result, the outflow of conduction noise to the system side was suppressed.

_{In FIG. 16, the capacitances of the line capacitors C X1-1} to C _X1-3 on the system side are reduced without changing the capacitance of the line capacitors on the inverter side (0.15 [μF]) from the reference conditions of FIG. It is a figure which shows an example of the simulation result when the ratio is increased by this. FIG. 17 is an enlarged view of a part of the frequency range of FIG. The resonance frequency at the resonance peak in FIG. 17 moves in a higher direction as the capacitance of the line capacitor on the system side becomes smaller than the capacitance of the line capacitor on the inverter side. Further, the amount of reduction of the resonance peak is also larger than that in FIG. The capacitance of the line capacitor can be made smaller than the standard condition.

In FIG. 18, the ratio is increased by reducing the capacitances of _{the line capacitors C X1-1} to C _X1-3 on the system side based on the condition that the capacitances of all the line capacitors are fixed to 0.3 [μF]. It is a figure which shows an example of the simulation result in the case of. FIG. 19 is an enlarged view of a part of the frequency range of FIG. Similar to FIG. 17, the resonance frequency at the resonance peak in FIG. 19 moves in a higher direction as the capacitance of the line capacitor on the system side becomes smaller than the capacitance of the line capacitor on the inverter side. When each capacitance of the line capacitor on the system side is 1/10 times (0.03 [μF]), the resonance frequency is 500 [kHz] or less, which can be sufficiently reduced from the quasi-peak value regulation value. You can check it.

In this way, in the noise filter 18C, the _{capacitances of the line capacitors C X1-1} to C _X1-3 on the system side are made smaller than the capacitances of the line capacitors C _X2-1 to C _{X2-3 on the inverter side.} By selecting the above, it is possible to appropriately reduce the components that are converted from the common mode component to the differential mode component. As a result, both standard satisfaction and filter miniaturization can be achieved.

Although the noise filter and the power conversion device have been described above by the embodiment, the present invention is not limited to the above embodiment. Various modifications and improvements, such as combinations and substitutions with some or all of the other embodiments, are possible within the scope of the present invention.

16 Neutral point 17 Ground wire 18 Noise filter 19 Converter part 20 Inverter part 90 AC power supply 91 Input cable 93 Reference ground surface 94 Motor 100 Motor drive system 200 Inverter 300 Power converter
L _C common mode choke coil

Claims (5)

Common mode choke coil and
A plurality of input capacitors delta-connected to the input side of the common mode choke coil, and
A plurality of output capacitors star-connected to the output side of the common mode choke coil, and
A noise filter comprising a grounding capacitor connected between the neutral points of the plurality of output capacitors and ground. The noise filter according to claim 1, wherein the resonance frequency of the parallel resonant circuit formed by the common mode choke coil, the plurality of input capacitors, the plurality of output capacitors, and the grounded capacitor is lower than 150 kHz. The noise filter according to claim 1 or 2, wherein each capacitance of the plurality of input capacitors is smaller than each capacitance of the plurality of output capacitors. The noise filter according to claim 3, wherein each capacitance of the plurality of output capacitors is at least three times the capacitance of each of the plurality of input capacitors. The noise filter according to any one of claims 1 to 4 and an inverter connected to the noise filter are provided.
The inverter is a power conversion device having a converter unit connected to the noise filter and an inverter unit connected to the converter unit.

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