CN109980978B - Converter and modulation method thereof - Google Patents
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/14—Arrangements for reducing ripples from dc input or output
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1582—Buck-boost converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
Abstract
The invention discloses a converter and a modulation method thereof, and belongs to the technical field of power electronic converters. Wherein one end of the DC side of the converter and the inductor LinIs connected to an inductor LinThe other end of the power switch tube S1The terminal 1 is connected with the conducting end of the switch device, and the cut-off end of the switch device is respectively connected with the power switch tube S2And S4Terminal 1 and capacitor C1One terminal of (C), a capacitor1The other ends of the two are respectively connected with a power switch tube S3And S5Terminal 2, the other end of the converter on the DC side, power switch tube S1And S2Terminal 2 and power switch tube S3The terminal 1 and one end of the AC side are both grounded, and the power switch tube S5Terminal 1 and power switch tube S4The terminal 2 is connected to the other end on the ac side. In addition, a corresponding modulation method, a bidirectional converter and a corresponding modulation method are also provided. Aiming at the problem of high-frequency leakage current of the traditional boost inverter, the boost inverter is high in integration level and small in size, and can effectively solve the problem of high-frequency leakage current.
Description
Technical Field
The invention relates to the technical field of power electronic converters, in particular to a converter and a modulation method thereof.
Background
With the development of the world economy, a large amount of non-renewable fossil energy is rapidly consumed, so that the problems of energy crisis, environmental pollution and the like are caused. The electric vehicle, as a new energy vehicle developed first, can alleviate the above technical problems, but the storage battery is a key factor restricting its development, and a vehicle-mounted charging system, which is one of the storage battery charging systems, generally includes 2 converters working in cascade, a bidirectional rectifier for charging the storage battery, an inverter for driving a motor, and the converter generally includes a large inductor and a large capacitor. This configuration undoubtedly increases the weight and cost of the electric vehicle, and occupies a limited space within the electric vehicle. Therefore, on the basis of ensuring good charging characteristics of the storage battery of the electric automobile, foreign scholars propose to research an integrated hybrid topological structure of the driving system of the electric automobile and the storage battery, namely, a hardware structure of the traction driving system is reconstructed into a storage battery charging device, and a converter is controlled by optimizing a topology and a control strategy to respectively complete rectification, inversion and power factor correction, so that the functions of motor driving, high-power factor charging, harmonic wave treatment and the like are realized, and the advantages of the vehicle-mounted charging system in charging quality, weight and cost are hopefully improved.
The distributed photovoltaic power generation system is an effective method for coping with energy crisis and realizing cleanness, environmental protection and low-carbon economy, however, the output voltage of a photovoltaic cell panel in the system is generally lower, and a commonly adopted solution in the prior art is to adopt a two-stage converter in a low-voltage renewable energy grid-connected power generation system, namely, a DC/DC module is added between a traditional inverter and the photovoltaic cell panel so as to improve the input end voltage of the traditional inverter. However, the method is not only complex in control, but also multiple in devices and low in conversion efficiency.
In the modulation method for realizing boosting inversion by combining a DC/DC converter and an inverter in series in the prior art, a strategy of separately modulating the pre-stage boosting and the post-stage inversion is adopted, the modulation method is complex and difficult to control, and the ripple variation of the voltage at the direct current input end seriously affects the quality of the voltage at the alternating current output end.
In order to overcome the problem of low voltage of the traditional inverter, a Z-source inverter is proposed in the prior art, the defects of the traditional voltage source inverter are overcome, the direct current bus voltage at the input side of the inverter is improved by utilizing the controllable direct connection of power switching tubes of upper and lower bridge arms, so that the output alternating current voltage is improved, in grid-connected application, the isolation transformer is generally required to be connected with a power grid, but the volume and the cost of a system are increased due to the connection of the isolation transformer, and the power generation efficiency is reduced. When the isolation transformer is not used, the problem of high-frequency leakage current exists, and the harmonic content of a power grid and the system loss are increased.
The article, "dual-mode control wide input voltage combined boost inverter" is disclosed in the journal of the Chinese Motor engineering, volume 28, No. 7, published as 2018, 4 and 5, and the authors: the circuit topology structure is formed by cascading a T-shaped network and a Buck-type inverter bridge, wherein the T-shaped network is equivalent to a Buck-Boost direct current converter. The inverter has the voltage boosting and reducing capacity by utilizing the lifting capacity of the T-shaped network, can adapt to the input direct-current voltage with wide range change, and is suitable for application occasions with wide range fluctuation of the input voltage; under two independent control loops, saturation and zero setting of duty ratio signals are used as switching time points, and smooth switching of two modes of voltage boosting and voltage reducing is achieved. The Buck inverter bridge has the disadvantages that the circuit needs more components, so that the loss is large, the conversion efficiency is low, and the Buck inverter bridge can generate large high-frequency common-mode voltage in a unipolar modulation mode, so that high-frequency leakage current is generated, and the harmonic content of a power grid and the system loss are increased.
Disclosure of Invention
1. Technical problem to be solved
The invention provides a converter and a modulation method thereof, aiming at the problems that the voltage of the traditional boost inverter is not high, the existing improved technology has more components and parts, larger loss, low conversion efficiency, high-frequency leakage current and complex control scheme and the like.
2. Technical scheme
The purpose of the invention is realized by the following technical scheme.
A converter comprises a power switch tube S1、S2、S3、S4And S5Inductance LinAnd a capacitor C1And further comprising a switching device; one end of the DC side of the converter and an inductor LinIs connected to an inductor LinThe other end of the power switch tube S1Is connected to the conducting terminal of a switching device, the switching deviceThe cut-off ends of the two are respectively connected with a power switch tube S2And S4Terminal 1 and capacitor C1One terminal of (C), a capacitor1The other ends of the two are respectively connected with a power switch tube S3And S5A terminal 2; the other end of the DC side, a power switch tube S1And S2Terminal 2, power switch tube S3The terminal 1 and one end of the AC side of the converter are both grounded; power switch tube S5Terminal 1 and power switch tube S4The terminal 2 is connected to the other end on the ac side.
Further, a capacitor C1The capacitance value of (A) is smaller and is a polar capacitor or a non-polar capacitor.
Further, the switching device may be a diode D or a power switch S6The conducting end of the switch device is the anode of a diode D or a power switch tube S6Terminal 1, switching device cut-off terminal is diode D cathode or power switch tube S6And a terminal 2.
Furthermore, the power switch tube S2、S3、S4And S5Are connected in anti-parallel with a diode at both ends, i.e. power switch tube S2、S3、S4And S5The terminals 2 are all connected with the anode of the diode and the power switch tube S2、S3、S4And S5Are connected to the cathode of the diode.
Furthermore, the power switch tube S3And S5Are connected in anti-parallel with a diode at both ends, i.e. power switch tube S3And S5The terminals 2 are all connected with the anode of the diode and the power switch tube S3And S5Are connected to the cathode of the diode.
Furthermore, an improved converter circuit is provided, the conducting end of the switch device is a power switch tube S6Terminal 2, the cut-off end of the switching device is a power switch tube S6The terminal 1 of (1); power switch tube S3、S4、S5And S6Are connected in anti-parallel with diodes at both ends, i.e. power switch tube S3、S4、S5And S6Are connected to the anode of a diode, a power switchPipe S3、S4、S5And S6Are connected to the cathode of the diode.
Further, a bidirectional converter, a power switch S1And S2Diodes with both ends antiparallel, i.e. power switching tubes S1And S2The terminals 2 are all connected with the anode of the diode and the power switch tube S1And S2Are connected to the cathode of the diode.
A modulation method of a converter, according to the converter, in the whole power frequency period, a power switch tube S1Working in an SPWM state all the time; in the positive half period of power frequency, the power switch tube S1And S4Synchronous, power switching tube S3In the on state, the power switch tube S2And S5In an off state; in the negative half period of the power frequency, the power switch tube S3And S4Are all in an off state, and a power switch tube S5In the on state, the power switch tube S2And S1The working states are consistent.
Furthermore, when the conducting end of the switch device is the power switch tube S6The cut-off end of the switching device is a power switch tube S6Terminal 2, power switch tube S6And a power switch tube S1The working state is opposite.
Furthermore, when the conducting end of the switch device is the power switch tube S6Terminal 2 of the switching device is a power switch tube S6The terminal 1 of (1); power switch tube S3、S4、S5And S6When the two ends of the power switch tube S are connected with the diodes in an anti-parallel mode, the power switch tube S6Always operating in the off state.
Furthermore, in the case of a bidirectional inverter, the power switch tube S is operated in a forward inversion mode6Is always in the off state; during reverse rectification operation, the power switch tube S is in the whole power frequency period1Always working in SPWM state and power switch tube S3And S4And S1Synchronous, power switching tube S2、S5And S6And S1The state is reversed.
3. Advantageous effects
Compared with the prior art, the invention has the advantages that:
(1) according to the scheme, the direct current side and the alternating current side are grounded together, so that common mode interference is avoided, and high-frequency leakage current does not exist;
(2) compared with a full-bridge inverter, the scheme only adds 1 power switching tube, and compared with some circuit structures for restraining high-frequency leakage current such as H6 and the like, the number of the power switching tubes is small, so that the use of the power switching tubes is reduced, the loss is reduced, and the conversion efficiency of a system is improved;
(3) according to the modulation method, in each working mode, at most three power switching tubes work, so that the conduction loss of the switching tubes is reduced;
(4) compared with a full-bridge inverter, the bidirectional converter in the scheme only increases 2 power switching tubes, and compared with some circuit structures for restraining high-frequency leakage current such as H6 and the like, the bidirectional converter has the advantages that the number of the power switching tubes is small, so that the use of the power switching tubes is reduced, the loss is reduced, and the conversion efficiency of a system is improved;
(5) the scheme is that the capacitor C1An energy storage element for converting energy and outputting AC, and a capacitor C1The value of the capacitor is flexible, and a non-polar thin film capacitor can be used, so that the circuit works reliably, and the service life of the circuit is prolonged;
(6) compared with the traditional two-stage boost inverter or grid-connected inverter which needs two groups of filters, the scheme has no high-frequency leakage current, only needs one filter and occupies small space;
(7) this scheme can carry out reactive compensation in certain extent, can not cause the influence to the direct current side moreover.
Drawings
FIG. 1 is a circuit configuration according to an embodiment of the present invention;
FIG. 2 illustrates a modulation strategy according to an embodiment of the present invention;
FIG. 3 is a diagram illustrating the distribution of operating modes in a switching cycle in the DCM state according to an embodiment of the present invention;
FIG. 4 is a schematic diagram illustrating a circuit operation mode according to an embodiment of the present invention;
FIG. 5 is a diagram illustrating a second circuit operation mode according to an embodiment of the present invention;
FIG. 6 is a schematic diagram of a third exemplary embodiment of a circuit operation mode;
FIG. 7 is a diagram illustrating four exemplary modes of operation of a circuit according to an embodiment of the present invention;
FIG. 8 is a five schematic diagram illustrating a circuit operation mode according to an embodiment of the present invention;
FIG. 9 is a six-schematic diagram illustrating a circuit operation mode according to an embodiment of the invention;
FIG. 10 is a diagram illustrating the distribution of operating modes in a switching cycle in a CCM state according to an embodiment of the present invention;
FIG. 11 is a graph of voltage and current simulation waveforms in full DCM according to an embodiment of the present invention;
FIG. 12 is a waveform illustrating the simulation of the inductor current in the full DCM of one embodiment of the present invention;
FIG. 13 is a diagram of simulated voltage and current waveforms in a CCM and DCM coexistence state according to an embodiment of the present invention;
FIG. 14 is a waveform diagram illustrating inductor current simulation in a CCM and DCM coexistence state according to an embodiment of the present invention;
FIG. 15 is a diagram of a main simulation waveform when no power is output according to an embodiment of the present invention;
FIG. 16 shows a capacitor C according to an embodiment of the present invention1When the current is 10 mu F, outputting a voltage and current simulation oscillogram;
FIG. 17 shows a capacitor C according to an embodiment of the present invention1When the voltage is 10 mu F, the THD value of the output voltage is obtained;
FIG. 18 shows a capacitor C according to an embodiment of the present invention1When the current is 100 mu F, outputting a voltage and current simulation oscillogram;
FIG. 19 shows a capacitor C according to an embodiment of the present invention1When the output voltage is 100 mu F, the THD value of the output voltage is obtained;
fig. 20 is a circuit configuration of a unidirectional converter of the present invention;
fig. 21 is a circuit configuration of another unidirectional converter of the present invention;
fig. 22 shows a circuit configuration of a bidirectional converter according to the present invention.
Detailed Description
For a further understanding of the present invention, reference will now be made in detail to the embodiments illustrated in the drawings.
The present application will be described in further detail with reference to the following drawings and examples. It is to be understood that the specific embodiments described herein are merely illustrative of the relevant invention and not restrictive of the invention. It should be noted that, for convenience of description, only the portions related to the present invention are shown in the drawings.
The terms first, second, one end, the other end and the like in the present invention are provided for convenience of describing the technical scheme of the present invention, have no specific limiting effect, are all general terms, and do not limit the technical scheme of the present invention.
It should be noted that the embodiments and features of the embodiments in the present application may be combined with each other without conflict. The present application will be described in detail below with reference to the embodiments with reference to the attached drawings.
The power switch tube of the invention comprises a power switch tube S1、S2、S3、S4、S5And S6The power switch tube is a MOSFET device, an IGBT or a triode and other power switch tubes. When using IGBT or triode, power switch tube S1-S6The terminals 1, 3 and 2 respectively represent the power switch tube S1-S6Collector, base and emitter of (1), power switching tube S when MOSFET is used1-S6The terminals 1, 3 and 2 respectively represent the power switch tube S1-S6Drain, gate and source.
The filter output voltage, i.e. the voltage across the load or the network, is denoted as the ac output voltage uoCorresponding to an AC output voltage of amplitude UomThe filter output current, i.e. the current flowing through the load or the network, is denoted as output current ioCorresponding to the amplitude of the output current of the filter being IomThe filter input voltage, i.e. the voltage between circuit nodes E and F, isUEF。
Example 1
As shown in fig. 1, the converter of the present embodiment includes a power switch tube S1、S2、S3、S4And S5Inductance LinAnd a capacitor C1And further comprises a diode D; one end of the DC side of the converter and an inductor LinIs connected to an inductor LinThe other end of the power switch tube S1The terminal 1 is connected with the anode of a diode D, and the cathode of the diode D is respectively connected with a power switch tube S2And S4Terminal 1 and capacitor C1One terminal of (C), a capacitor1The other ends of the two are respectively connected with a power switch tube S3And S5Terminal 2, the other end of the converter on the DC side, power switch tube S1And S2Terminal 2, power switch tube S3The terminal 1 and one end of the AC side of the converter are grounded; power switch tube S5Terminal 1 and power switch tube S4The terminal 2 is connected to the other end of the inverter on the ac side. Wherein, the power switch tube S2、S3、S4And S5Both ends of the diode are connected with the diode in an anti-parallel mode, and the function of follow current is achieved.
Further, as shown in fig. 1, the dc side is a dc power supply UinIn practical application, the output voltage of the photovoltaic cell panel is connected with the AC side, nodes E and F are connected in parallel at the input end of a filter, the output end of the filter is connected in parallel with a load or a power grid (the converted electric energy is directly fed back to the power grid), and the voltage U of the filter is measured by a voltage measuring circuitEFFiltering is performed to remove harmonic interference, the filtering can be selected according to an actual application scenario, for example, an LC filter or an LCL filter can be shown in fig. 1, the output of the filter can be connected to a power grid or a load, the load characteristic can also be selected according to the actual application scenario, the filtering can be a resistive load, an inductive load or a capacitive load, and when a non-resistive load is loaded, reactive compensation can be performed on the output with a power factor smaller than 1.
Further, a capacitor C1The capacitance value of the capacitor is smaller, and the capacitor is a non-polar capacitor or a polar capacitor, so that the space occupied by the circuit is further reduced.
Further, when nodes E and F are connected in parallel with the LC filter, as shown in FIG. 1, node E is connected to filter inductor L0Is connected to node F and capacitor C0Is connected to one end of a filter inductor L0Another terminal of (1) and a filter capacitor C0Is connected with one end of a current source, a filter capacitor C0The other end of the current source and the other end of the current source are both grounded.
In order to realize the effect of boosting and inverting the direct current power supply on the direct current side, most of the prior art adopts the technical scheme of combining a DC/DC converter and an inverter in series, but still has the problems of more power switching devices, large loss, conversion efficiency and the likeinThe amplitude of the output voltage at the AC side is larger than that of the DC power supply U for conversioninThe power supply has the advantages of low switching loss and circuit cost due to the fact that the number of components is small, especially the number of power switching devices is small, high boosting transformation ratio, small size and light weight, and can be widely popularized and applied.
Example 2
According to the inverter described in embodiment 1, this embodiment proposes a modulation method of the inverter, as shown in fig. 2, the modulation wave is an absolute value of a sine wave, and an SPWM wave is obtained by comparing the absolute value with a triangular carrier wave as a power switch tube S1The switching signal of (1); in the whole power frequency period, the power switch tube S1Working in an SPWM state all the time; in the positive half period of power frequency, the power switch tube S1And S4Synchronous, power switching tube S3In the on state, the power switch tube S2And S5In an off state; in the negative half period of the power frequency, the power switch tube S3And S4Are all in an off state, and a power switch tube S5In the on state, the power switch tube S2And S1The working states are consistent.
2.1 principle of operation analysis
2.1.1 working principle analysis in DCM State
Table 1 shows the switching timing corresponding to the modulation method in fig. 3, which is the switching timing of the converter of this embodiment in a DCM state within one switching period.
TABLE 1 switching sequence of power switching tube in DCM state
The working principle of the converter is analyzed in detail in connection with table 1. According to the direction of the output current, the flow from left to right through the filter inductor L is defined0The direction of current flow in the positive direction is divided into six modes, as shown in fig. 4-9. The output filter may be selected from an LC filter, an LCL filter, a single inductor filter, or the like. An LC filter is preferred as the output filter in this embodiment.
The output current being greater than zero, i.e. at the AC output voltage uoIn the positive half period, the mode one, the mode two and the mode three alternately operate and work; the output current being less than zero, i.e. at the AC output voltage uoIn the negative half period, the mode four, the mode five and the mode six alternately operate and work.
Mode one [ t ]0-t1]
As shown in fig. 3, at t0Before the moment, the power switch tube S3Conduction, inductance LinCurrent iLin(t) is zero; at t0Time of day, power switch tube S1And S4Start of conduction, S3Maintain conduction, power switch tube S2And S5And is turned off, the diode D is turned off, and the current flow path is shown by the dotted line in fig. 4.
At [ t ]0-t1]In the working phase of (1), the DC power supply UinInductor LinAnd a power switch tube S1Forming a closed loop, inductor LinThe voltage at both ends is a DC power supply UinDC power supply UinTo the inductance LinCharging energy storage, flowing through inductance LinCurrent i ofLin(t) increases linearly until t1At that time, modality one ends.
Capacitor C1Power switch tube S3Power switch tube S4Filter inductor L0A positive closed loop formed by the current source and a capacitor C1Discharging is carried out, the capacitor C1The stored energy is released to charge the current source, and the AC side outputs a voltage uoRise while due to input of power UinAnd an AC output voltage uoCommon ground avoids common mode interference and high frequency leakage current is zero.
Modal two [ t ]1-t2]
As shown in fig. 3, at t1Time of day, power switch tube S1And S4Turn-off, power switch tube S3Maintain conduction, power switch tube S2And S5Remains off, diode D is turned on and the current flow path is shown by the dashed line in fig. 5. At [ t ]1-t2]In the working phase of (1), the DC power supply UinInductor LinDiode D and capacitor C1And a power switch tube S3Forming a closed loop, inductor LinSubject to reverse voltage Uin-UC1Discharge to the capacitor C1Charging and storing energy, flowing through inductor LinCurrent i ofLin(t) a linear decrease; power switch tube S3Power switch tube S5Reverse diode and filter inductor L with two ends connected in parallel0A current source to form a follow current loop, a filter inductor L0Cannot be abruptly changed to freewheel the current source until t2Time of day, inductance LinCurrent i ofLin(t) linearly decreases to zero and mode two ends.
It should be added that: power switch tube S3On the one hand, a capacitance C1Charging; on the other hand through the power switch tube S3And the diodes with two anti-parallel ends output follow current for the alternating current side so as to obtain a sine waveform.
Modal three [ t ]2-t3]
As shown in fig. 3, at t2Time of day, inductance LinCurrent i ofLin(t) linear reduction to zero, power switch tube S3Maintain conduction, power switch tube S1、S2、S4And S5Maintain shutdownWhen the diode D is turned off, the current flow path is shown by the dotted line in FIG. 6, and the power switch tube S3、S5Reverse diode and filter inductor L with two ends connected in parallel0The current source still forms a follow current loop, and the filter inductor L0The current source is still freewheeling until t3At time, modality three ends. The modes one to three are then cycled in DCM state during the positive half cycle of the sinusoidal modulation wave.
Modal four [ t ]4-t5]
As shown in fig. 3, at t4Before the moment, the power switch tube S5Conduction, inductance LinCurrent iLin(t) is zero; at t4Time of day, power switch tube S1And S2Starting to conduct, the power switch tube S5Maintain conduction, power switch tube S3And S4And is turned off, the diode D is turned off, and the current flow path is shown by the dotted line in fig. 7. At [ t ]4-t5]In the working phase, the DC power supply UinInductor LinAnd a power switch tube S1Forming a closed loop, inductor LinThe voltage at both ends is DC input voltage UinDC power supply UinTo the inductance LinCharging energy storage, flowing through inductance LinCurrent i ofLin(t) linear growth.
Capacitor C1Power switch tube S2Power switch tube S5Filter inductor L0A reverse closed loop formed by the capacitor C and the load or the power grid1Discharge, capacitance C1Releasing the stored energy, reverse charging the current source, and outputting an AC output voltage uoRise due to the DC power supply UinAnd an AC output voltage uoCommon ground, common mode interference is avoided, and high-frequency leakage current is zero until t5At time, modality four ends.
Modal five [ t ]5-t6]
As shown in fig. 3, at t5Time of day, power switch tube S1And S2Turn-off, power switch tube S5Maintain conduction, power switch tube S3And S4Keep off, diode D is ledThe path of the current flow is shown by the dotted line in fig. 8. At [ t ]5-t6]In the working phase, the DC power supply UinInductor LinDiode D and capacitor C1And a power switch tube S3A closed loop consisting of two parallel reverse diodes, inductor LinSubject to reverse voltage Uin-UC1Discharge to the capacitor C1Charging and storing energy, flowing through inductor LinCurrent i ofLin(t) decreases linearly.
Power switch tube S3Reverse diode and power switch tube S with two ends connected in parallel5Filter inductor L0A reverse follow current loop formed by the inductor L and the load or the power grid0The current on the current source can not be suddenly changed, and the current source is reversely fed with follow current until t6At the moment, the current i of the inductorLin(t) decreases linearly to zero and mode five ends.
Mode six [ t ]7-t8]
As shown in fig. 3, at t7Time of day, inductance LinCurrent i ofLin(t) linear reduction to zero, power switch tube S5Maintain conduction, power switch tube S1、S2、S3And S4Keeping off, the diode D is turned off, the current flow path is shown by the dotted line in FIG. 9, and the power switch tube S3Reverse diode and power switch tube S with two ends connected in parallel5Filter inductor L0The current source and the current source still form a reverse follow current loop, and the filter inductor L0The current source is still reversely fed with current until t8At time, modality six ends. And then circulating the modes from four to six in a DCM state of the negative half period of the sinusoidal modulation wave.
2.1.2 CCM operating principle analysis
Table 2 shows the switching timing corresponding to the modulation method in fig. 10, which is the switching timing of the converter in this embodiment in a switching cycle in the CCM state.
TABLE 2 switching sequence of power switching tube in CCM state
The operation principle of the converter in CCM state is analyzed in detail with reference to table 2. According to the direction of the output current, the flow from left to right through the filter inductor L is defined0Is a positive direction, divided into four modes, as shown in fig. 4, 5, 7 and 8.
Mode 1[ t ]0-t1]
At t0Before the moment, the power switch tube S3Conduction, inductance LinCurrent iLin(t) is not zero; at t0Time of day, power switch tube S1And S4Start of conduction, S3Maintain conduction, power switch tube S2And S5And is turned off, the diode D is turned off, and the current flow path is shown by the dotted line in fig. 4. At [ t ]0-t1]In the working phase, the DC power supply UinInductor LinAnd a power switch tube S1Forming a closed loop, inductor LinThe voltage at both ends is DC input voltage UinDC power supply UinTo the inductance LinCharging energy storage, flowing through inductance LinCurrent i ofLin(t) linear growth.
Capacitor C1Power switch tube S3Power switch tube S4Filter inductor L0A positive closed loop formed by the current source and a capacitor C1Discharging is carried out, the capacitor C1The stored energy is released and charges a current source to output an AC output voltage uoRise due to the DC power supply UinAnd an AC output voltage uoCommon ground, common mode interference is avoided, and high-frequency leakage current is zero until t1At that time, modality 1 ends.
Mode 2[ t ]1-t2]
At t1Time of day, power switch tube S1And S4Turn-off, power switch tube S3Maintain conduction, power switch tube S2And S5Remains off, diode D is turned on and the current flow path is shown by the dashed line in fig. 5. At [ t ]1-t2]In the working phase, the DC power supply U on the DC sideinInductor LinDiode D and capacitor C1And a power switch tube S3Forming a closed loop, inductor LinSubjected to reverse voltage Uin-UC1Discharge to the capacitor C1Charging and storing energy, flowing through inductor LinCurrent i ofLin(t) decreases linearly.
Power switch tube S3Power switch tube S5Reverse diode and filter inductor L with two ends connected in parallel0A current source to form a follow current loop, a filter inductor L0The current on the current source can not be suddenly changed, and the current source is continuously supplied until t2At the time, mode 2 ends, and then modes 1 to 2 are cycled in CCM state for the positive half cycle of the sinusoidal modulation wave.
Mode 3[ t ]3-t4]
At t3Before the moment, the power switch tube S5Conduction, inductance LinCurrent iLin(t) is not zero; at t3Time of day, power switch tube S1And S2Start of conduction, S5Maintain conduction, power switch tube S3And S4And is turned off, the diode D is turned off, and the current flow path is shown by the dotted line in fig. 7. At [ t ]3-t4]In the working phase, the DC power supply UinInductor LinAnd a power switch tube S1Forming a closed loop, inductor LinThe voltage at both ends is DC voltage UinDC power supply UinTo the inductance LinCharging energy storage, flowing through inductance LinCurrent i ofLin(t) linear growth.
Capacitor C1Power switch tube S2Power switch tube S5Filter inductor L0A reverse closed loop formed by the current source and a capacitor C1Discharging is carried out, the capacitor C1The stored energy is released to charge the current source reversely and output voltage uoRise while due to input of power UinAnd an AC output voltage uoCommon ground, common mode interference is avoided, and high-frequency leakage current is zero until t4At that time, modality 3 ends.
Mode 4[ t ]4-t5]
At t4Time of day, power switch tube S1And S2Turn-off, power switch tube S5Maintain conduction, power switch tube S3And S4The diode D is turned on with the diode D kept off, and the current flow path is shown by the dotted line in fig. 8. At [ t ]4-t5]In the working phase, the DC power supply UinInductor LinDiode D and capacitor C1And a power switch tube S3A closed loop consisting of two parallel reverse diodes, inductor LinSubjected to reverse voltage Uin-UC1Discharge to the capacitor C1Charging and storing energy, flowing through inductor LinCurrent i ofLin(t) decreases linearly.
Power switch tube S3Reverse diode and power switch tube S with two ends connected in parallel5Filter inductor L0A reverse follow current loop formed by the current source and a filter inductor L0Cannot be abruptly changed to reversely freewheel the current source until t5Time of day, inductance LinCurrent i ofLin(t) decreases linearly to zero and mode 4 ends. The modes 3 to 4 are then cycled in the CCM state of the negative half cycle of the sinusoidal modulation wave.
As shown in fig. 2 and 10, only two high-frequency power switching tubes operate in the positive half period and the negative half period, so that the loss of the power switching tubes is reduced, the loss of the total devices is reduced, and the conversion efficiency is improved.
2.2 boost ratio analysis
By the modulation method of the present embodiment, as shown in fig. 1, the dc power supply UinDiode D and inductor LinAnd a power switch tube S1Combined, to a capacitor C1Charging is carried out, and a capacitor C1Storing energy, discharging to AC side, obtaining SPWM waveform voltage at front end of filter (between nodes E and F), blocking capacitor C by diode D1To a DC power supply UinEnergy is fed back, and then follow current is carried out through the power switch tube, and finally the processes of voltage boosting and inversion can be realized.
In combination with the above, according toFig. 4-9 correspond to six working modes, and the present embodiment is directed to the inductor LinCurrent iLin(t) in the case of DCM (inductance L)inCurrent iLin(t) in the case of CCM, a further analysis can be carried out, the analysis principle is similar and is not described here), a calculation analysis is carried out by combining the volt-second equilibrium principle, and the transformation ratio G is:
wherein, UomFor AC output voltage amplitude, UinIs the amplitude of the DC power supply; m is an amplitude modulation ratio, and m is more than or equal to 0 and less than or equal to 1; poIs the output power; l isinIs an inductor; f is power switch tube S1The switching frequency of (1).
The AC output voltage amplitude U can be obtained from the formula that m is more than or equal to 0 and less than or equal to 1omCan be larger than the amplitude U of the direct current power supplyinOr less than the amplitude U of the DC power supplyinThe converter of the embodiment can realize boosting inversion and also can realize step-down inversion.
Converter with combination of BOOST and full-bridge inversion, inductor LinThe energy storage function is the same as that of the BOOST converter; inductor LinCurrent i ofLin(t) either in an intermittent or continuous state; while in the present application the inductance L isinCurrent i ofLin(t) can be operated in an intermittent state, a continuous state, or an intermittent and continuous alternating state. In intermittent, continuously alternating or intermittent states, to avoid inductance LinThe energy on the power switch tube S is continuously accumulated to influence or damage the power switch tube S1. But current i in the present applicationLin(t) zero crossing, in the continuous state, the current iLinThe effect (t) is not good in the intermittent, continuous alternate state, or the intermittent state.
2.3 reactive compensation analysis
In the converter of the embodiment, after the harmonic wave on the alternating current side is filtered by the filter, the alternating current side can be directly connected in parallel to a power grid and properly adjustedThe converter can absorb the reactive current or energy in the power grid to the capacitor C by saving the phase and the amplitude of the output voltage at the alternating current side1Or sending out current corresponding to reactive current (power) in the power grid to offset the reactive current (power), thereby realizing the purpose of dynamic reactive compensation; meanwhile, the diode D in the converter structure can prevent reactive current from entering the direct current input source so as to avoid influencing or damaging a direct current power supply on the direct current side, and therefore safety is improved.
2.4 simulation verification
In order to verify the working performance of the transducer, the embodiment establishes a 1000W transducer model in Matlab/Simulink software for verification, and the main parameters of a prototype are shown in table 3.
TABLE 3 selection of devices
The abscissa in FIGS. 11-19 represents time t (10ms/div), and each bin represents 10 ms; the ordinate represents the voltage or current, corresponding to the designation of the respective parameter. As shown in fig. 11 and 12, the waveforms of the ac output voltage, the output current, the dc input voltage and the input inductor current in the full DCM state, and as can be seen from fig. 11 and 12, the converter has a boost characteristic, which can meet the requirement of grid connection of the photovoltaic panel. As shown in fig. 13 and 14, in the CCM and DCM coexistence state: the waveforms of the ac output voltage, output current and input voltage and input inductor current, and as can be seen from fig. 13 and 14, the converter has a boost characteristic, which can meet the requirement of grid connection of the photovoltaic cell panel. As shown in fig. 15, the waveforms of the output voltage and the output current during reactive compensation, and it can be seen from fig. 15 that the converter has the function of reactive compensation.
As can be seen from FIGS. 16 and 18, the capacitance C1Can be a capacitor with small capacitance value and hardly generates output voltage and output currentThe influence is caused, the reliability of the circuit is greatly improved, and as can be seen from fig. 17 and 19, by adopting the modulation method of the present application, a capacitor with a smaller capacitance value, such as a non-polar capacitor, can be adopted, the THD requirement of the output voltage can be met, and the simulation result well verifies the working principle and the steady-state characteristic of the converter.
By adopting the modulation method, the power switch tube S1And S4After synchronization, the ripple fluctuation of the direct-current power supply on the direct-current side does not affect the alternating-current output voltage on the alternating-current side, an additional control strategy is not needed to be adopted for the ripple fluctuation of the direct-current power supply, and a control method is simplified. In order to achieve the same THD level, the capacitor connected in parallel with the DC bus voltage in most of the BOOST and full-bridge inversion combined BOOST inverters adopted in the prior art and the capacitor C in the application are connected in parallel1The values are the same, but the existing BOOST and full-bridge inversion combined BOOST inverter also needs to play a role in stabilizing direct-current power supply fluctuation, an electrolytic capacitor with a larger capacitance value needs to be selected, and the capacitor C of the application1During operation, the non-polar capacitor with a small capacitance value can be selected to be used only for energy transfer, so that the size and the weight of the converter are reduced.
2.5 modulation method analysis
Power switch tube S1And a power switch tube S2、S3、S4And S5All have a certain logic relation, and in the positive half period, the power switch tube S4Working state of and power switch tube S1In unison, and during the negative half-cycle, the power switch tube S2Working state of and power switch tube S1In line, the power switch tube S3And S5Then is in power frequency state, and is connected with power switch tube S1The SPWM wave period of (a) corresponds to: the modulation wave is the absolute value of sine wave, and is compared with triangular wave carrier to obtain SPWM wave as power switch tube S1The switching signal of (1); power switch tube S4The positive half-cycle switching signal and the power switch tube S1Is consistent with the switching signal of the power switch tube S4The negative half-cycle switching signal of (2) is in a normally-off state; power switch tube S2Is turning toThe half-cycle switching signal is in a normally-off state, and the power switch tube S2Negative half-cycle switching signal and power switch tube S1The switching signals of (1) are consistent; power switch tube S3And S5The switching signal of (A) is a power frequency signal, and in the positive half period, the power switch tube S3The switching signal is in the normal on state, the power switch tube S5The switching signal is in a normally-off state, and in a negative half period, the power switch tube S5The switching signal is in the normal on state, the power switch tube S3The switching signal is in a normally off state. By the control mode, the bus voltage does not need to be stabilized, namely the ripple fluctuation of the direct current bus voltage on the direct current side does not influence the output voltage on the alternating current side, and the control mode is simplified. Most of the existing boost inverters need to feed back the voltage of the direct current bus in a control method, so that the control difficulty is increased, the number of required components is increased, the size and the weight are increased, the failure rate is higher, the stability of the whole system is poor, and the popularization is limited.
Example 3
The converter of this embodiment, which has substantially the same effect as that achieved in embodiment 1, includes a power switch S as shown in fig. 201、S2、S3、S4、S5And S6Inductance LinAnd a capacitor C1(ii) a One end of the DC side and the inductor LinIs connected to an inductor LinThe other end of the power switch tube S1Terminal 1 and power switch tube S6Terminal 1 is connected to power switch tube S6The terminals 2 are respectively connected with a power switch tube S2And S4Terminal 1, and capacitor C1One terminal of (C), a capacitor1The other ends of the two are respectively connected with a power switch tube S3And S5A terminal 2; the other end of the DC side, a power switch tube S1And S2Terminal 2, power switch tube S3The terminal 1, and one end of the AC side are grounded; power switch tube S5Terminal 1 and power switch tube S4The terminal 2 is connected with the other end of the alternating current side; wherein, the power switch tube S2、S3、S4And S5Are connected in anti-parallel with twoA pole tube; further, the capacitance C1The capacitance value of (A) is smaller and is a non-polar capacitor or a polar capacitor.
The effect achieved by embodiment 2 is basically the same, and the modulation method corresponding to this embodiment is as follows: within the whole power frequency period, the power switch tube S1Always working in SPWM state, power switch tube S6And a power switch tube S1The working state is opposite; in the positive half period of power frequency, the power switch tube S1And S4Synchronous, power switching tube S3In the on state, the power switch tube S2And S5In an off state; in the negative half period of the power frequency, the power switch tube S3And S4Are all in an off state, and a power switch tube S5In the on state, the power switch tube S2And S1The working states are consistent.
Example 4
A converter of this embodiment, as shown in FIG. 21, includes a power switch S1、S2、S3、S4、S5And S6Inductance LinAnd a capacitor C1(ii) a One end of the DC side and the inductor LinIs connected to an inductor LinThe other end of the power switch tube S1Terminal 1 and power switch tube S6Is connected with the terminal 2, the power switch tube S6The terminals 1 are respectively connected with a power switch tube S2And S4Terminal 1, electric capacity C1One terminal of (C), a capacitor1The other ends of the two are respectively connected with a power switch tube S3And S5A terminal 2; the other end of the DC side, a power switch tube S1And S2Terminal 2, power switch tube S3The terminal 1, and one end of the AC side are grounded; power switch tube S5Terminal 1 and power switch tube S4The terminal 2 is connected with the other end of the alternating current side; wherein, the power switch tube S3、S4、S5And S6Are connected in anti-parallel with the diode. Further, the capacitance C1The capacitance value of (a) is small, and the capacitance value is a non-polar capacitance or a polar capacitance, and the effect is basically the same as that realized by the embodiment 1.
The effect achieved by embodiment 2 is basically the same, and the modulation method corresponding to this embodiment is as follows: in the whole power frequency period, the power switch tube S1Always works in SPWM state, and the power switch tube S6Working in an off state all the time; in the positive half period of power frequency, the power switch tube S1And S4Synchronous, power switching tube S3In the on state, the power switch tube S2And S5In an off state; in the negative half period of the power frequency, the power switch tube S3And S4Are all in an off state, and a power switch tube S5In the on state, the power switch tube S2And S1The working states are consistent.
EXAMPLE 5 bidirectional converter and modulation method thereof
As shown in FIG. 22, a converter, a bidirectional converter, includes a power switch S1、S2、S3、S4、S5And S6Inductance LinAnd a capacitor C1(ii) a One end of the DC side and the inductor LinIs connected to an inductor LinThe other end of the power switch tube S1Terminal 1 and power switch tube S6Is connected with the terminal 2, the power switch tube S6The terminals 1 are respectively connected with a power switch tube S2And S4Terminal 1 and capacitor C1One terminal of (C), a capacitor1The other ends of the two are respectively connected with a power switch tube S3And S5A terminal 2; the other end of the DC side, a power switch tube S1And S2Terminal 2, power switch tube S3The terminal 1, and one end of the AC side are grounded; power switch tube S5Terminal 1 and power switch tube S4The terminal 2 is connected with the other end of the alternating current side;
wherein, the power switch tube S1、S2、S3、S4、S5And S6Both ends of the diode are connected with the diode in an anti-parallel mode, and the diode plays a role of follow current in the working process of the embodiment so as to realize energy flow and conversion between a direct current side and an alternating current side. The embodiment can realize the voltage boosting inversion, the voltage reducing inversion, the rectification voltage boosting and the rectification voltage reducingAnd (4) the action of pressure. When inverting, the converter has the function of reactive compensation.
Further, the capacitance C1Has a small capacitance value, and is a non-polar capacitor, or a polar capacitor, a capacitor C1During operation, the non-polar capacitor with a small capacitance value can be selected to be used only for energy transfer, so that the size and the weight of the converter are reduced.
As shown in FIG. 22, during the step-up/step-down inversion, the DC side is the DC power supply UinThe output voltage of the photovoltaic cell panel is used in an actual application scene; on the AC side, nodes E and F are connected in parallel to the input of a filter, the output of which is connected in parallel to a load or the grid (the converted electrical energy is fed directly back to the grid), wherein the filter is connected to a voltage UEFFiltering is performed to remove harmonic interference, the filtering may be selected according to an actual application scenario, and may be an LC filter (as shown in fig. 22), an LCL filter, or the like, an output of the filter may be connected to a power grid or a load, a load characteristic may also be selected according to an actual application scenario, and may be a resistive load, an inductive load, a capacitive load, or the like, and when the load is not a resistive load, reactive compensation may be performed on an output having a power factor smaller than 1.
When nodes E and F are connected in parallel with the LC filter, as shown in FIG. 22, node E is connected to filter inductor L0Is connected to node F and capacitor C0Is connected to one end of a filter inductor L0Another terminal of (1) and a filter capacitor C0Is connected with one end of a current source, a filter capacitor C0The other end of the current source and the other end of the current source are both grounded. The input side (DC side) and the output side (AC side) are grounded, so that common mode interference is avoided, and high-frequency leakage current does not exist.
This embodiment provides a modulation method corresponding to a bidirectional converter, in which a power switch tube S is used during a whole power frequency period during forward inversion operation1Always working in SPWM state, power switch tube S6Is always in the off state; in the positive half period of power frequency, the power switch tube S1And S4Synchronous, power switching tube S3In the on state, the power switch tube S2And S5In an off state; in the negative half period of the power frequency, the power switch tube S3And S4Are all in an off state, and a power switch tube S5In the on state, the power switch tube S2And S1The working states are consistent; during reverse rectification operation, the power switch tube S is in the whole power frequency period1Always working in SPWM state and power switch tube S3And S4And S1Synchronous, power switching tube S2、S5And S6And S1The state is reversed. Modulation method (power switch tube S) adopting the application1And S4After synchronization), the ripple fluctuation of the direct-current power supply on the direct-current side does not affect the alternating-current output voltage on the alternating-current side, and an additional control strategy does not need to be adopted for the ripple fluctuation of the direct-current power supply, so that the control method is simplified.
The foregoing description is only exemplary of the preferred embodiments of this application and is presented for the purpose of illustration and description of the principles of the technology used. The specification and description are not to be taken in a limiting sense, and the drawings are illustrative of only one embodiment of the invention and are not intended to limit the actual construction. It will be appreciated by those skilled in the art that the scope of the invention herein disclosed is not limited to the particular combination of features described above, but also encompasses other arrangements formed by any combination of the above features or their equivalents without departing from the spirit of the present application. For example, the above features may be replaced with (but not limited to) features having similar functions disclosed in the present application. Therefore, if the person skilled in the art receives the teaching, without departing from the spirit of the invention, the person skilled in the art shall not inventively design the similar structural modes and embodiments to the technical solution, but shall fall within the scope of the invention.
The invention and its embodiments have been described above schematically, without limitation, and the invention can be embodied in other specific forms without departing from the spirit or essential characteristics thereof. The representation in the drawings is only one of the embodiments of the invention, the actual construction is not limited thereto, and any reference signs in the claims shall not limit the claims concerned. Therefore, if the person skilled in the art receives the teaching, without departing from the spirit of the present invention, the person skilled in the art shall not inventively design the similar structural modes and embodiments to the technical solution, but shall fall within the protection scope of the present invention. Furthermore, the word "comprising" does not exclude other elements or steps, and the word "a" or "an" preceding an element does not exclude the presence of a plurality of such elements. Several of these means may be embodied by one and the same item of software or hardware. The terms first, second, etc. are used to denote names, but not any particular order.
Claims (7)
1. A transducer, characterized by: comprising a power switch tube S1、S2、S3、S4And S5Inductance LinAnd a capacitor C1And further comprising a switching device; the switching device is a diode D or a power switch tube S6The conducting end of the switch device is the anode of a diode D or a power switch tube S6Terminal 1, switching device cut-off terminal is diode D cathode or power switch tube S6A terminal 2; one end of the DC side of the converter and an inductor LinIs connected to an inductor LinThe other end of the power switch tube S1The terminal 1 is connected with the conducting end of the switch device, and the cut-off end of the switch device is respectively connected with the power switch tube S2And S4Terminal 1 and capacitor C1One terminal of (C), a capacitor1The other ends of the two are respectively connected with a power switch tube S3And S5The terminal 2 of (1); the other end of the DC side, a power switch tube S1And S2Terminal 2, power switch tube S3The terminal 1 and one end of the alternating current side are both grounded; power switch tube S5Terminal 1 and power switch tube S4The terminal 2 is connected with the other end of the alternating current side;
when the switching device is a diode D, the power switch tube comprises S1、S2、S3、S4And S5And a power switch tube S1、S2、S3、S4And S5Is a MOSFET device, an IGBT or a triode;when the power switch tube is an IGBT, the terminals 1, 3 and 2 of the power switch tube respectively represent the collector, the grid and the emitter of the IGBT, when the power switch tube is a triode, the terminals 1, 3 and 2 of the power switch tube respectively represent the collector, the base and the emitter of the triode, when the power switch tube is an MOSFET, the terminals 1, 3 and 2 of the power switch tube respectively represent the drain, the grid and the source of the MOSFET, and simultaneously, the power switch tube S2、S3、S4And S5And also are reversely connected with a diode and a power switch tube S in parallel2、S3、S4And S5The terminals 2 are all connected with the anode of the diode and the power switch tube S2、S3、S4And S5The terminals 1 of the diodes are all connected with the cathode of the diode;
the switching device being a power switching tube S6The power switch tube comprises S1、S2、S3、S4、S5And S6And a power switch tube S1、S2、S3、S4、S5And S6When the power switch tube is an IGBT, the terminals 1, 3 and 2 of the power switch tube respectively represent the collector, grid and emitter of the IGBT, when the power switch tube is an IGBT, the terminals 1, 3 and 2 of the power switch tube respectively represent the collector, base and emitter of the triode, when the power switch tube is an MOSFET, the terminals 1, 3 and 2 of the power switch tube respectively represent the drain, grid and source of the MOSFET, and simultaneously, the power switch tube S3And S5And also are reversely connected with a diode and a power switch tube S in parallel3And S5The terminals 2 are all connected with the anode of the diode and the power switch tube S3And S5Are connected to the cathode of the diode.
2. A transducer according to claim 1, characterized in that: capacitor C1The capacitor is a polar capacitor or a non-polar capacitor.
3. A transducer, characterized by: comprises a power switch tubeS1、S2、S3、S4And S5Inductance LinAnd a capacitor C1And further comprising a switching device; the switching device being a power switching tube S6The conducting end of the switch device is a power switch tube S6Terminal 2 of the switching device is a power switch tube S6The terminal 1 of (1); one end of the DC side of the converter and an inductor LinIs connected to an inductor LinThe other end of the power switch tube S1The terminal 1 is connected with the conducting end of the switch device, and the cut-off end of the switch device is respectively connected with the power switch tube S2And S4Terminal 1 and capacitor C1One terminal of (C), a capacitor1The other ends of the two are respectively connected with a power switch tube S3And S5The terminal 2 of (1); the other end of the DC side, a power switch tube S1And S2Terminal 2, power switch tube S3The terminal 1 and one end of the alternating current side are both grounded; power switch tube S5Terminal 1 and power switch tube S4The terminal 2 is connected with the other end of the alternating current side;
the power switch tube comprises S1、S2、S3、S4、S5And S6And a power switch tube S1、S2、S3、S4、S5And S6When the power switch tube is an IGBT, the terminals 1, 3 and 2 of the power switch tube respectively represent the collector, the grid and the emitter of the IGBT, when the power switch tube is an triode, the terminals 1, 3 and 2 of the power switch tube respectively represent the collector, the base and the emitter of the triode, when the power switch tube is an MOSFET, the terminals 1, 3 and 2 of the power switch tube respectively represent the drain, the grid and the source of the MOSFET, and the power switch tube S3、S4、S5And S6And also are reversely connected with a diode and a power switch tube S in parallel3、S4、S5And S6The terminals 2 are all connected with the anode of the diode and the power switch tube S3、S4、S5And S6Are connected to the cathode of the diode.
4. A transducer according to claim 3, wherein: power switch tube S1And S2And also are reversely connected with a diode and a power switch tube S in parallel1And S2The terminals 2 are all connected with the anode of the diode and the power switch tube S1And S2Are connected to the cathode of the diode.
5. A method of modulating a converter, characterized by: a converter as claimed in claim 1, wherein the power switch S is arranged to operate during the entire power frequency cycle1Working in an SPWM state all the time; in the positive half period of power frequency, the power switch tube S1And S4Synchronous, power switching tube S3In the on state, the power switch tube S2And S5In an off state; in the negative half period of the power frequency, the power switch tube S3And S4Are all in an off state, and a power switch tube S5In the on state, the power switch tube S2And S1The working states are consistent; when the switching device is a power switch tube S6In the whole power frequency period, the power switch tube S6And a power switch tube S1The working state is opposite.
6. A method of modulating a converter, characterized by: a converter as claimed in claim 3, wherein the power switch S is arranged to operate during the entire power frequency cycle1Always working in SPWM state, power switch tube S6Working in an off state all the time; in the positive half period of power frequency, the power switch tube S1And S4Synchronous, power switching tube S3In the on state, the power switch tube S2And S5In an off state; in the negative half period of the power frequency, the power switch tube S3And S4Are all in an off state, and a power switch tube S5In the on state, the power switch tube S2And S1The working states are consistent.
7. Method for modulating converter, and converterIs characterized in that: a converter as claimed in claim 4, wherein the power switch S is operated in forward-to-reverse operation for the entire power frequency cycle1Always working in SPWM state, power switch tube S6Is always in the off state; in the positive half period of power frequency, the power switch tube S1And S4Synchronous, power switching tube S3In the on state, the power switch tube S2And S5In an off state; in the negative half period of the power frequency, the power switch tube S3And S4Are all in an off state, and a power switch tube S5In the on state, the power switch tube S2And S1The working states are consistent; during reverse rectification operation, the power switch tube S is in the whole power frequency period1Always working in SPWM state and power switch tube S3And S4And S1Synchronous, power switching tube S2、S5And S6And S1The state is reversed.
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| CN111446874A (en) * | 2020-05-06 | 2020-07-24 | 西南交通大学 | Single-phase boost common-mode inverter and modulation method thereof |
| CN112003474B (en) * | 2020-09-04 | 2023-12-05 | 国网福建省电力有限公司电力科学研究院 | Cascade Buck-Boost high-gain converter |
| CN114696611A (en) * | 2020-12-28 | 2022-07-01 | 圣邦微电子（北京）股份有限公司 | Power converter and control method thereof |
| CN112737293B (en) * | 2021-02-02 | 2022-03-29 | 安徽工业大学 | Control method of non-isolated integrated boost DC/AC converter |
| CN113489363B (en) * | 2021-07-07 | 2022-05-24 | 国网湖北省电力有限公司电力科学研究院 | Bidirectional H6 photovoltaic grid-connected converter and modulation method thereof |
| CN113726199B (en) * | 2021-09-03 | 2023-09-22 | 安徽工业大学 | Low-output ripple boost rectifier and control method thereof |
| CN116404864B (en) * | 2023-06-07 | 2023-08-08 | 西南交通大学 | Power decoupling step-up and step-down common-ground power factor correction method and topological structure |
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