WO2023233446A1 - Filter circuit and power conversion device - Google Patents

Filter circuit and power conversion device Download PDF

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Publication number
WO2023233446A1
WO2023233446A1 PCT/JP2022/021877 JP2022021877W WO2023233446A1 WO 2023233446 A1 WO2023233446 A1 WO 2023233446A1 JP 2022021877 W JP2022021877 W JP 2022021877W WO 2023233446 A1 WO2023233446 A1 WO 2023233446A1
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WIPO (PCT)
Prior art keywords
winding
filter circuit
power
conversion device
power conversion
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PCT/JP2022/021877
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French (fr)
Japanese (ja)
Inventor
哲 村上
光 中川
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三菱電機株式会社
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Application filed by 三菱電機株式会社 filed Critical 三菱電機株式会社
Priority to PCT/JP2022/021877 priority Critical patent/WO2023233446A1/en
Publication of WO2023233446A1 publication Critical patent/WO2023233446A1/en

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
    • H03H7/01—Frequency selective two-port networks
    • H03H7/09—Filters comprising mutual inductance

Definitions

  • This application relates to a filter circuit and a power converter.
  • a power conversion device that is installed in electrical equipment such as home appliances and automobile-related equipment and performs power control includes a power conversion circuit that performs power conversion.
  • a power conversion circuit performs voltage conversion and the like by switching semiconductor elements, but noise is generated due to this switching.
  • a power conversion device including the following filter circuit has been disclosed.
  • a filter circuit in a conventional power conversion device includes a magnetic core in which one through hole is formed, one end is connected to the power conversion circuit, and the filter circuit is connected to the power conversion circuit from one first opening of the through hole to the second opening of the other through hole.
  • a first wiring that penetrates through the through hole and has its other end drawn out from the second opening, and a first wiring that has one end connected to the other end of the first wiring that penetrates from the second opening to the first opening of the through hole and has the other end connected to the other end of the first wiring. is a second wiring drawn out from the first opening as a filter output end, a first capacitor provided between the connection part of the first wiring and the second wiring, and the ground, and the other end of the second wiring. and a second capacitor provided between the ground and the ground (for example, see Patent Document 1).
  • the first wiring, the second wiring, and the first capacitor are made of magnetic material so that the magnetic flux due to the first wiring and the magnetic flux due to the second wiring due to DC components cancel each other out and become zero. Wired to the core.
  • the composite magnetic flux of the magnetic flux due to the first wiring and the magnetic flux due to the second wiring does not become zero, and the first wiring and the second wiring function as an inductor and a noise filter for the AC component. .
  • This realizes a compact filter circuit that suppresses the occurrence of magnetic saturation in the magnetic core due to an increase in magnetic flux density due to an increase in the output current of the power converter.
  • the present application discloses a technique for solving the above-mentioned problems, and aims to obtain a filter circuit with improved noise attenuation characteristics and a power conversion device equipped with this filter circuit.
  • the filter circuit disclosed in this application includes: a first capacitive element having one end connected to a transmission path for transmitting DC power and the other end connected to a reference potential; and a second capacitive element having one end connected to the transmission path and the other end connected to the reference potential.
  • a series body of a winding and a second capacitive element, The second winding is provided to be magnetically coupled to the transmission line, It is something.
  • the power converter device disclosed in this application includes: comprising a first DC circuit that outputs DC power and a second DC circuit that inputs DC power, The first DC circuit and the second DC circuit are connected to each other by a positive DC bus and a negative DC bus as the transmission line, A filter circuit configured as described above is provided on each of the positive DC bus and the negative DC bus, It is something.
  • a filter circuit and power conversion device with improved noise attenuation characteristics can be obtained.
  • FIG. 1 is a circuit diagram showing a schematic configuration of a power conversion device according to Embodiment 1.
  • FIG. FIG. 2 is a perspective view showing the structure of a filter circuit of the power conversion device according to the first embodiment.
  • FIG. 2 is a circuit diagram showing a schematic configuration of a power conversion device according to a comparative example.
  • 3 is a diagram showing noise attenuation characteristics of the power conversion device according to the first embodiment.
  • FIG. FIG. 3 is a diagram showing current-voltage phase characteristics of the power conversion device according to the first embodiment.
  • FIG. 3 is a diagram showing current-voltage phase characteristics of the power conversion device according to the first embodiment.
  • FIG. 2 is a circuit diagram showing a schematic configuration of a power conversion device according to a comparative example.
  • FIG. 3 is a diagram showing noise attenuation characteristics of the power conversion device according to the first embodiment.
  • FIG. FIG. 2 is a perspective view showing the structure of a filter circuit of the power conversion device according to the first embodiment.
  • FIG. 2 is a circuit diagram showing a schematic configuration of a power conversion device according to a second embodiment.
  • 7 is a diagram showing noise attenuation characteristics of a power conversion device according to a second embodiment.
  • FIG. 7 is a diagram showing current-voltage phase characteristics of a power converter according to a second embodiment.
  • FIG. 7 is a diagram showing current-voltage phase characteristics of a power converter according to a second embodiment.
  • FIG. 3 is a circuit diagram showing a schematic configuration of a power conversion device according to a third embodiment.
  • FIG. 1 is a circuit diagram showing a schematic configuration of a power conversion device 100 according to a first embodiment.
  • the power converter 100 of this embodiment includes a power converter 1 as a first DC circuit that outputs DC power, a load 2 such as a battery as a second DC circuit into which DC power is input, and the power converter 1 as a first DC circuit that outputs DC power.
  • the filter circuit 10 is provided on a power line B serving as a transmission line connecting the device 1 and the load 2. The filter circuit 10 is thus provided between the power converter 1 and the load 2, and attenuates the noise component contained in the DC power supplied from the power converter 1 to the load 2.
  • the filter circuit 10 includes a first inductor 3A in the power line B, a capacitor 3B as a first capacitive element whose one end is connected to the power line B and the other end is connected to the ground as a reference potential, and a second capacitor 3B. It includes a series circuit 3S as a series body in which a winding 3C as a wire and a capacitor 3D as a second capacitive element are connected in series.
  • the series circuit 3S has one end connected between the output side of the first induction section 3A and one end of the capacitor 3B, and the other end connected to ground.
  • the circuit symbol of the winding connected in series to the power line B is used as the first induction section 3A, but as will be explained below using FIG.
  • the guide portion 3A is not limited to a spiral winding.
  • FIG. 2 is a perspective view showing the structure of filter circuit 10 of power conversion device 100 according to this embodiment.
  • the left side in this figure is the power converter 1 side, and the right side is the load 2 side.
  • the filter circuit 10 has a core 4 as a magnetic material.
  • the first guide portion 3A is a portion of the power line B having a set length L.
  • the core 4 has a through hole 4H, into which the power line B is inserted, and is provided so as to surround the outer peripheral surface of the first guide portion 3A of the power line B.
  • the winding 3C has a cross-sectional area smaller than that of the power line B through which the main power from the power converter 1 is transferred, and is wound around the core 4. In this way, the winding 3C and the first induction section 3A are magnetically coupled to each other via the common core 4.
  • the self-inductance of the first induction section 3A is L1
  • the noise current flowing into the filter circuit 10 is i1
  • the voltage across the first induction section 3A caused by this noise current is V1
  • the self-inductance of the winding 3C is L2
  • the winding 3C Let i2 be the noise current flowing through the winding 3C, V2 be the voltage across the winding 3C, and M be the mutual inductance between the first induction section 3A and the winding 3C.
  • the voltage V1 across the first induction portion 3A is expressed by the following (Formula 1)
  • the voltage V2 across the winding 3C is expressed by the following (Formula 2).
  • the magnetic coupling relationship between the first induction section 3A and the winding 3C is such that when the capacitor 3D is short-circuited and a DC current is caused to flow through the first induction section 3A and the winding 3C, a direct current is generated in the core 4 due to the respective currents.
  • Winding the wires so that the magnetic fluxes strengthen each other is called a forward coupling configuration (K, M are positive), and winding the wires so that the magnetic fluxes weaken each other is called a reverse coupling configuration (K and M are negative).
  • 1 has a sequential connection configuration. That is, in the configuration of the filter circuit 10 shown in FIG. 2, the winding direction of the winding 3C with respect to the core 4 is adjusted so that the magnetic coupling with the first induction part 3A of the power line B is forward coupling. It has been wound.
  • the input voltage of the noise component generated by the noise current superimposed on the DC current output from the power converter 1 and flowing into the filter circuit 10 is Vn_in
  • the output voltage of the noise component output from the filter circuit 10 is Vn_out. do.
  • the input/output voltage ratio of noise indicating the attenuation effect of the filter is expressed by the following (Formula 4) using (Formula 1).
  • the first induction section 3A has a voltage drop due to the noise current i2 flowing through the winding 3C and the mutual inductance M due to magnetic coupling with the winding 3C.
  • the effects are added. Therefore, compared to the case where the magnetic bodies of the first induction section 3A and the winding 3C are configured separately, that is, when they are not magnetically coupled, the input/output voltage ratio of noise becomes larger. The noise attenuation effect can be enhanced.
  • the series circuit 3S of the winding 3C and the capacitor 3D resonates in series at a specific frequency, and the series impedance is ideally zero at the time of resonance. filter) can be configured.
  • the characteristics of the BEF are shown in the second term of (Equation 2), and the winding 3C has a noise current i1 flowing through the first induction section 3A due to magnetic coupling with the first induction section 3A. The effect of voltage drop due to mutual inductance M is added.
  • the filter circuit 10 of the present embodiment the BEF by the series circuit 3S of the winding 3C and the capacitor 3D, and the LPF (Low-pass filter) of the first induction part 3A and the capacitor 3B are integrated. It has the following configuration.
  • the power converter 100EX1 of the comparative example shown in FIG. 3 is obtained by removing the winding 3C and the capacitor 3D from the configuration of the filter circuit 10 of the power converter 100 of the present embodiment shown in FIG.
  • a filter circuit 10EX1 having only an LPF constituted by an inductor 3AEX1 and a capacitor 3BEX1 is provided.
  • FIG. 4 is a diagram for showing a specific effect of the amount of noise attenuation, and shows a comparison of the frequency characteristics of the power conversion device 100 of the present embodiment and the power conversion device 100EX1 of the comparative example.
  • the horizontal axis represents the frequency and the vertical axis represents the input/output voltage ratio on a logarithmic scale.
  • the filter circuit 10 of the power converter 100 of the present embodiment the first inductor 3A, the winding 3C are 2uH, the capacitor 3B is 2uF, the capacitor 3D is 0.4uF, and the coupling coefficient K
  • the frequency characteristic W1 is shown when is set to +0.5.
  • the filter circuit 10EX1 of the power conversion device 100EX1 of the comparative example the frequency characteristic W2 is shown when the first induction section 3A is set to 2uH and the capacitor 3B is set to 2uF.
  • a resonance point fres1 is generated by the first induction section 3A and the capacitor 3B, and the frequency is higher than this resonance point fres1. It has a characteristic that the amount of noise attenuation becomes larger in the second-order LPF.
  • the filter circuit 10 of this embodiment having a configuration in which an LPF and a BEF are integrated, a resonance point fres1I occurs mainly due to the first induction section 3A and the capacitor 3B. Due to the influence of the mutual inductance M shown by (Equation 1), the resonance point fres1I has a value lower than the resonance point fres1 of the filter circuit 10EX1 of the comparative example. Therefore, the filter circuit 10 of the present embodiment achieves a noise attenuation effect from lower frequencies than the filter circuit 10EX1 of the comparative example. Further, since series resonance occurs between the winding 3C and the capacitor 3D, and a BEF is formed around this resonance frequency fres2, it is possible to attenuate noise at a specific frequency.
  • the filter circuit 10 of this embodiment has a smaller noise attenuation amount than the filter circuit 10EX1 of the comparative example. This is because, with the resonant frequency fres2 as the boundary, the current phase flowing through the winding 3C changes from a 90-degree delayed current to a 90-degree advanced current phase, and the current phases of the first induction section 3A and the winding 3C are reversed. This is because the terms including the mutual inductance M shown in equations 1) to 2 are negative. A characteristic change in the amount of noise attenuation of the filter circuit 10 around the resonance frequency fres2 due to the term including the mutual inductance M becoming negative will be explained in comparison with the filter circuit 10EX1 of a comparative example.
  • FIG. 5 is a diagram showing the current phase of the first induction section 3A and the winding 3C in a frequency range (100 kHz) lower than the resonance frequency fres2 in the power converter 100 of the present embodiment, and the power converter of the present embodiment.
  • 100 is a diagram showing the voltage amplitude of the first induction section 3A of the power converter 100 and the first induction section 3AEX1 of the power conversion device 100EX1 of the comparative example.
  • FIG. 6 is a diagram showing the current phase of the first induction section 3A and the winding 3C in a frequency range (600 kHz) higher than the resonance frequency fres2 in the power converter 100 of the present embodiment, and the power converter of the present embodiment.
  • 100 is a diagram showing the voltage amplitude of the first induction section 3A of the power converter 100 and the first induction section 3AEX1 of the power conversion device 100EX1 of the comparative example.
  • the current phase of the first induction section 3A and the winding 3C of the filter circuit 10 is in phase, but as shown in FIG.
  • the current phases of the first induction section 3A and the winding 3C are opposite, and the polarity of the term including the mutual inductance M is reversed with the series resonance frequency fres2 as the boundary.
  • Equation 1 the influence of this polarity reversal appears on the voltage amplitude of the first induction section 3A. That is, as shown in FIG. 5, it can be seen that at a frequency lower than fres2, the voltage V1 of the first induction section 3A is higher than the voltage V1EX1 of the first induction section 3AEX1 of the filter circuit 10EX1 of the comparative example. On the other hand, as shown in FIG. 6, at a frequency higher than fres2, the voltage V1 of the first induction section 3A is higher than the voltage V1EX1 of the first induction section 3AEX1 of the filter circuit 10EX1 of the comparative example due to the influence of current phase inversion. It can be seen that it has become smaller.
  • the filter circuit 10 has a characteristic that the amount of noise attenuation decreases at frequencies higher than fres2, and a characteristic that obtains a high amount of noise attenuation in a low frequency band around fres1, which is lower than fres2.
  • FIG. 7 is a diagram showing the configuration of a power conversion device 100EX2 including a filter circuit 10EX2 of a comparative example.
  • the first induction section 3AEX2 and the winding 3CEX2 are each wound around separate magnetic cores, and are not magnetically coupled to each other. That is, in the filter circuit 10EX2, an LPF composed of the first induction section 3AEX2 and the winding 3BEX2, and a BEF composed of the windings 3CEX2 and 3DEX2 constitute separate filters.
  • FIG. 8 is a diagram showing a comparison of the frequency characteristics of the power conversion device 100 of the present embodiment and the power conversion device 100EX2 of the comparative example, in order to show a specific effect of the amount of noise attenuation.
  • a frequency characteristic W1 of the power converter 100 of the present embodiment and a frequency characteristic W3 of the power converter 100EX2 of the comparative example are shown.
  • the resonant frequency fres1II mainly composed of the first induction section 3AEX2 and the capacitor 3BEX2, and the resonant frequency fres2I mainly composed of the winding 3CEX2 and the capacitor 3DEX2, are both the electric power of this embodiment. It is higher than the resonant frequencies fres1I and fres2 of the conversion device 100. This shows that the power conversion device 100 of this embodiment, which is integrally configured by combining an LPF and a BEF, can obtain a higher noise attenuation effect in the low frequency range than the power conversion device 100EX2 of the comparative example. ing.
  • the filter circuit 10 included in the power converter 100 of the present embodiment has the LPF effect obtained by the first induction section 3A and the capacitor 3B, and the BEF effect obtained by the winding 3C and the capacitor 3D. Furthermore, since the LPF and BEF are coupled by magnetic coupling, high noise is produced in the low frequency region centered on fres1, where it is generally difficult to obtain a noise attenuation effect, and in the specific frequency region centered on fres2. This provides a damping effect.
  • the power converter 1 is shown as the 1st DC circuit which the power converter 100 has, and the load 2 is shown as the 2nd DC circuit, it is not limited to this.
  • the first DC circuit included in the power converter 100 may be a DC voltage source that outputs DC power
  • the second DC circuit may be a power converter.
  • the power converter 1 may be any converter that inputs and outputs a DC voltage to the filter circuit 10, and may be either an AC/DC or DC/DC converter, and may have an insulated/non-insulated configuration.
  • the first guiding portion 3A is a straight portion of the power line B having a set length L, and is not spirally wound, but is not limited to this.
  • the first guide portion 3A may be a spirally wound coil.
  • the core 4 may be provided so as to penetrate through the center of the coil serving as the first induction section.
  • the first induction section 3A may be configured to be magnetically coupled to the winding 3C with a mutual inductance M.
  • winding 3C is provided so as to be magnetically coupled to the first induction portion 3A
  • a configuration may be adopted in which the common core 4 is not provided.
  • FIG. 9 is a perspective view showing another example of the structure of the filter circuit 10 of the power conversion device 100 according to the present embodiment.
  • the power line B includes a recess 3A1 that is recessed in a direction perpendicular to the longitudinal direction.
  • the core 4 is provided so as to surround the outer peripheral surface of the power line B within the recess 3A1. With such a configuration, the core 4 can be stably installed even on the linear power line B.
  • the filter circuit of this embodiment configured as described above is a first capacitive element having one end connected to a transmission path for transmitting DC power and the other end connected to a reference potential; and a second capacitive element having one end connected to the transmission path and the other end connected to the reference potential.
  • a series body of a winding and a second capacitive element, The second winding is provided to be magnetically coupled to the transmission line, It is something.
  • the first capacitive element is connected to the transmission path to configure an LPF. Furthermore, a BEF of a series body of a second winding and a second capacitive element connected to the transmission line is provided. The second winding is provided so as to be magnetically coupled to the transmission line.
  • the noise attenuation characteristics of the filter itself can be improved without the need to provide an extra power line for the purpose of canceling DC magnetic flux and functioning as a noise filter.
  • the second winding and the transmission path are magnetically coupled to each other via a common magnetic body forming a closed magnetic path surrounding the outer peripheral surface of the transmission path. It is something.
  • the cross-sectional area of the second winding is configured to be smaller than the cross-sectional area of the transmission line. It is something.
  • the second winding is a thin wire through which only the noise current flows without passing the main power.
  • the transmission line is thick due to the large current capacity and it is not possible to increase the inductance by winding the transmission line multiple times in a magnetic material, or when the DC bus voltage of the power converter is high and the capacitor is used. If a high withstand voltage is required, a large capacitor is installed. Such an increase in the capacitance of the capacitor greatly affects the increase in the volume of the entire filter circuit.
  • a second winding configured to be smaller than the cross-sectional area of the transmission line, which does not allow the main power to pass through but only the noise current, is provided and is magnetically coupled to the transmission line, so that the inductance is reduced by the noise current. Therefore, a high noise attenuation effect can be obtained from lower frequencies without increasing loss or increasing the size.
  • FIG. 10 is a circuit diagram showing a schematic configuration of a power conversion device 200 according to the second embodiment.
  • the magnetic coupling relationship between the first induction portion 3A and the winding 3C in the filter circuit 210 is an inverse coupling configuration.
  • FIG. 11 shows frequency characteristics W4 of power conversion device 200 of this embodiment, frequency characteristics W1 of power conversion device 100 of Embodiment 1, and frequency characteristics W2 of power conversion device 100EX1 of a comparative example having only an LPF configuration. It is a figure showing. In this figure, the same specifications used to obtain the frequency characteristic W1 of the power converter 100 of the first embodiment are used for the first induction parts 3A, 3C and the capacitor 3B of the power converter 200 of the present embodiment. A frequency characteristic W4 applied to 3D is shown.
  • the frequency is lower than that of fres2 compared to the filter circuit 10EX1 of the comparative example using only an LPF. While a high noise attenuation effect was obtained in a low frequency band, the noise attenuation effect was reduced in a frequency band higher than the resonance point of fres2.
  • the filter circuit 210 of the power conversion device 200 of the second embodiment has a higher noise attenuation effect in a frequency band higher than the resonance point of fres2, compared to the filter circuit 10EX1 having an LPF-only configuration and the filter circuit 10 having a forward coupling configuration. is obtained.
  • the first term is positive. Since the second term is negative, the magnitude of di2/dt of the first term and di1/dt of the second term do not change, but depending on the magnitude relationship, V2, which is the sum of the first and second terms, may change.
  • the polarity can be positive or negative. For example, at a frequency lower than fres2, the impedance of capacitor 3D is large and the proportion of i2 in i1 is small, so the positive value of the first term is small and the negative value of the second term is large, so V2 is negative. becomes.
  • FIG. 12 shows the current phase of the first induction section 3A and the winding 3C in the frequency range (100 kHz) lower than the resonance frequency fres2 in the power converter 200 of the present embodiment, and the current phase of the first induction part 3A and the winding 3C of the power converter 200 of the present embodiment.
  • 1 is a diagram showing the voltage amplitude of the first induction section 3A and the winding 3C, the noise output voltage Vn_out of the power converter 200 of the present embodiment, and the voltage amplitude of the winding 3C.
  • FIG. 13 shows the current phase of the first induction section 3A and the winding 3C in a frequency range (700 kHz) higher than the resonance frequency fres2 in the power conversion device 200 of this embodiment, and the current phase of the first induction section 3A and the winding 3C of the power conversion device 200 of this embodiment.
  • 1 is a diagram showing the voltage amplitude of the first induction section 3A and the winding 3C, the noise output voltage Vn_out of the power converter 200 of the present embodiment, and the voltage amplitude of the winding 3C.
  • the first inductive part 3A and the winding 3C have the same phase. Current flows. However, as shown in FIG. 13, at a frequency higher than fres2, the winding 3C has a voltage that is in the same phase as the first induction section 3A, and in this case, it is in reverse phase with respect to the noise output voltage Vn_out, canceling the voltage. work like that. Therefore, as shown in the frequency characteristic W4 of FIG. 11, a high noise attenuation effect can be obtained in a frequency region (700 kHz) higher than the resonance frequency fres2.
  • the second winding is wound with a winding direction adjusted relative to the magnetic body so that the magnetic coupling with the transmission line is forward coupling or reverse coupling. It is something.
  • the second winding has a configuration in which the winding direction with respect to the magnetic body is adjusted so that the magnetic coupling with the transmission line is forward coupling or reverse coupling, thereby producing a high noise attenuation effect. It becomes possible to switch the frequency domain between a frequency domain higher than the resonance frequency fres2 and a frequency domain lower than the resonance frequency fres2. Therefore, if the magnetic coupling between the second winding and the transmission line is reversely coupled, a high noise attenuation effect can be obtained in a high frequency range. In this way, it is possible to effectively reduce noise in accordance with the specifications, installation environment, etc. of the applied power converter 100.
  • FIG. 14 is a circuit diagram showing a schematic configuration of power conversion device 300 according to the third embodiment.
  • the damping effect on normal mode noise was shown by the configuration in which the first induction section 3A and the winding 3C were magnetically coupled, but when the first induction section 3A and the winding 3C were A similar noise attenuation effect can be obtained with a common mode filter having a forward coupling configuration, and an example of the configuration is shown in FIG.
  • the power converter 1 and the load 2 are connected to each other by a positive DC bus BP and a negative DC bus BN, and two filter circuits 10 are provided on each of the positive DC bus BP and the negative DC bus BN. It will be done.
  • the first induction portion 3A and the winding 3C in the filter circuit 10 connected to the positive DC bus BP are magnetically coupled via a common core 4P.
  • the first induction section 3A and the winding 3C in the filter circuit 10 connected to the negative DC bus BN are magnetically coupled via a common core 4N. Note that the core 4P and the core 4N are separate magnetic bodies.
  • FIG. 15 is a diagram showing a comparison of the frequency characteristics of the power conversion device 300 of the present embodiment and the power conversion device 100EX1 of the comparative example, in order to show a specific effect of the amount of noise attenuation.
  • the frequency characteristic W2 of the filter circuit 10EX1 shown in the first embodiment in which the winding 3C and the capacitor 3D are removed and only the LPF is configured, and the frequency characteristic W5 of the power conversion device 300 of the present embodiment are shown. It shows.
  • the same specifications used to obtain the frequency characteristic W1 of the power converter 100 of the first embodiment are applied to the first induction parts 3A, 3C and the capacitors 3B, 3D of the power converter 300 of the present embodiment. There is.
  • the power conversion device of this embodiment comprising a first DC circuit that outputs DC power and a second DC circuit that inputs DC power
  • the first DC circuit and the second DC circuit are connected to each other by a positive DC bus and a negative DC bus as the transmission line
  • the filter circuit shown in Embodiment 1 or Embodiment 2 is provided on each of the positive DC bus and the negative DC bus, It is something.
  • FIG. 16 is a circuit diagram showing a schematic configuration of a power conversion device 400 including a filter circuit 410 according to the fourth embodiment.
  • the configuration includes only one series circuit 3S.
  • a plurality of series circuits 3S are provided, each of which is connected to the power line B.
  • a BEF having a plurality of specific attenuation ranges can be configured using one core 4 while lowering the resonance point fres2 of the original LPF characteristics and obtaining a high amount of noise attenuation.
  • the self-inductance of the winding 3G is L3, the flowing current is i3, the voltage at both ends is V3, the mutual inductance between the first induction section 3A and the winding 3C is M12, the mutual inductance between the first induction section 3A and the winding 3G is If the inductance is M13, and the mutual inductance between the winding 3C and the winding 3G is M23, the voltages V1, V2, and V3 across the first inductor 3A, the winding 3C, and the winding 3G are as follows (Equation 5) ⁇ It is shown by (Formula 7).

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Abstract

A filter circuit (10) comprises: a first capacitive element (3B) connected at one end to a transmission line (B) through which DC power is transmitted and at the other end to a reference potential; and a serial body, of a second winding (3C) and a second capacitive element (3D), connected at one end to the transmission line (B) and at the other end to the reference potential. The second winding (3C) is provided so as to be magnetically coupled with the transmission line (B).

Description

フィルタ回路および電力変換装置Filter circuits and power converters
　本願は、フィルタ回路および電力変換装置に関するものである。 This application relates to a filter circuit and a power converter.
　家電、自動車関連機器等の電機機器に設けられて電力制御を行う電力変換装置は、電力変換を行う電力変換回路を備えている。電力変換回路は、半導体素子のスイッチングにより電圧変換等を行うが、このスイッチングに伴いノイズが発生する。このノイズを効果的に抑制するために、以下のようなフィルタ回路を備えた電力変換装置が開示されている。 A power conversion device that is installed in electrical equipment such as home appliances and automobile-related equipment and performs power control includes a power conversion circuit that performs power conversion. A power conversion circuit performs voltage conversion and the like by switching semiconductor elements, but noise is generated due to this switching. In order to effectively suppress this noise, a power conversion device including the following filter circuit has been disclosed.
　即ち、従来の電力変換装置におけるフィルタ回路は、一つの貫通孔が形成された磁性体コアと、一端が電力変換回路に接続され、貫通孔の一方の第１開口から他方の第２開口へと貫通して、他端が第２開口から引き出されている第１配線と、一端が第１配線の他端に接続され、貫通孔の第２開口から第１開口へと貫通して、他端がフィルタ出力端として第１開口から引き出されている第２配線と、第１配線と第２配線との接続部とグランドとの間に設けられた第１コンデンサと、第２配線の他端とグランドとの間に設けられた第２コンデンサと、を備える（例えば、特許文献１参照）。 That is, a filter circuit in a conventional power conversion device includes a magnetic core in which one through hole is formed, one end is connected to the power conversion circuit, and the filter circuit is connected to the power conversion circuit from one first opening of the through hole to the second opening of the other through hole. A first wiring that penetrates through the through hole and has its other end drawn out from the second opening, and a first wiring that has one end connected to the other end of the first wiring that penetrates from the second opening to the first opening of the through hole and has the other end connected to the other end of the first wiring. is a second wiring drawn out from the first opening as a filter output end, a first capacitor provided between the connection part of the first wiring and the second wiring, and the ground, and the other end of the second wiring. and a second capacitor provided between the ground and the ground (for example, see Patent Document 1).
特開２０１５－４１９５９号公報JP 2015-41959 Publication
　上記のような従来のフィルタ回路において、第１配線、第２配線および第１コンデンサは、直流成分による第１配線による磁束と第２配線による磁束とが互いに打ち消しあってゼロとなるように磁性体コアに対して配線されている。一方、交流成分に関しては、第１配線による磁束と第２配線による磁束との合成磁束がゼロはならず、第１配線および第２配線が交流成分に対してインダクタとして機能しノイズフィルタとして機能する。これにより、電力変換器の出力電流増加に伴う磁束密度の増加に対する磁性体コアの磁気飽和の発生を抑制する小型のフィルタ回路を実現している。
　しかしながら、このような従来フィルタ回路においては、磁気飽和を抑制することで小型化は可能となるものの、一般的なフィルタ回路と比較してそのフィルタの減衰特性そのものを向上させるものではなく、所望のノイズ減衰特性が得られない場合があるという課題があった。 In the conventional filter circuit as described above, the first wiring, the second wiring, and the first capacitor are made of magnetic material so that the magnetic flux due to the first wiring and the magnetic flux due to the second wiring due to DC components cancel each other out and become zero. Wired to the core. On the other hand, regarding the AC component, the composite magnetic flux of the magnetic flux due to the first wiring and the magnetic flux due to the second wiring does not become zero, and the first wiring and the second wiring function as an inductor and a noise filter for the AC component. . This realizes a compact filter circuit that suppresses the occurrence of magnetic saturation in the magnetic core due to an increase in magnetic flux density due to an increase in the output current of the power converter.
However, although it is possible to reduce the size of such conventional filter circuits by suppressing magnetic saturation, this does not improve the filter's attenuation characteristics itself compared to general filter circuits, and it does not improve the desired attenuation characteristics. There was a problem that noise attenuation characteristics could not be obtained in some cases.
　本願は、上記のような課題を解決するための技術を開示するものであり、ノイズ減衰特性を向上したフィルタ回路と、このフィルタ回路を備えた電力変換装置とを得ることを目的としている。 The present application discloses a technique for solving the above-mentioned problems, and aims to obtain a filter circuit with improved noise attenuation characteristics and a power conversion device equipped with this filter circuit.
　本願に開示されるフィルタ回路は、
直流電力を伝送する伝送路に一端が接続され、他端が基準電位に接続された第１容量素子と、前記伝送路に一端が接続され、他端が前記基準電位に接続された、第２巻線と第２容量素子との直列体と、を備え、
前記第２巻線は、前記伝送路と相互に磁気結合するように設けられる、
ものである。
　また、本願に開示される電力変換装置は、
直流電力を出力する第１直流回路と、直流電力が入力される第２直流回路とを備え、
前記第１直流回路と前記第２直流回路とが、前記伝送路としての正側直流母線と負側直流母線とにより互いに接続され、
上記のように構成されたフィルタ回路が、前記正側直流母線と前記負側直流母線のそれぞれに設けられる、
ものである。 The filter circuit disclosed in this application includes:
a first capacitive element having one end connected to a transmission path for transmitting DC power and the other end connected to a reference potential; and a second capacitive element having one end connected to the transmission path and the other end connected to the reference potential. A series body of a winding and a second capacitive element,
The second winding is provided to be magnetically coupled to the transmission line,
It is something.
Furthermore, the power converter device disclosed in this application includes:
comprising a first DC circuit that outputs DC power and a second DC circuit that inputs DC power,
The first DC circuit and the second DC circuit are connected to each other by a positive DC bus and a negative DC bus as the transmission line,
A filter circuit configured as described above is provided on each of the positive DC bus and the negative DC bus,
It is something.
　本願に開示されるフィルタ回路および電力変換装置によれば、ノイズ減衰特性を向上したフィルタ回路および電力変換装置が得られる。 According to the filter circuit and power conversion device disclosed in the present application, a filter circuit and power conversion device with improved noise attenuation characteristics can be obtained.
実施の形態１による電力変換装置の概略構成を示す回路図である。1 is a circuit diagram showing a schematic configuration of a power conversion device according to Embodiment 1. FIG. 実施の形態１による電力変換装置のフィルタ回路の構造を示す斜視図である。FIG. 2 is a perspective view showing the structure of a filter circuit of the power conversion device according to the first embodiment. 比較例による電力変換装置の概略構成を示す回路図である。FIG. 2 is a circuit diagram showing a schematic configuration of a power conversion device according to a comparative example. 実施の形態１による電力変換装置のノイズ減衰特性を示す図である。3 is a diagram showing noise attenuation characteristics of the power conversion device according to the first embodiment. FIG. 実施の形態１による電力変換装置の電流電圧位相の特性を示す図である。FIG. 3 is a diagram showing current-voltage phase characteristics of the power conversion device according to the first embodiment. 実施の形態１による電力変換装置の電流電圧位相の特性を示す図である。FIG. 3 is a diagram showing current-voltage phase characteristics of the power conversion device according to the first embodiment. 比較例による電力変換装置の概略構成を示す回路図である。FIG. 2 is a circuit diagram showing a schematic configuration of a power conversion device according to a comparative example. 実施の形態１による電力変換装置のノイズ減衰特性を示す図である。3 is a diagram showing noise attenuation characteristics of the power conversion device according to the first embodiment. FIG. 実施の形態１による電力変換装置のフィルタ回路の構造を示す斜視図である。FIG. 2 is a perspective view showing the structure of a filter circuit of the power conversion device according to the first embodiment. 実施の形態２による電力変換装置の概略構成を示す回路図である。FIG. 2 is a circuit diagram showing a schematic configuration of a power conversion device according to a second embodiment. 実施の形態２による電力変換装置のノイズ減衰特性を示す図である。7 is a diagram showing noise attenuation characteristics of a power conversion device according to a second embodiment. FIG. 実施の形態２による電力変換装置の電流電圧位相の特性を示す図である。7 is a diagram showing current-voltage phase characteristics of a power converter according to a second embodiment. FIG. 実施の形態２による電力変換装置の電流電圧位相の特性を示す図である。7 is a diagram showing current-voltage phase characteristics of a power converter according to a second embodiment. FIG. 実施の形態３による電力変換装置の概略構成を示す回路図である。3 is a circuit diagram showing a schematic configuration of a power conversion device according to a third embodiment. FIG. 実施の形態３による電力変換装置のノイズ減衰特性を示す図である。FIG. 7 is a diagram showing noise attenuation characteristics of the power conversion device according to Embodiment 3; 実施の形態４による電力変換装置の概略構成を示す回路図である。12 is a circuit diagram showing a schematic configuration of a power conversion device according to a fourth embodiment. FIG.
実施の形態１．
　図１は、実施の形態１による電力変換装置１００の概略構成を示す回路図である。
　本実施の形態の電力変換装置１００は、直流電力を出力する第１直流回路としての電力変換器１と、直流電力が入力される第２直流回路としてのバッテリ等の負荷２と、この電力変換器１と負荷２とを接続する伝送路としての電力ラインＢに設けられるフィルタ回路１０と、を備える。
　フィルタ回路１０は、このように電力変換器１と負荷２との間に設けられ、電力変換器１から負荷２に対して供給される直流電力に含まれるノイズ成分を減衰させる。 Embodiment 1.
FIG. 1 is a circuit diagram showing a schematic configuration of a power conversion device 100 according to a first embodiment.
The power converter 100 of this embodiment includes a power converter 1 as a first DC circuit that outputs DC power, a load 2 such as a battery as a second DC circuit into which DC power is input, and the power converter 1 as a first DC circuit that outputs DC power. The filter circuit 10 is provided on a power line B serving as a transmission line connecting the device 1 and the load 2.
The filter circuit 10 is thus provided between the power converter 1 and the load 2, and attenuates the noise component contained in the DC power supplied from the power converter 1 to the load 2.
　フィルタ回路１０は、電力ラインＢにおける第１誘導部３Ａと、一端が電力ラインＢに接続され、他端が基準電位としてのグランドに接続された第１容量素子としてのコンデンサ３Ｂと、第２巻線としての巻線３Ｃと第２容量素子としてのコンデンサ３Ｄとを直列接続した直列体としての直列回路３Ｓと、を備える。
　直列回路３Ｓは、その一端が第１誘導部３Ａの出力側とコンデンサ３Ｂの一端との間に接続され、その他端がグランドに接続される。 The filter circuit 10 includes a first inductor 3A in the power line B, a capacitor 3B as a first capacitive element whose one end is connected to the power line B and the other end is connected to the ground as a reference potential, and a second capacitor 3B. It includes a series circuit 3S as a series body in which a winding 3C as a wire and a capacitor 3D as a second capacitive element are connected in series.
The series circuit 3S has one end connected between the output side of the first induction section 3A and one end of the capacitor 3B, and the other end connected to ground.
　なお、図１に示す回路図では、第１誘導部３Ａとして、電力ラインＢに直列接続された巻線の回路記号を用いているが、以下に図２を用いて説明するように、第１誘導部３Ａは、螺旋形状の巻線に限定するものではない。 In the circuit diagram shown in FIG. 1, the circuit symbol of the winding connected in series to the power line B is used as the first induction section 3A, but as will be explained below using FIG. The guide portion 3A is not limited to a spiral winding.
　以下、上記フィルタ回路１０の構造を図を用いて説明する。
　図２は、本実施の形態による電力変換装置１００のフィルタ回路１０の構造を示す斜視図である。
　本図における左側が電力変換器１側であり、右側が負荷２側である。 Hereinafter, the structure of the filter circuit 10 will be explained using the drawings.
FIG. 2 is a perspective view showing the structure of filter circuit 10 of power conversion device 100 according to this embodiment.
The left side in this figure is the power converter 1 side, and the right side is the load 2 side.
　図２に示すように、フィルタ回路１０は、磁性体としてのコア４を有する。
　第１誘導部３Ａは、電力ラインＢにおける設定された長さＬを有する部分である。
　コア４は、貫通孔４Ｈを有しており、この貫通孔４Ｈに電力ラインＢが挿通され、電力ラインＢの第１誘導部３Ａの外周面を取り囲むように設けられる。
　巻線３Ｃは、その断面積が、電力変換器１からの主電力が移送される電力ラインＢの断面積よりも小さく構成されており、コア４に巻回される。こうして、巻線３Ｃと第１誘導部３Ａとは、共通のコア４を介して相互に磁気結合される。 As shown in FIG. 2, the filter circuit 10 has a core 4 as a magnetic material.
The first guide portion 3A is a portion of the power line B having a set length L.
The core 4 has a through hole 4H, into which the power line B is inserted, and is provided so as to surround the outer peripheral surface of the first guide portion 3A of the power line B.
The winding 3C has a cross-sectional area smaller than that of the power line B through which the main power from the power converter 1 is transferred, and is wound around the core 4. In this way, the winding 3C and the first induction section 3A are magnetically coupled to each other via the common core 4.
　以下、本実施の形態の電力変換装置１００が有するフィルタ回路１０によるノイズ減衰効果について説明する。
　第１誘導部３Ａの自己インダクタンスをＬ１、フィルタ回路１０に流入するノイズ電流をｉ１、このノイズ電流により生じる第１誘導部３Ａにおける両端電圧をＶ１、巻線３Ｃの自己インダクタンスをＬ２、巻線３Ｃに流れるノイズ電流をｉ２、巻線３Ｃにおける両端電圧をＶ２とし、第１誘導部３Ａと巻線３Ｃとの間の相互インダクタンスをＭとする。
　この場合、第１誘導部３Ａの両端電圧Ｖ１は、以下の（式１）により示され、巻線３Ｃの両端電圧Ｖ２は、以下（式２）により示される。 Hereinafter, the noise attenuation effect by the filter circuit 10 included in the power conversion device 100 of this embodiment will be explained.
The self-inductance of the first induction section 3A is L1, the noise current flowing into the filter circuit 10 is i1, the voltage across the first induction section 3A caused by this noise current is V1, the self-inductance of the winding 3C is L2, the winding 3C Let i2 be the noise current flowing through the winding 3C, V2 be the voltage across the winding 3C, and M be the mutual inductance between the first induction section 3A and the winding 3C.
In this case, the voltage V1 across the first induction portion 3A is expressed by the following (Formula 1), and the voltage V2 across the winding 3C is expressed by the following (Formula 2).
Figure JPOXMLDOC01-appb-M000001
Figure JPOXMLDOC01-appb-M000001
Figure JPOXMLDOC01-appb-M000002
Figure JPOXMLDOC01-appb-M000002
　また、第１誘導部３Ａ、巻線３Ｃ間の結合係数をＫ（―１＜Ｋ＜＋１）とすると、相互インダクタンスＭは、以下（式３）により示される。 Further, assuming that the coupling coefficient between the first inductive portion 3A and the winding 3C is K (−1<K<+1), the mutual inductance M is expressed by the following (Equation 3).
Figure JPOXMLDOC01-appb-M000003
Figure JPOXMLDOC01-appb-M000003
　第１誘導部３Ａと巻線３Ｃとの磁気結合の関係は、コンデンサ３Ｄをショートし、直流電流を第１誘導部３Ａ、巻線３Ｃに流した時、それぞれの電流により生じるコア４内の直流磁束が強め合うように巻線されることを順結合（Ｋ、Ｍが正）構成、弱めあうように巻線されることを逆結合（Ｋ、Ｍが負）構成と称し、本実施の形態１では順結合構成とする。
　即ち、図２に示されるフィルタ回路１０の構成において、巻線３Ｃは、電力ラインＢの第１誘導部３Ａとの間の磁気結合が順結合となるように、コア４に対する巻回方向が調整されて巻回されている。 The magnetic coupling relationship between the first induction section 3A and the winding 3C is such that when the capacitor 3D is short-circuited and a DC current is caused to flow through the first induction section 3A and the winding 3C, a direct current is generated in the core 4 due to the respective currents. Winding the wires so that the magnetic fluxes strengthen each other is called a forward coupling configuration (K, M are positive), and winding the wires so that the magnetic fluxes weaken each other is called a reverse coupling configuration (K and M are negative). 1 has a sequential connection configuration.
That is, in the configuration of the filter circuit 10 shown in FIG. 2, the winding direction of the winding 3C with respect to the core 4 is adjusted so that the magnetic coupling with the first induction part 3A of the power line B is forward coupling. It has been wound.
　ここで、電力変換器１から出力される直流電流に重畳され、フィルタ回路１０に流入するノイズ電流により生じるノイズ成分の入力電圧をＶｎ＿ｉｎ、フィルタ回路１０から出力されるノイズ成分の出力電圧をＶｎ＿ｏｕｔとする。
　フィルタの減衰効果を示すノイズの入出力電圧比は、（式１）を用いると、以下の（式４）により示される。 Here, the input voltage of the noise component generated by the noise current superimposed on the DC current output from the power converter 1 and flowing into the filter circuit 10 is Vn_in, and the output voltage of the noise component output from the filter circuit 10 is Vn_out. do.
The input/output voltage ratio of noise indicating the attenuation effect of the filter is expressed by the following (Formula 4) using (Formula 1).
Figure JPOXMLDOC01-appb-M000004
Figure JPOXMLDOC01-appb-M000004
　上記（式４）の括弧内における第２項に示す通り、第１誘導部３Ａには、巻線３Ｃとの磁気結合により、巻線３Ｃに流れたノイズ電流ｉ２と相互インダクタンスＭによる電圧降下の影響が加算されている。そのため、第１誘導部３Ａと巻線３Ｃの磁性体がそれぞれ別体に構成されている場合、即ち、磁気結合されていない場合に比較して、ノイズの入出力電圧比が大きくなることから、ノイズ減衰効果を高めることができる。 As shown in the second term in the parentheses of the above (Equation 4), the first induction section 3A has a voltage drop due to the noise current i2 flowing through the winding 3C and the mutual inductance M due to magnetic coupling with the winding 3C. The effects are added. Therefore, compared to the case where the magnetic bodies of the first induction section 3A and the winding 3C are configured separately, that is, when they are not magnetically coupled, the input/output voltage ratio of noise becomes larger. The noise attenuation effect can be enhanced.
　また、巻線３Ｃとコンデンサ３Ｄとの直列回路３Ｓは、特定周波数で直列共振し、共振時においてその直列インピーダンスは理想的にゼロとなることから、特定周波数帯域における信号を減衰させるＢＥＦ（Ｂａｎｄ　ｅｌｉｍｉｎａｔｉｏｎ　ｆｉｌｔｅｒ）を構成できる。
　そのＢＥＦの特性は、（式２）の第２項に示されており、巻線３Ｃには、第１誘導部３Ａとの間の磁気結合により第１誘導部３Ａに流れたノイズ電流ｉ１と相互インダクタンスＭによる電圧降下の影響が加算される。 Further, the series circuit 3S of the winding 3C and the capacitor 3D resonates in series at a specific frequency, and the series impedance is ideally zero at the time of resonance. filter) can be configured.
The characteristics of the BEF are shown in the second term of (Equation 2), and the winding 3C has a noise current i1 flowing through the first induction section 3A due to magnetic coupling with the first induction section 3A. The effect of voltage drop due to mutual inductance M is added.
　このように第１誘導部３Ａと巻線３Ｃとが磁気結合している構成の場合、磁気結合していない構成の場合と比較して、第１誘導部３Ａ、巻線３Ｃの両端電圧Ｖ１、Ｖ２が大きくなることから、インダクタンスが増加したことと等価となる効果が得られる。
　このように、本実施の形態のフィルタ回路１０は、巻線３Ｃとコンデンサ３Ｄとの直列回路３ＳによるＢＥＦと、第１誘導部３Ａとコンデンサ３ＢとのＬＰＦ（Ｌｏｗ－ｐａｓｓ　ｆｉｌｔｅｒ）と、が統合された構成を有する。 In the case of the configuration in which the first induction part 3A and the winding 3C are magnetically coupled, the voltage V1 across the first induction part 3A and the winding 3C, Since V2 becomes larger, an effect equivalent to an increase in inductance can be obtained.
In this way, in the filter circuit 10 of the present embodiment, the BEF by the series circuit 3S of the winding 3C and the capacitor 3D, and the LPF (Low-pass filter) of the first induction part 3A and the capacitor 3B are integrated. It has the following configuration.
　以下、上記のように構成した本実施の形態のフィルタ回路１０のノイズ減衰効果について、比較例のフィルタ回路と比較して示す。
　図３に示す比較例の電力変換装置１００ＥＸ１は、図１に示した本実施の形態の電力変換装置１００のフィルタ回路１０の構成から、巻線３Ｃとコンデンサ３Ｄを取り除いたものであり、第１誘導部３ＡＥＸ１とコンデンサ３ＢＥＸ１とにより構成されるＬＰＦのみを有するフィルタ回路１０ＥＸ１を備える。 Hereinafter, the noise attenuation effect of the filter circuit 10 of the present embodiment configured as described above will be described in comparison with a filter circuit of a comparative example.
The power converter 100EX1 of the comparative example shown in FIG. 3 is obtained by removing the winding 3C and the capacitor 3D from the configuration of the filter circuit 10 of the power converter 100 of the present embodiment shown in FIG. A filter circuit 10EX1 having only an LPF constituted by an inductor 3AEX1 and a capacitor 3BEX1 is provided.
　図４は、ノイズ減衰量の具体的効果を示すための図であり、本実施の形態の電力変換装置１００と比較例の電力変換装置１００ＥＸ１の周波数特性を比較して示す。
　本図において、横軸を周波数、縦軸を入出力電圧比として対数スケールで表している。 FIG. 4 is a diagram for showing a specific effect of the amount of noise attenuation, and shows a comparison of the frequency characteristics of the power conversion device 100 of the present embodiment and the power conversion device 100EX1 of the comparative example.
In this figure, the horizontal axis represents the frequency and the vertical axis represents the input/output voltage ratio on a logarithmic scale.
　図４においては、一例として、本実施の形態の電力変換装置１００のフィルタ回路１０において、第１誘導部３Ａ、巻線３Ｃを２ｕＨ、コンデンサ３Ｂを２ｕＦ、コンデンサ３Ｄを０．４ｕＦ、結合係数Ｋを＋０．５とした場合の周波数特性Ｗ１を示している。
　また、比較例の電力変換装置１００ＥＸ１のフィルタ回路１０ＥＸ１において、第１誘導部３Ａを２ｕＨ、コンデンサ３Ｂを２ｕＦとした場合の周波数特性Ｗ２を示している。 In FIG. 4, as an example, in the filter circuit 10 of the power converter 100 of the present embodiment, the first inductor 3A, the winding 3C are 2uH, the capacitor 3B is 2uF, the capacitor 3D is 0.4uF, and the coupling coefficient K The frequency characteristic W1 is shown when is set to +0.5.
Further, in the filter circuit 10EX1 of the power conversion device 100EX1 of the comparative example, the frequency characteristic W2 is shown when the first induction section 3A is set to 2uH and the capacitor 3B is set to 2uF.
　周波数特性Ｗ２として示すように、ＬＰＦのみを有する比較例の電力変換装置１００ＥＸ１のフィルタ回路１０ＥＸ１においては、第１誘導部３Ａとコンデンサ３Ｂとにより共振点ｆｒｅｓ１が生じ、周波数がこの共振点ｆｒｅｓ１より高くなるにつれて２次のＬＰＦでノイズ減衰量が大きくなる特性を有する。 As shown as frequency characteristic W2, in the filter circuit 10EX1 of the power conversion device 100EX1 of the comparative example having only an LPF, a resonance point fres1 is generated by the first induction section 3A and the capacitor 3B, and the frequency is higher than this resonance point fres1. It has a characteristic that the amount of noise attenuation becomes larger in the second-order LPF.
　これに対し、周波数特性Ｗ１として示されるように、ＬＰＦとＢＥＦを統合した構成の本実施の形態のフィルタ回路１０においては、主として第１誘導部３Ａとコンデンサ３Ｂによる共振点ｆｒｅｓ１Ｉが生じる。そして（式１）により示した相互インダクタンスＭの影響により、共振点ｆｒｅｓ１Ｉは、比較例のフィルタ回路１０ＥＸ１の共振点ｆｒｅｓ１より低い値となっている。そのため、本実施の形態のフィルタ回路１０は、比較例のフィルタ回路１０ＥＸ１と比較して、より低い周波数からノイズ減衰効果が得られている。
　さらに、巻線３Ｃ、コンデンサ３Ｄとによる直列共振が生じ、この共振周波数ｆｒｅｓ２を中心にＢＥＦが形成されるため、特定周波数のノイズを減衰させることが可能となる。 On the other hand, as shown by the frequency characteristic W1, in the filter circuit 10 of this embodiment having a configuration in which an LPF and a BEF are integrated, a resonance point fres1I occurs mainly due to the first induction section 3A and the capacitor 3B. Due to the influence of the mutual inductance M shown by (Equation 1), the resonance point fres1I has a value lower than the resonance point fres1 of the filter circuit 10EX1 of the comparative example. Therefore, the filter circuit 10 of the present embodiment achieves a noise attenuation effect from lower frequencies than the filter circuit 10EX1 of the comparative example.
Further, since series resonance occurs between the winding 3C and the capacitor 3D, and a BEF is formed around this resonance frequency fres2, it is possible to attenuate noise at a specific frequency.
　なお、この巻線３Ｃ、コンデンサ３Ｄによる共振周波数ｆｒｅｓ２よりも高い周波数領域では、本実施の形態のフィルタ回路１０は、比較例のフィルタ回路１０ＥＸ１に比べてノイズ減衰量が小さくなっている。これは、共振周波数ｆｒｅｓ２を境に、巻線３Ｃに流れる電流位相が９０度遅れの電流から９０度進みの電流位相となり、第１誘導部３Ａと巻線３Ｃの電流位相が反転するため、（式１）～（式２）に示す相互インダクタンスＭを含む項がマイナスになるためである。
　この相互インダクタンスＭを含む項がマイナスになることによる、共振周波数ｆｒｅｓ２近辺を境にした、フィルタ回路１０のノイズ減衰量の特性変化について、比較例のフィルタ回路１０ＥＸ１と比較して説明する。 Note that in a frequency region higher than the resonance frequency fres2 caused by the winding 3C and the capacitor 3D, the filter circuit 10 of this embodiment has a smaller noise attenuation amount than the filter circuit 10EX1 of the comparative example. This is because, with the resonant frequency fres2 as the boundary, the current phase flowing through the winding 3C changes from a 90-degree delayed current to a 90-degree advanced current phase, and the current phases of the first induction section 3A and the winding 3C are reversed. This is because the terms including the mutual inductance M shown in equations 1) to 2 are negative.
A characteristic change in the amount of noise attenuation of the filter circuit 10 around the resonance frequency fres2 due to the term including the mutual inductance M becoming negative will be explained in comparison with the filter circuit 10EX1 of a comparative example.
　図５は、本実施の形態の電力変換装置１００における共振周波数ｆｒｅｓ２より低い周波数領域（１００ｋＨｚ）における第１誘導部３Ａ、巻線３Ｃの電流位相を示す図と、本実施の形態の電力変換装置１００の第１誘導部３Ａおよび比較例の電力変換装置１００ＥＸ１の第１誘導部３ＡＥＸ１の電圧振幅と、を示す図である。
　図６は、本実施の形態の電力変換装置１００における共振周波数ｆｒｅｓ２より高い周波数領域（６００ｋＨｚ）における第１誘導部３Ａ、巻線３Ｃの電流位相を示す図と、本実施の形態の電力変換装置１００の第１誘導部３Ａおよび比較例の電力変換装置１００ＥＸ１の第１誘導部３ＡＥＸ１の電圧振幅と、を示す図である。 FIG. 5 is a diagram showing the current phase of the first induction section 3A and the winding 3C in a frequency range (100 kHz) lower than the resonance frequency fres2 in the power converter 100 of the present embodiment, and the power converter of the present embodiment. 100 is a diagram showing the voltage amplitude of the first induction section 3A of the power converter 100 and the first induction section 3AEX1 of the power conversion device 100EX1 of the comparative example.
FIG. 6 is a diagram showing the current phase of the first induction section 3A and the winding 3C in a frequency range (600 kHz) higher than the resonance frequency fres2 in the power converter 100 of the present embodiment, and the power converter of the present embodiment. 100 is a diagram showing the voltage amplitude of the first induction section 3A of the power converter 100 and the first induction section 3AEX1 of the power conversion device 100EX1 of the comparative example.
　図５に示すように、ｆｒｅｓ２より低い周波数（１００ｋＨｚ）では、フィルタ回路１０の第１誘導部３Ａと巻線３Ｃの電流位相は同相だが、図６に示すように、ｆｒｅｓ２よりも高い周波数では第１誘導部３Ａと巻線３Ｃの電流位相は逆相となり、直列共振周波数ｆｒｅｓ２を境に相互インダクタンスＭを含む項の極性が反転する。 As shown in FIG. 5, at a frequency lower than fres2 (100 kHz), the current phase of the first induction section 3A and the winding 3C of the filter circuit 10 is in phase, but as shown in FIG. The current phases of the first induction section 3A and the winding 3C are opposite, and the polarity of the term including the mutual inductance M is reversed with the series resonance frequency fres2 as the boundary.
　（式１）に示した通り、この極性反転の影響が、第１誘導部３Ａの電圧振幅に現れる。即ち、図５に示すようにｆｒｅｓ２より低い周波数では、第１誘導部３Ａの電圧Ｖ１が、比較例のフィルタ回路１０ＥＸ１の第１誘導部３ＡＥＸ１の電圧Ｖ１ＥＸ１よりも高くなっていることが判る。
　これに対し、図６に示すようにｆｒｅｓ２より高い周波数では、電流位相反転の影響で、第１誘導部３Ａの電圧Ｖ１が、比較例のフィルタ回路１０ＥＸ１の第１誘導部３ＡＥＸ１の電圧Ｖ１ＥＸ１よりも小さくなっていることが判る。 As shown in (Equation 1), the influence of this polarity reversal appears on the voltage amplitude of the first induction section 3A. That is, as shown in FIG. 5, it can be seen that at a frequency lower than fres2, the voltage V1 of the first induction section 3A is higher than the voltage V1EX1 of the first induction section 3AEX1 of the filter circuit 10EX1 of the comparative example.
On the other hand, as shown in FIG. 6, at a frequency higher than fres2, the voltage V1 of the first induction section 3A is higher than the voltage V1EX1 of the first induction section 3AEX1 of the filter circuit 10EX1 of the comparative example due to the influence of current phase inversion. It can be seen that it has become smaller.
　こうして、フィルタ回路１０は、ｆｒｅｓ２より高い周波数ではノイズ減衰量が低下する特性を有し、ｆｒｅｓ２より低いｆｒｅｓ１付近の低周波数帯においては高いノイズ減衰量を得る特性を有する。 In this way, the filter circuit 10 has a characteristic that the amount of noise attenuation decreases at frequencies higher than fres2, and a characteristic that obtains a high amount of noise attenuation in a low frequency band around fres1, which is lower than fres2.
　次に、第１誘導部３Ａと巻線３Ｃとの磁気結合に伴うノイズ減衰効果について説明する。
　図７は、比較例のフィルタ回路１０ＥＸ２を備える電力変換装置１００ＥＸ２の構成を示す図である。
　フィルタ回路１０ＥＸ２は、第１誘導部３ＡＥＸ２と巻線３ＣＥＸ２とがそれぞれ別体の磁性体コアに巻線されており、相互に磁気結合していない。即ち、フィルタ回路１０ＥＸ２においては、第１誘導部３ＡＥＸ２と巻線３ＢＥＸ２とで構成されるＬＰＦと、巻線３ＣＥＸ２と３ＤＥＸ２とで構成されるＢＥＦとがそれぞれ別体のフィルタを構成している。 Next, the noise attenuation effect accompanying the magnetic coupling between the first induction section 3A and the winding 3C will be explained.
FIG. 7 is a diagram showing the configuration of a power conversion device 100EX2 including a filter circuit 10EX2 of a comparative example.
In the filter circuit 10EX2, the first induction section 3AEX2 and the winding 3CEX2 are each wound around separate magnetic cores, and are not magnetically coupled to each other. That is, in the filter circuit 10EX2, an LPF composed of the first induction section 3AEX2 and the winding 3BEX2, and a BEF composed of the windings 3CEX2 and 3DEX2 constitute separate filters.
　図８は、ノイズ減衰量の具体的効果を示すための、本実施の形態の電力変換装置１００と比較例の電力変換装置１００ＥＸ２の周波数特性を比較して示す図である。
　本図において、本実施の形態の電力変換装置１００の周波数特性Ｗ１と、比較例の電力変換装置１００ＥＸ２の周波数特性Ｗ３とを示している。 FIG. 8 is a diagram showing a comparison of the frequency characteristics of the power conversion device 100 of the present embodiment and the power conversion device 100EX2 of the comparative example, in order to show a specific effect of the amount of noise attenuation.
In this figure, a frequency characteristic W1 of the power converter 100 of the present embodiment and a frequency characteristic W3 of the power converter 100EX2 of the comparative example are shown.
　図８に示されるように、主として第１誘導部３ＡＥＸ２とコンデンサ３ＢＥＸ２とから構成される共振周波数ｆｒｅｓ１ＩＩ、巻線３ＣＥＸ２とコンデンサ３ＤＥＸ２とから構成される共振周波数ｆｒｅｓ２Ｉが、いずれも本実施の形態の電力変換装置１００の共振周波数ｆｒｅｓ１Ｉ、ｆｒｅｓ２よりも高くなっている。
　このことは、ＬＰＦとＢＥＦとを結合して一体構成した本実施の形態の電力変換装置１００が、比較例の電力変換装置１００ＥＸ２よりも、低周波数域において高いノイズ減衰効果が得られることを示している。 As shown in FIG. 8, the resonant frequency fres1II mainly composed of the first induction section 3AEX2 and the capacitor 3BEX2, and the resonant frequency fres2I mainly composed of the winding 3CEX2 and the capacitor 3DEX2, are both the electric power of this embodiment. It is higher than the resonant frequencies fres1I and fres2 of the conversion device 100.
This shows that the power conversion device 100 of this embodiment, which is integrally configured by combining an LPF and a BEF, can obtain a higher noise attenuation effect in the low frequency range than the power conversion device 100EX2 of the comparative example. ing.
　以上のように、本実施の形態の電力変換装置１００が有するフィルタ回路１０は、第１誘導部３Ａとコンデンサ３Ｂとにより得られるＬＰＦ効果と、巻線３Ｃとコンデンサ３ＤとによるＢＥＦ効果とを備え、更に、磁気結合によりＬＰＦとＢＥＦとを結合した構成としているため、一般的にノイズ減衰効果を得られにくいｆｒｅｓ１を中心とした低周波数領域と、ｆｒｅｓ２を中心とした特定周波数領域において、高いノイズ減衰効果を得るものである。 As described above, the filter circuit 10 included in the power converter 100 of the present embodiment has the LPF effect obtained by the first induction section 3A and the capacitor 3B, and the BEF effect obtained by the winding 3C and the capacitor 3D. Furthermore, since the LPF and BEF are coupled by magnetic coupling, high noise is produced in the low frequency region centered on fres1, where it is generally difficult to obtain a noise attenuation effect, and in the specific frequency region centered on fres2. This provides a damping effect.
　なお、電力変換装置１００が有する第１直流回路として電力変換器１を示し、第２直流回路として負荷２を示したがこれに限定するものではない。例えば、電力変換装置１００が有する第１直流回路は、直流電力を出力する直流電圧源でもよく、第２直流回路が電力変換器でもよい。
　なお、電力変換器１はフィルタ回路１０に直流電圧を入出力する変換器であればよく、ＡＣ／ＤＣ、ＤＣ／ＤＣ変換器のいずれでもよく、絶縁／非絶縁の構成についても問わない。 In addition, although the power converter 1 is shown as the 1st DC circuit which the power converter 100 has, and the load 2 is shown as the 2nd DC circuit, it is not limited to this. For example, the first DC circuit included in the power converter 100 may be a DC voltage source that outputs DC power, and the second DC circuit may be a power converter.
Note that the power converter 1 may be any converter that inputs and outputs a DC voltage to the filter circuit 10, and may be either an AC/DC or DC/DC converter, and may have an insulated/non-insulated configuration.
　また、上記では、第１誘導部３Ａは、電力ラインＢにおける設定された長さＬの直線部分とし、螺旋状に巻回されていない構成のものを示したが、これに限定するものではない。例えば第１誘導部３Ａは、螺旋状に巻回されたコイルでもよい。この場合、コア４を第１誘導部としてのコイルの中心を貫通するように設けてもよい。第１誘導部３Ａは、巻線３Ｃと相互インダクタンスＭを有して磁気結合される構成であればよい。 Further, in the above description, the first guiding portion 3A is a straight portion of the power line B having a set length L, and is not spirally wound, but is not limited to this. . For example, the first guide portion 3A may be a spirally wound coil. In this case, the core 4 may be provided so as to penetrate through the center of the coil serving as the first induction section. The first induction section 3A may be configured to be magnetically coupled to the winding 3C with a mutual inductance M.
　また、巻線３Ｃが第１誘導部３Ａに対して磁気結合可能に設けられるのであれば、共通のコア４を設けない構成としてもよい。 Further, as long as the winding 3C is provided so as to be magnetically coupled to the first induction portion 3A, a configuration may be adopted in which the common core 4 is not provided.
　なお、伝送路には、以下図９に示すように、凹部を設ける構成としてもよい。
　図９は、本実施の形態による電力変換装置１００のフィルタ回路１０の構造の別例を示す斜視図である。
　電力ラインＢは、長手方向に対して垂直な方向に窪む凹部３Ａ１を備えている。
　そして、コア４は、この凹部３Ａ１内において電力ラインＢの外周面を取り囲むように設けられる。このような構成とすることにより、直線状の電力ラインＢに対しても、安定的にコア４を設置させることができる。 Note that the transmission path may have a configuration in which a recess is provided as shown in FIG. 9 below.
FIG. 9 is a perspective view showing another example of the structure of the filter circuit 10 of the power conversion device 100 according to the present embodiment.
The power line B includes a recess 3A1 that is recessed in a direction perpendicular to the longitudinal direction.
The core 4 is provided so as to surround the outer peripheral surface of the power line B within the recess 3A1. With such a configuration, the core 4 can be stably installed even on the linear power line B.
　上記のように構成された本実施の形態のフィルタ回路は、
直流電力を伝送する伝送路に一端が接続され、他端が基準電位に接続された第１容量素子と、前記伝送路に一端が接続され、他端が前記基準電位に接続された、第２巻線と第２容量素子との直列体と、を備え、
前記第２巻線は、前記伝送路と相互に磁気結合するように設けられる、
ものである。 The filter circuit of this embodiment configured as described above is
a first capacitive element having one end connected to a transmission path for transmitting DC power and the other end connected to a reference potential; and a second capacitive element having one end connected to the transmission path and the other end connected to the reference potential. A series body of a winding and a second capacitive element,
The second winding is provided to be magnetically coupled to the transmission line,
It is something.
　このように、伝送路に対して第１容量素子が接続されてＬＰＦが構成されている。更に、伝送路に接続された第２巻線と第２容量素子との直列体のＢＥＦを備えている。そして、第２巻線は、伝送路と相互に磁気結合するように設けられる。
　これにより、一般的にノイズ減衰効果を得られにくいｆｒｅｓ１を中心とした低周波数領域と、ｆｒｅｓ２を中心とした特定周波数領域において、高いノイズ減衰効果を得ることができる。 In this way, the first capacitive element is connected to the transmission path to configure an LPF. Furthermore, a BEF of a series body of a second winding and a second capacitive element connected to the transmission line is provided. The second winding is provided so as to be magnetically coupled to the transmission line.
As a result, it is possible to obtain a high noise attenuation effect in a low frequency region centered on fres1, where it is generally difficult to obtain a noise attenuation effect, and in a specific frequency region centered on fres2.
　また、例えば、直流磁束をキャンセルしてノイズフィルタとして機能させることを目的とした電力線を電力ラインに余分に設ける必要なく、フィルタのノイズ減衰特性そのものを向上できる。 Further, for example, the noise attenuation characteristics of the filter itself can be improved without the need to provide an extra power line for the purpose of canceling DC magnetic flux and functioning as a noise filter.
　また、上記のように構成された本実施の形態のフィルタ回路においては、
前記第２巻線と前記伝送路とは、該伝送路の外周面を取り囲むように閉磁路を形成する共通の磁性体を介して相互に磁気結合される、
ものである。 Furthermore, in the filter circuit of this embodiment configured as described above,
The second winding and the transmission path are magnetically coupled to each other via a common magnetic body forming a closed magnetic path surrounding the outer peripheral surface of the transmission path.
It is something.
　このように、伝送路の外周面を取り囲むように閉磁路を形成する磁性体を設け、この磁性体を介して伝送路と第２巻線とが相互に磁気結合される構成とすることで、主電力が移送される伝送路をコイル状に磁性体に巻回する必要がなくなる。これにより、フィルタ回路の設置において、伝送路を切断する必要がなく、作業性が向上すると共に、伝送路が巻回される構成に比較して、伝送路における損失を低減することができる。
　更には、伝送路と第２巻線とは共通の磁性体を有するため、それぞれ別体の磁性体を備える構成に比較して、使用する磁性体の数を削減して低コスト化が図れると共に、省スペース化を図れる。 In this way, by providing a magnetic body that forms a closed magnetic path so as to surround the outer peripheral surface of the transmission line, and by configuring the transmission line and the second winding to be magnetically coupled to each other via this magnetic body, There is no need to wind the transmission path through which the main power is transferred around a magnetic material in a coil shape. This eliminates the need to cut the transmission line when installing the filter circuit, improving workability and reducing loss in the transmission line compared to a configuration in which the transmission line is wound.
Furthermore, since the transmission line and the second winding have a common magnetic material, compared to a configuration in which each has separate magnetic materials, the number of magnetic materials used can be reduced and costs can be reduced. , space saving can be achieved.
　また、上記のように構成された本実施の形態のフィルタ回路においては、
前記第２巻線の断面積は、前記伝送路の断面積よりも小さく構成される、
ものである。 Furthermore, in the filter circuit of this embodiment configured as described above,
The cross-sectional area of the second winding is configured to be smaller than the cross-sectional area of the transmission line.
It is something.
　このように、第２巻線の断面積を伝送路の断面積よりも小さく構成することで、第２巻線には、主電力が通過せずノイズ電流のみが流れる細線としている。
　一般的に、電流容量が大きいために伝送路が太く、伝送路を磁性体に複数回巻線してインダクタンスを増加することができない場合、あるいは、電力変換器の直流母線電圧が高電圧でコンデンサに耐圧が必要な場合では、大型のコンデンサが設けられる。このようなコンデンサの容量増加は、フィルタ回路全体の体積増加に大きく影響してしまう。
　このような場合でも、主電力が通過せずノイズ電流だけを通過させる、伝送路の断面積よりも小さく構成された第２巻線を設けて、伝送路と磁気結合させ、ノイズ電流によってインダクタンスを増加することができるため、損失増加、大型化することなく、より低い周波数から高いノイズ減衰効果を得ることができる。 In this way, by configuring the cross-sectional area of the second winding to be smaller than the cross-sectional area of the transmission line, the second winding is a thin wire through which only the noise current flows without passing the main power.
Generally, when the transmission line is thick due to the large current capacity and it is not possible to increase the inductance by winding the transmission line multiple times in a magnetic material, or when the DC bus voltage of the power converter is high and the capacitor is used. If a high withstand voltage is required, a large capacitor is installed. Such an increase in the capacitance of the capacitor greatly affects the increase in the volume of the entire filter circuit.
Even in such a case, a second winding configured to be smaller than the cross-sectional area of the transmission line, which does not allow the main power to pass through but only the noise current, is provided and is magnetically coupled to the transmission line, so that the inductance is reduced by the noise current. Therefore, a high noise attenuation effect can be obtained from lower frequencies without increasing loss or increasing the size.
実施の形態２．
　以下、本願の実施の形態２を、上記実施の形態１と異なる箇所を中心に図を用いて説明する。上記実施の形態１と同様の部分は同一符号を付して説明を省略する。
　図１０は、実施の形態２による電力変換装置２００の概略構成を示す回路図である。
　本実施の形態の電力変換装置２００は、フィルタ回路２１０における第１誘導部３Ａと巻線３Ｃとの磁気結合関係を逆結合構成としたものである。 Embodiment 2.
Embodiment 2 of the present application will be described below with reference to the drawings, focusing on the differences from Embodiment 1 described above. The same parts as in the first embodiment are given the same reference numerals, and the description thereof will be omitted.
FIG. 10 is a circuit diagram showing a schematic configuration of a power conversion device 200 according to the second embodiment.
In the power conversion device 200 of this embodiment, the magnetic coupling relationship between the first induction portion 3A and the winding 3C in the filter circuit 210 is an inverse coupling configuration.
　図１１は、本実施の形態の電力変換装置２００の周波数特性Ｗ４と、実施の形態１の電力変換装置１００の周波数特性Ｗ１と、ＬＰＦのみの構成の比較例の電力変換装置１００ＥＸ１の周波数特性Ｗ２と示す図である。
　本図においては、実施の形態１の電力変換装置１００の周波数特性Ｗ１を得るために用いた同じ諸元を、本実施の形態の電力変換装置２００の第１誘導部３Ａ、３Ｃとコンデンサ３Ｂ、３Ｄに適用した周波数特性Ｗ４を示している。 FIG. 11 shows frequency characteristics W4 of power conversion device 200 of this embodiment, frequency characteristics W1 of power conversion device 100 of Embodiment 1, and frequency characteristics W2 of power conversion device 100EX1 of a comparative example having only an LPF configuration. It is a figure showing.
In this figure, the same specifications used to obtain the frequency characteristic W1 of the power converter 100 of the first embodiment are used for the first induction parts 3A, 3C and the capacitor 3B of the power converter 200 of the present embodiment. A frequency characteristic W4 applied to 3D is shown.
　実施の形態１の電力変換装置１００においては、第１誘導部３Ａと巻線３Ｃとの磁気結合が順結合構成である場合は、ＬＰＦのみの比較例のフィルタ回路１０ＥＸ１と比較して、ｆｒｅｓ２より低い周波数帯においてノイズの高減衰効果が得られていた一方、ｆｒｅｓ２の共振点よりも高い周波数帯においてはノイズ減衰効果が低下していた。
　本実施の形態２の電力変換装置２００のフィルタ回路２１０では、ＬＰＦのみの構成のフィルタ回路１０ＥＸ１および順結合構成のフィルタ回路１０と比較して、ｆｒｅｓ２の共振点より高い周波数帯において高いノイズ減衰効果が得られる。 In the power conversion device 100 of the first embodiment, when the magnetic coupling between the first induction section 3A and the winding 3C is a forward coupling configuration, the frequency is lower than that of fres2 compared to the filter circuit 10EX1 of the comparative example using only an LPF. While a high noise attenuation effect was obtained in a low frequency band, the noise attenuation effect was reduced in a frequency band higher than the resonance point of fres2.
The filter circuit 210 of the power conversion device 200 of the second embodiment has a higher noise attenuation effect in a frequency band higher than the resonance point of fres2, compared to the filter circuit 10EX1 having an LPF-only configuration and the filter circuit 10 having a forward coupling configuration. is obtained.
　本実施の形態の電力変換装置２００における逆結合構成においても、実施の形態１に示した（式１）～（式３）は成立し、（式２）を見れば、第１項は正で第２項は負のため、第１項のｄｉ２／ｄｔと第２項のｄｉ１／ｄｔの大きさは変わらないが、その大小関係によっては、第１項と第２項の和であるＶ２の極性が正にもなり負にもなりえる。
　例えば、ｆｒｅｓ２よりも低い周波数においては、コンデンサ３Ｄのインピーダンスが大きくｉ１におけるｉ２の占める割合が小さいため、第１項の正の値が小さく第２項の負の値が大きくなるため、Ｖ２は負となる。 Even in the inversely coupled configuration of power conversion device 200 of this embodiment, (Formula 1) to (Formula 3) shown in Embodiment 1 hold true, and looking at (Formula 2), the first term is positive. Since the second term is negative, the magnitude of di2/dt of the first term and di1/dt of the second term do not change, but depending on the magnitude relationship, V2, which is the sum of the first and second terms, may change. The polarity can be positive or negative.
For example, at a frequency lower than fres2, the impedance of capacitor 3D is large and the proportion of i2 in i1 is small, so the positive value of the first term is small and the negative value of the second term is large, so V2 is negative. becomes.
　これに対し、ｆｒｅｓ２より周波数が高い場合、コンデンサ３Ｄのインピーダンスが小さくｉ１におけるｉ２の占める割合が大きくなるため、第２項の負の値より第１項の正の値が大きくなりＶ２が正となり、第１誘導部３Ａと巻線３Ｃの電圧極性が同相になる。 On the other hand, when the frequency is higher than fres2, the impedance of capacitor 3D is small and the proportion of i2 in i1 becomes large, so the positive value of the first term becomes larger than the negative value of the second term, and V2 becomes positive. , the voltage polarities of the first induction section 3A and the winding 3C are in phase.
　図１２は、本実施の形態の電力変換装置２００における共振周波数ｆｒｅｓ２より低い周波数領域（１００ｋＨｚ）における第１誘導部３Ａ、巻線３Ｃの電流位相と、本実施の形態の電力変換装置２００の第１誘導部３Ａ、巻線３Ｃの電圧振幅と、本実施の形態の電力変換装置２００のノイズ出力電圧Ｖｎ＿ｏｕｔと、巻線３Ｃの電圧振幅と、を示す図である。
　図１３は、本実施の形態の電力変換装置２００における共振周波数ｆｒｅｓ２より高い周波数領域（７００ｋＨｚ）における第１誘導部３Ａ、巻線３Ｃの電流位相と、本実施の形態の電力変換装置２００の第１誘導部３Ａ、巻線３Ｃの電圧振幅と、本実施の形態の電力変換装置２００のノイズ出力電圧Ｖｎ＿ｏｕｔと、巻線３Ｃの電圧振幅と、を示す図である。 FIG. 12 shows the current phase of the first induction section 3A and the winding 3C in the frequency range (100 kHz) lower than the resonance frequency fres2 in the power converter 200 of the present embodiment, and the current phase of the first induction part 3A and the winding 3C of the power converter 200 of the present embodiment. 1 is a diagram showing the voltage amplitude of the first induction section 3A and the winding 3C, the noise output voltage Vn_out of the power converter 200 of the present embodiment, and the voltage amplitude of the winding 3C.
FIG. 13 shows the current phase of the first induction section 3A and the winding 3C in a frequency range (700 kHz) higher than the resonance frequency fres2 in the power conversion device 200 of this embodiment, and the current phase of the first induction section 3A and the winding 3C of the power conversion device 200 of this embodiment. 1 is a diagram showing the voltage amplitude of the first induction section 3A and the winding 3C, the noise output voltage Vn_out of the power converter 200 of the present embodiment, and the voltage amplitude of the winding 3C.
　電力ラインＢにおける第１誘導部３Ａとコンデンサ３Ｂの一端の接続点とグランドとの間の電圧で見ると、ｆｒｅｓ２よりも高い周波数においても、第１誘導部３Ａと巻線３Ｃには同位相の電流が流れる。しかしながら、図１３に示すように、ｆｒｅｓ２よりも高い周波数では、巻線３Ｃは第１誘導部３Ａと同位相の電圧となり、この場合ノイズの出力電圧Ｖｎ＿ｏｕｔに対して逆相となり、電圧をキャンセルするように働く。そのため、図１１の周波数特性Ｗ４に示されるように、共振周波数ｆｒｅｓ２よりも高い周波数領域（７００ｋＨｚ）において、高いノイズ減衰効果が得られることになる。 Looking at the voltage between the connection point of the first inductive part 3A and one end of the capacitor 3B in power line B and the ground, even at frequencies higher than fres2, the first inductive part 3A and the winding 3C have the same phase. Current flows. However, as shown in FIG. 13, at a frequency higher than fres2, the winding 3C has a voltage that is in the same phase as the first induction section 3A, and in this case, it is in reverse phase with respect to the noise output voltage Vn_out, canceling the voltage. work like that. Therefore, as shown in the frequency characteristic W4 of FIG. 11, a high noise attenuation effect can be obtained in a frequency region (700 kHz) higher than the resonance frequency fres2.
　上記のように構成された本実施の形態のフィルタ回路においては、
前記第２巻線は、前記伝送路との磁気結合が、順結合あるいは逆結合となるように前記磁性体に対する巻回方向が調整されて巻回される、
ものである。 In the filter circuit of this embodiment configured as described above,
The second winding is wound with a winding direction adjusted relative to the magnetic body so that the magnetic coupling with the transmission line is forward coupling or reverse coupling.
It is something.
　このように、第２巻線は、伝送路との磁気結合が、順結合あるいは逆結合となるように磁性体に対する巻回方向が調整された構成とすることで、高いノイズ減衰効果を生じさせる周波数領域を、共振周波数ｆｒｅｓ２よりも高い周波数領域と、共振周波数ｆｒｅｓ２よりも低い周波数領域との間で切り替えることが可能になる。そのため、第２巻線の伝送路との磁気結合を逆結合とすれば、高周波数域において高いノイズ減衰効果を得られる。
　こうして、適用される電力変換装置１００の仕様、設置環境等にあわせた、効果的なノイズ低減が可能となる。 In this way, the second winding has a configuration in which the winding direction with respect to the magnetic body is adjusted so that the magnetic coupling with the transmission line is forward coupling or reverse coupling, thereby producing a high noise attenuation effect. It becomes possible to switch the frequency domain between a frequency domain higher than the resonance frequency fres2 and a frequency domain lower than the resonance frequency fres2. Therefore, if the magnetic coupling between the second winding and the transmission line is reversely coupled, a high noise attenuation effect can be obtained in a high frequency range.
In this way, it is possible to effectively reduce noise in accordance with the specifications, installation environment, etc. of the applied power converter 100.
実施の形態３．
　以下、本願の実施の形態３を、上記実施の形態１と異なる箇所を中心に図を用いて説明する。上記実施の形態１と同様の部分は同一符号を付して説明を省略する。
　図１４は、実施の形態３による電力変換装置３００の概略構成を示す回路図である。
　実施の形態１、２では、第１誘導部３Ａと巻線３Ｃとを磁気結合させた構成によるノーマルモードのノイズに対しての減衰効果を示したが、第１誘導部３Ａと巻線３Ｃとを順結合構成したコモンモードフィルタにおいても同様のノイズ減衰効果が得られ、その構成例を図１４に示す。 Embodiment 3.
Embodiment 3 of the present application will be described below with reference to the drawings, focusing on the differences from Embodiment 1 described above. The same parts as in the first embodiment are given the same reference numerals, and the description thereof will be omitted.
FIG. 14 is a circuit diagram showing a schematic configuration of power conversion device 300 according to the third embodiment.
In Embodiments 1 and 2, the damping effect on normal mode noise was shown by the configuration in which the first induction section 3A and the winding 3C were magnetically coupled, but when the first induction section 3A and the winding 3C were A similar noise attenuation effect can be obtained with a common mode filter having a forward coupling configuration, and an example of the configuration is shown in FIG.
　電力変換器１と負荷２とは、正側直流母線ＢＰと負側直流母線ＢＮとにより互いに接続されており、２つのフィルタ回路１０が正側直流母線ＢＰと負側直流母線ＢＮのそれぞれに設けられる。
　正側直流母線ＢＰに接続されるフィルタ回路１０における第１誘導部３Ａと巻線３Ｃは、共通のコア４Ｐを介して磁気結合される。
　負側直流母線ＢＮに接続されるフィルタ回路１０における第１誘導部３Ａと巻線３Ｃは、共通のコア４Ｎを介して磁気結合される。
　なお、コア４Ｐとコア４Ｎは別体の磁性体である。 The power converter 1 and the load 2 are connected to each other by a positive DC bus BP and a negative DC bus BN, and two filter circuits 10 are provided on each of the positive DC bus BP and the negative DC bus BN. It will be done.
The first induction portion 3A and the winding 3C in the filter circuit 10 connected to the positive DC bus BP are magnetically coupled via a common core 4P.
The first induction section 3A and the winding 3C in the filter circuit 10 connected to the negative DC bus BN are magnetically coupled via a common core 4N.
Note that the core 4P and the core 4N are separate magnetic bodies.
　図１５は、ノイズ減衰量の具体的効果を示すための、本実施の形態の電力変換装置３００と比較例の電力変換装置１００ＥＸ１の周波数特性を比較して示す図である。
　本図において、巻線３Ｃとコンデンサ３Ｄを取り除き、ＬＰＦだけの構成とした実施の形態１に示したフィルタ回路１０ＥＸ１の周波数特性Ｗ２と、本実施の形態の電力変換装置３００の周波数特性Ｗ５とを示している。
　実施の形態１の電力変換装置１００の周波数特性Ｗ１を得るために用いた同じ諸元を、本実施の形態の電力変換装置３００の第１誘導部３Ａ、３Ｃとコンデンサ３Ｂ、３Ｄに適用している。 FIG. 15 is a diagram showing a comparison of the frequency characteristics of the power conversion device 300 of the present embodiment and the power conversion device 100EX1 of the comparative example, in order to show a specific effect of the amount of noise attenuation.
In this figure, the frequency characteristic W2 of the filter circuit 10EX1 shown in the first embodiment, in which the winding 3C and the capacitor 3D are removed and only the LPF is configured, and the frequency characteristic W5 of the power conversion device 300 of the present embodiment are shown. It shows.
The same specifications used to obtain the frequency characteristic W1 of the power converter 100 of the first embodiment are applied to the first induction parts 3A, 3C and the capacitors 3B, 3D of the power converter 300 of the present embodiment. There is.
　図１４に示されるように、本実施の形態の電力変換装置３００においても、一般的にノイズ減衰効果を得られにくいｆｒｅｓ１を中心とした低周波数領域と、ｆｒｅｓ２を中心とした特定周波数領域において、高いノイズ減衰効果を得ることができる。 As shown in FIG. 14, in the power conversion device 300 of this embodiment as well, in the low frequency region centered on fres1, where it is generally difficult to obtain a noise attenuation effect, and in the specific frequency region centered on fres2, A high noise attenuation effect can be obtained.
　上記のように構成された本実施の形態の電力変換装置においては、
直流電力を出力する第１直流回路と、直流電力が入力される第２直流回路とを備え、
前記第１直流回路と前記第２直流回路とが、前記伝送路としての正側直流母線と負側直流母線とにより互いに接続され、
実施の形態１あるいは実施の形態２に示されるフィルタ回路が、前記正側直流母線と前記負側直流母線のそれぞれに設けられる、
ものである。 In the power conversion device of this embodiment configured as described above,
comprising a first DC circuit that outputs DC power and a second DC circuit that inputs DC power,
The first DC circuit and the second DC circuit are connected to each other by a positive DC bus and a negative DC bus as the transmission line,
The filter circuit shown in Embodiment 1 or Embodiment 2 is provided on each of the positive DC bus and the negative DC bus,
It is something.
　これにより、コモンモードノイズに対しても、一般的にノイズ減衰効果を得られにくいｆｒｅｓ１を中心とした低周波数領域と、ｆｒｅｓ２を中心とした特定周波数領域において、高いノイズ減衰効果を得ることができる。 As a result, it is possible to obtain a high noise attenuation effect against common mode noise in the low frequency region centered on fres1, where it is generally difficult to obtain a noise attenuation effect, and in the specific frequency region centered on fres2. .
実施の形態４．
　以下、本願の実施の形態４を、上記実施の形態１と異なる箇所を中心に図を用いて説明する。上記実施の形態１と同様の部分は同一符号を付して説明を省略する。
　図１６は、実施の形態４によるフィルタ回路４１０を備えた電力変換装置４００の概略構成を示す回路図である。
　実施の形態１では、直列回路３Ｓを一つのみ備える構成としていた。本実施の形態ではこの直列回路３Ｓを複数備え、それぞれを電力ラインＢに接続する構成としている。
　これにより、元々のＬＰＦの特性の共振点ｆｒｅｓ２を下げて高いノイズ減衰量を得ながら、複数の特定の減衰域を持つＢＥＦを一つのコア４により構成することができる。 Embodiment 4.
Embodiment 4 of the present application will be described below with reference to the drawings, focusing on the differences from Embodiment 1 described above. The same parts as in the first embodiment are given the same reference numerals, and the description thereof will be omitted.
FIG. 16 is a circuit diagram showing a schematic configuration of a power conversion device 400 including a filter circuit 410 according to the fourth embodiment.
In the first embodiment, the configuration includes only one series circuit 3S. In this embodiment, a plurality of series circuits 3S are provided, each of which is connected to the power line B.
As a result, a BEF having a plurality of specific attenuation ranges can be configured using one core 4 while lowering the resonance point fres2 of the original LPF characteristics and obtaining a high amount of noise attenuation.
　なお、図では巻線３Ｃとコンデンサ３Ｄとによる直列回路３Ｓと、巻線３Ｇとコンデンサ３Ｈとによる直列回路３Ｓの、２つの直列回路３Ｓを備える例を示したが、直列回路３Ｓの数は２以上であっても構わない。
　ここで、巻線３Ｇの自己インダクタンスをＬ３、流れる電流をｉ３、両端電圧をＶ３、第１誘導部３Ａ、巻線３Ｃ間の相互インダクタンスをＭ１２、第１誘導部３Ａ、巻線３Ｇ間の相互インダクタンスをＭ１３、巻線３Ｃ、巻線３Ｇ間の相互インダクタンスをＭ２３とすると、第１誘導部３Ａ、巻線３Ｃ、巻線３Ｇの両端電圧Ｖ１、Ｖ２、Ｖ３はそれぞれ以下の（式５）～（式７）により示される。 Note that although the figure shows an example including two series circuits 3S: a series circuit 3S consisting of a winding 3C and a capacitor 3D, and a series circuit 3S consisting of a winding 3G and a capacitor 3H, the number of series circuits 3S is 2. It does not matter if it is more than that.
Here, the self-inductance of the winding 3G is L3, the flowing current is i3, the voltage at both ends is V3, the mutual inductance between the first induction section 3A and the winding 3C is M12, the mutual inductance between the first induction section 3A and the winding 3G is If the inductance is M13, and the mutual inductance between the winding 3C and the winding 3G is M23, the voltages V1, V2, and V3 across the first inductor 3A, the winding 3C, and the winding 3G are as follows (Equation 5) ~ It is shown by (Formula 7).
Figure JPOXMLDOC01-appb-M000005
Figure JPOXMLDOC01-appb-M000005
Figure JPOXMLDOC01-appb-M000006
Figure JPOXMLDOC01-appb-M000006
Figure JPOXMLDOC01-appb-M000007
Figure JPOXMLDOC01-appb-M000007
　上記（式５）～（式７）によりで示されるように、相互インダクタンスによって、第１誘導部３Ａによるノイズの電圧降下が増加して共振点が下がり、ＢＥＦを複数点構成しつつ実施の形態１同様のノイズ低減効果が得られる。 As shown by the above (Formula 5) to (Formula 7), due to the mutual inductance, the voltage drop of the noise due to the first induction section 3A increases and the resonance point lowers. A noise reduction effect similar to No. 1 can be obtained.
　本願は、様々な例示的な実施の形態及び実施例が記載されているが、１つ、または複数の実施の形態に記載された様々な特徴、態様、及び機能は特定の実施の形態の適用に限られるのではなく、単独で、または様々な組み合わせで実施の形態に適用可能である。
従って、例示されていない無数の変形例が、本願に開示される技術の範囲内において想定される。例えば、少なくとも１つの構成要素を変形する場合、追加する場合または省略する場合、さらには、少なくとも１つの構成要素を抽出し、他の実施の形態の構成要素と組み合わせる場合が含まれるものとする。 Although this application describes various exemplary embodiments and examples, various features, aspects, and functions described in one or more embodiments may be applicable to a particular embodiment. The present invention is not limited to, and can be applied to the embodiments alone or in various combinations.
Therefore, countless variations not illustrated are envisioned within the scope of the technology disclosed herein. For example, this includes cases where at least one component is modified, added, or omitted, and cases where at least one component is extracted and combined with components of other embodiments.
１　電力変換器（第１直流回路）、２　負荷（第２直流回路）、３Ｂ　コンデンサ（第１容量素子）、３Ｃ　巻線（第２巻線）、３Ｄ　コンデンサ（第２容量素子）、３Ｓ　直列回路（直列体）、４　コア（磁性体）、１０，２１０，４１０　フィルタ回路、Ｂ　電力ライン（伝送路）、ＢＰ　正側直流母線（伝送路）、ＢＮ　負側直流母線（伝送路）、１００，２００，３００，４００　電力変換装置。 1 Power converter (first DC circuit), 2 Load (second DC circuit), 3B capacitor (first capacitive element), 3C winding (second winding), 3D capacitor (second capacitive element), 3S series Circuit (series body), 4 Core (magnetic material), 10,210,410 Filter circuit, B Power line (transmission line), BP Positive side DC bus (transmission line), BN Negative side DC bus (transmission line), 100 , 200, 300, 400 power conversion device.

Claims (7)

  1. 直流電力を伝送する伝送路に一端が接続され、他端が基準電位に接続された第１容量素子と、前記伝送路に一端が接続され、他端が前記基準電位に接続された、第２巻線と第２容量素子との直列体と、を備え、
    前記第２巻線は、前記伝送路と相互に磁気結合するように設けられる、
    フィルタ回路。     a first capacitive element having one end connected to a transmission path for transmitting DC power and the other end connected to a reference potential; and a second capacitive element having one end connected to the transmission path and the other end connected to the reference potential. A series body of a winding and a second capacitive element,
    The second winding is provided to be magnetically coupled to the transmission line,
    filter circuit.
  2. 前記第２巻線と前記伝送路とは、該伝送路の外周面を取り囲むように閉磁路を形成する共通の磁性体を介して相互に磁気結合される、
    請求項１に記載のフィルタ回路。     The second winding and the transmission path are magnetically coupled to each other via a common magnetic body forming a closed magnetic path surrounding the outer peripheral surface of the transmission path.
    The filter circuit according to claim 1.
  3. 前記第２巻線は、前記伝送路との磁気結合が、順結合あるいは逆結合となるように前記磁性体に対する巻回方向が調整されて巻回される、
    請求項２に記載のフィルタ回路。     The second winding is wound with a winding direction adjusted relative to the magnetic body so that the magnetic coupling with the transmission line is forward coupling or reverse coupling.
    The filter circuit according to claim 2.
  4. 前記第２巻線の断面積は、前記伝送路の断面積よりも小さく構成される、
    請求項１から請求項３のいずれか１項に記載のフィルタ回路。     The cross-sectional area of the second winding is configured to be smaller than the cross-sectional area of the transmission line.
    The filter circuit according to any one of claims 1 to 3.
  5. 複数の前記直列体を備え、
    各前記直列体は、前記伝送路と前記基準電位との間で並列接続して設けられる、
    請求項１から請求項４のいずれか１項に記載のフィルタ回路。     comprising a plurality of the series bodies,
    Each of the series bodies is connected in parallel between the transmission path and the reference potential,
    The filter circuit according to any one of claims 1 to 4.
  6. 前記伝送路は、該伝送路の長手方向に対して垂直な方向に窪む凹部を備え、
    前記磁性体は、前記凹部内において前記伝送路の外周面を取り囲むように設けられる、
    請求項２または請求項３に記載のフィルタ回路。     The transmission path includes a recess that is recessed in a direction perpendicular to the longitudinal direction of the transmission path,
    The magnetic material is provided in the recess so as to surround an outer peripheral surface of the transmission path.
    The filter circuit according to claim 2 or 3.
  7. 直流電力を出力する第１直流回路と、直流電力が入力される第２直流回路とを備え、
    前記第１直流回路と前記第２直流回路とが、前記伝送路としての正側直流母線と負側直流母線とにより互いに接続され、
    請求項１から請求項６のいずれか１項に記載のフィルタ回路が、前記正側直流母線と前記負側直流母線のそれぞれに設けられる、
    電力変換装置。     comprising a first DC circuit that outputs DC power and a second DC circuit that inputs DC power,
    The first DC circuit and the second DC circuit are connected to each other by a positive DC bus and a negative DC bus as the transmission line,
    The filter circuit according to any one of claims 1 to 6 is provided on each of the positive DC bus and the negative DC bus,
    Power converter.
PCT/JP2022/021877 2022-05-30 2022-05-30 Filter circuit and power conversion device WO2023233446A1 (en)

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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2004297551A (en) * 2003-03-27 2004-10-21 Tdk Corp Noise filter device and switching power supply
JP2016100618A (en) * 2014-11-18 2016-05-30 北川工業株式会社 Output noise reduction device
JP2021141487A (en) * 2020-03-06 2021-09-16 株式会社豊田中央研究所 Noise filter

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2004297551A (en) * 2003-03-27 2004-10-21 Tdk Corp Noise filter device and switching power supply
JP2016100618A (en) * 2014-11-18 2016-05-30 北川工業株式会社 Output noise reduction device
JP2021141487A (en) * 2020-03-06 2021-09-16 株式会社豊田中央研究所 Noise filter

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