WO2021161945A1 - Power source stabilization device - Google Patents

Abstract

A power source stabilization device (20) positioned in series between a DC power source (11) and a power load (70). A current transformer (30) includes a primary winding (31) and a secondary winding (32). A control power source (50): calculates a voltage control item for offsetting ripple voltage of the DC voltage (Vb), and a current control item for injecting an offset current (Ict) to the secondary winding (32) so as to offset a DC current component that generates DC magnetomotive force in the primary winding (31); and causes an output voltage generation circuit (52) to operate on the basis of a voltage command value obtained by adding the voltage control item and the current control item, thereby applying an output voltage (Vout) to the secondary winding (32). A current sensor (40) detects the differential between the magnetomotive force caused by primary current (Iaux) flowing to the primary winding (31) and the magnetomotive force caused by the offset current (Ict) flowing to the sensor secondary winding. The control power source (50) calculates the current control item so as to bring the differential between the magnetomotive forces close to 0.

Description

Power stabilizer Cross-reference of related applications

This application is based on Patent Application No. 2020-021067 filed on February 11, 2020, and the contents of the description are incorporated herein by reference.

This disclosure relates to a power supply stabilizer.

Conventionally, a device that compensates for the ripple component of the DC voltage supplied from the power supply to the DC system and stabilizes the DC voltage is known. For example, the DC reactor device disclosed in Patent Document 1 includes a DC reactor main winding, a DC reactor auxiliary winding that is magnetically coupled to the DC reactor main winding via a DC reactor core, a voltage source, and a control means. And. The DC reactor main winding is connected between the rectifier circuit that rectifies the AC power supply and the smoothing capacitor on the load side. The voltage source is connected to the DC reactor auxiliary winding to generate an arbitrary voltage waveform. The control means controls the voltage source so as to suppress the magnetic saturation of the DC reactor and compensate for the DC ripple.

Japanese Patent No. 3763745

The DC reactor device of the embodiment corresponding to claim 2 of Patent Document 1 detects a DC primary winding current detection circuit that detects a DC current on the DC primary winding side and a DC current on the DC secondary winding side. It has a DC secondary winding current detection circuit. The magnetic saturation suppression control unit of the control unit calculates the difference between the primary winding current detection value and the secondary winding current detection value. Since this configuration requires two current sensors, there is a problem that the device becomes large.

An object of the present disclosure is to provide a power supply stabilizing device capable of simplifying the configuration of a current sensor and reducing the size of the device.

The power supply stabilizer of the present disclosure is arranged in series between a DC distribution line and a power load in a DC system that supplies power from a DC power source to a power load via a DC distribution line, and is applied to the power load from the DC power source. It stabilizes the DC voltage that is applied. This power supply stabilizer includes a current transformer, a current sensor, and a control power supply.

Current transformers include transformer cores, primary windings and secondary windings. The primary winding is connected in series between the DC distribution line and the power load and penetrates the transformer core. _{The secondary winding is wound around the transcore for N 1} turns, which is a predetermined number of turns, and is magnetically coupled to the primary winding via the transcore. The current sensor has a sensor core through which the primary winding penetrates. The secondary winding for the sensor connected in series with the secondary winding is wound around the sensor core in the same number of N ₁ turns as the predetermined number of turns.

The control power supply calculates the voltage control term and the current control term. The voltage control term is used to cancel the ripple voltage of the DC voltage. The current control term is used to inject a canceling current into the secondary winding so as to cancel the DC current component that generates a DC magnetomotive force in the primary winding among the primary currents flowing in the primary winding. Then, the control power supply applies an output voltage to the secondary winding by operating the output voltage generation circuit based on the voltage command value obtained by adding the voltage control term and the current control term.

The current sensor detects the difference between the magnetomotive force due to the primary current flowing in the primary winding and the magnetomotive force due to the canceling current flowing in the secondary winding for the sensor. The control power supply calculates the current control term so that the difference in the magnetomotive force approaches zero.

In the power supply stabilizer of the present disclosure, one current sensor detects the difference between the magnetomotive force due to the primary current and the magnetomotive force due to the canceling current, and the control circuit calculates the current control term based on the difference in the magnetomotive force. .. Therefore, the number of current sensors can be reduced and the device can be downsized as compared with the conventional technique of Patent Document 1 which requires two current sensors.

The above objectives and other objectives, features and advantages of the present disclosure will be clarified by the following detailed description with reference to the accompanying drawings. The drawing is
FIG. 1 is a schematic configuration diagram of a DC system including a power supply stabilizer according to an embodiment. FIG. 2 is a diagram of a voltage control block and a current control block of a control circuit. FIG. 3 is a frequency characteristic diagram of the HPF of the voltage control block and the LPF of the current control block. FIG. 4 is a configuration diagram of a current sensor according to an embodiment. FIG. 5 is a block diagram of the current sensor of another embodiment. FIG. 6 is a voltage control block diagram of another embodiment.

(One Embodiment)
The power supply stabilizing device according to the embodiment of the present disclosure will be described with reference to the drawings. As shown in FIG. 1, the power supply stabilizing device 20 is in series between the DC distribution line 12 and the power load 70 in the DC system 90 that supplies power from the DC power supply 11 to the power load 70 via the DC distribution line 12. Placed in. In the configuration in which the power supply stabilizing devices 20 are arranged in series, the shared voltage of the DC power supply 11 by the power supply stabilizing device 20 can be reduced, and the device capacity can be reduced.

There may be a plurality of power loads 70 regardless of the type. However, in general, the power load 70 has a reactor component and a capacitance component, and an LC resonance circuit is formed. For example, when a harmonic current is generated at the power load 70, this may act as an excitation source and cause LC resonance. Alternatively, resonance may be excited due to load fluctuation or the like. When a resonance phenomenon occurs, the DC voltage Vb fluctuates greatly due to the impedance drop of the DC distribution wire 12, or an excessive resonance current flows, causing an overvoltage or an overcurrent. Therefore, in order to prevent such an event, it is required to stabilize the DC voltage Vb in the DC system 90.

The power supply stabilizing device 20 of the present embodiment includes a current transformer 30, a current sensor 40, a control power supply 50, and the like. The power supply stabilizing device 20 compensates for the voltage ripple component of the DC voltage Vb by the output voltage Vout generated by the control power supply 50, and cancels the DC current component of the primary current Iaux. In this way, the power supply stabilizing device 20 stabilizes the DC voltage Vb applied from the DC power supply 11 to the power load 70. The DC voltage Vb is detected by a voltage sensor or the like and is acquired by the control circuit 60 of the control power supply 50.

The current transformer 30 includes a primary winding 31 and a secondary winding 32. The primary winding 31 is connected in series between the DC distribution line 12 and the power load 70. The secondary winding 32 is connected to the control power supply 50 and magnetically couples with the primary winding 31 via the transformer core 33. The number of turns of the primary winding 31 is one turn, and the number of turns of the secondary winding 32 is N ₁ turn. The value of the number of turns N ₁ turn of the secondary winding 32 can be arbitrarily set according to the DC voltage Vb of the DC power supply 11 and the current capacity and withstand voltage of the control power supply 50.

A primary current Iaux containing both a DC current component and a ripple current component flows through the primary winding 31 which is a path from the DC power supply 11 to the power load 70. The primary current Iaux generates a magnetic flux in the transformer core 33. Then, the dimension of the transformer core 33 is determined by the density of the magnetic flux. The direct current component of the primary current Iaux increases the magnetic flux density of the transcore 33 and acts in the direction of increasing the dimensions. Therefore, the current injected into the secondary winding 32 by the output voltage Vout of the control power supply 50 in order to reduce the size of the current transformer 30 and cancel the DC current component of the primary winding 31 is referred to as an offset current Ict.

The current sensor 40 _{detects the detected current Iss represented by "Iux-N 1} Ict" and notifies the control circuit 60 of it. The detailed configuration of the current sensor 40 will be described later with reference to FIG. 4, and the difference from the current detection configuration of the prior art will be described first. In the prior art, two current sensors are used: a current sensor that detects the primary current Iux flowing through the primary winding 31 and a current sensor that detects the canceling current Ict flowing through the secondary winding 32.

On the other hand, in the present embodiment, " _{a value obtained by multiplying the canceling current Ict by the number of turns N 1} turn of the secondary winding 32 multiplied by the primary current Iaux" is detected by one current sensor 40. Here, the value obtained by multiplying the current by the number of turns is the magnetomotive force (unit: [AT (ampere-turn)]). The primary value obtained by multiplying the number of turns one turn of the current Iaux the primary winding 31 is magnetomotive force Iaux [AT], offset current value obtained by multiplying the number of turns N ₁ turns of the secondary winding 32 to Ict magnetomotive force N ₁ Ict [AT].

That is, the detected current Isns (= Iux-N ₁ Ict) means the difference between the magnetomotive force due to the primary current Iaux flowing in the primary winding 31 and the magnetomotive force due to the canceling current Ict flowing in the secondary winding 32. The current sensor 40 of the present embodiment directly detects the difference in the magnetomotive force and outputs the difference to the control power supply 50.

The control power supply 50 includes an operating power supply 51, an output voltage generation circuit 52, and a control circuit 60. The operating power supply 51 may be configured by a dedicated DC power supply as shown in FIG. Alternatively, the DC voltage Vb of the DC power supply 11 may be used by being connected to the distribution line branched from the DC distribution line 12.

The output voltage generation circuit 52 operates in accordance with a command from the control circuit 60 to generate an output voltage Vout applied to the secondary winding 32. For example, the output voltage generation circuit 52 is composed of a high-speed PWM-controlled single-phase inverter. The single-phase inverter can be realized, for example, according to the half-bridge circuit described in FIG. 10 of Patent Document 1 or the full-bridge circuit described in FIG. By switching the plurality of switching elements constituting the bridge circuit by PWM control, the power supply voltage Einv of the operating power supply 51 is converted to generate the output voltage Vout.

The control circuit 60 calculates a duty ratio D for PWM control of the output voltage generation circuit 52 based on the DC voltage Vb, the power supply voltage Einv of the operating power supply 51, the output voltage Vout, and the detection current Isns of the current sensor 40. .. FIG. 2 shows the arithmetic configuration of the control circuit 60. The control circuit 60 includes a voltage control block 61, a current control block 62, an adder 63 and a divider 64.

The voltage control block 61 calculates a ^{voltage control term V *} _v for generating an output voltage Vout that cancels the voltage ripple of the DC voltage Vb. The voltage control block 61 having a feedback control configuration shown in FIG. 2 includes a high-pass filter (in the figure and hereinafter “HPF”) 611, a deviation calculator 612, and a voltage controller 613. HPF611 cuts the DC component in the DC voltage Vb including the ripple voltage and extracts the ^{ripple voltage component Vout *.}

The deviation calculator 612 calculates ^{the voltage deviation between the ripple voltage component Vout *} and the output voltage Vout. The voltage controller 613 calculates the ^{voltage control term V *} _v by multiplying the voltage deviation by the proportional control gain Kpv. ^{Alternatively, the voltage control term V *} _v may be calculated by the proportional integral control which is the proportional control plus the integral control. The illustration of the integration control block is omitted.

The current control block 62 calculates ^{the current control term V *} _i for controlling the DC current component of the current transformer 30. The current control block 62 shown in FIG. 2 includes a low-pass filter (in the figure and hereinafter “LPF”) 621, a deviation calculator 622, and a current controller 623. The LPF621 outputs a filtered detection current Isns_LPF that cuts off components other than the DC current component in the detection current Isns (= Iux-N _{1 Ict) detected by the current sensor 40.}

Here, assuming that the DC current component included in the primary current Iux is Ict ^* (not shown in FIG. 2), the detected current after filtering Isns_LPF is "Ict ^* -N ₁ Ict", that is, the DC current component of the primary current Iux. It corresponds to the difference between the magnetomotive force due to Ict ^{* and the magnetomotive force due to the canceling current Ict.}

The deviation calculator 622 calculates the current deviation between the detected current Isns_LPF after filtering and the target value 0. The current controller 623 calculates the ^{current control term V *} _i by multiplying the current deviation by the proportional control gain Kpi. The gain Kpi includes a coefficient of the resistance dimension that converts the value of the current dimension into the voltage dimension. ^{Alternatively, the current control term V *} _i may be calculated by the proportional integral control which is the proportional control plus the integral control. The illustration of the integration control block is omitted.

The adder 63 adds the voltage control term V ^* _v and the current control term V ^* _i to calculate ^{the voltage command value Vinv *} of the output voltage generation circuit 52. The divider 64 calculates the duty ratio D of the output voltage generation circuit 52 by dividing the voltage command value Vinv ^{* by the power supply voltage Einv of the operating power supply 51.} When the output voltage generation circuit 52 operates by PWM control based on the duty ratio D, the output voltage Vout is applied to the secondary winding 32 of the current transformer 30.

When the voltage control term V ^* _v of the voltage command value Vinv ^* is reflected in the output voltage Vout, the output voltage Vout corresponding to the ripple voltage included in the DC voltage Vb is generated in the primary winding 31, and the DC voltage Vb Fluctuations are compensated. In the present embodiment, since the fluctuation of the DC voltage Vb is detected and the output voltage Vout is controlled, accurate control is possible.

Further, the current control term V ^* _i of the voltage command value Vinv ^* is reflected in the output voltage Vout, so that a magnetic flux component that superimposes the canceling current Ict on the secondary winding 32 is supplied. Therefore, the canceling current Ict flowing in the secondary winding 32 cancels the DC magnetic flux component generated in the primary winding 31 of the current transformer 30. Therefore, the magnetic path cross-sectional area of the current transformer 30 can be reduced, and the current transformer 30 can be miniaturized.

Here, if the control response of the current control block 62 is faster than the control response of the voltage control block 61, the magnetic flux in the current transformer 30 is always controlled to zero, so that the output voltage Vout cannot be appropriately generated. .. That is, control interference between the ripple voltage cancellation and the DC magnetic flux control occurs. Therefore, control interference is prevented by configuring the control response of the voltage control block 61 to be faster than the control response of the current control block 62.

For example, as shown in FIG. 3, the cutoff frequency fcoL of the LPF621 of the current control block 62 is set lower than the cutoff frequency fcoH of the HPF611 of the voltage control block 61. Alternatively, by adjusting the relationship between the gain Kpi of the current controller 623 and the gain Kpv of the voltage controller 613, the control response of the voltage control block 61 can be made faster than the control response of the current control block 62. Is.

Next, with reference to FIG. 4, the detailed configuration of the current transformer 30 and the current sensor 40 will be described. In the current transformer 30, the primary winding 31 penetrates the ring-shaped transformer core 33, and the _{secondary winding 32 of N 1} turn is wound around the current transformer 30. The number of turns of the primary winding 31 penetrating the transformer core 33 is one turn. The secondary winding 32 is connected to the control power supply 50.

In the current sensor 40, the primary winding 31 penetrates the ring-shaped sensor core 43, and the same number of turns as the secondary winding 32 of the current transformer 30, that is, the sensor secondary winding 42 of _{N 1 turn is wound.} There is. The number of turns of the primary winding 31 penetrating the sensor core 43 is one turn. The sensor secondary winding 42 is connected in series with the secondary winding 32. A canceling current Ict flows through the secondary winding 32 and the sensor secondary winding 42 in a direction that cancels the magnetic flux created by the primary current Iaux.

_{With this configuration, the sensor core 43 has a difference (Iaux-N 1)} between the magnetomotive force due to the primary current Iaux flowing in the primary winding 31 of _one turn and the magnetomotive force due to the canceling current Ict flowing in the secondary winding 32 of N 1 turn. A magnetic flux proportional to Ict) is generated. A hall sensor 45 for detecting the amount of magnetic flux of the sensor core 43 is incorporated in the cross section of the sensor core 43. The amplifier (“AMP” in the figure) 46 amplifies the amount of magnetic flux detected by the Hall sensor 45.

The current sensor 40 illustrated in FIG. 4 is of a magnetic equilibrium type, and the amplified magnetic flux amount is _{applied to the N 2} turn feedback winding 47 wound around the sensor core 43. The detection voltage Vdcct of the shunt resistor 48 connected to the feedback winding 47 is input to the control power supply 50. The detected voltage Vdcct is expressed by the following equation.
Vdcct∝ (Iaux-N ₁ Ict) / N ₂

The detection voltage Vdcct in the control circuit 60 of the control power supply 50 _{is converted into the detection current "Iux-N 1} Ict (= Isns)" and input to the current control block 62. Then, as described above with reference to FIG. 2, the current control term V ^* _i is calculated so that the detected current Iss approaches the target value 0 in the current control block 62. That is, the control power supply 50 injects the canceling current Ict into the secondary winding 32 so that the magnetomotive force due to the primary current Iaux and the magnetomotive force due to the canceling current Ict match.

As described above, the current sensor 40 of the present embodiment does not detect the primary current Iux itself and the canceling current Ict itself, but rather the primary winding 31 penetrating the sensor core 43 and the sensor wound around the sensor core 43. The difference in the magnetomotive force of the current flowing through the secondary winding 42 is detected. Therefore, the comparison operation in the current control block 62 can be directly realized.

By the way, in the embodiment corresponding to claim 2 shown in FIGS. 4 and 5 of Patent Document 1 (Japanese Patent No. 376,745), the primary winding current detection value detected by the DC primary winding current detection circuit and the DC secondary winding current detection value are detected. The difference from the secondary winding current detection value detected by the secondary winding current detection circuit is calculated. The voltage source generates a voltage waveform based on the voltage command data based on this difference, so that the magnetic saturation of the iron core (trans core) is suppressed.

Since the configuration of Patent Document 1 requires two current sensors, there is a problem that the device becomes large. On the other hand, in the present embodiment, one current sensor 40 detects the difference between the magnetomotive force due to the primary current Iaux and the magnetomotive force due to the canceling current Ict, and the control circuit 60 detects the difference between the magnetomotive forces and the current control term V based on the difference in the magnetomotive forces. ^* Calculate _i. Therefore, the number of current sensors can be reduced and the device can be miniaturized.

Further, Japanese Patent Application Laid-Open No. 2018-74668 (hereinafter referred to as "references") discloses a technique for imparting a resistive impedance to the secondary winding of a current transformer and stabilizing the voltage of a DC distribution system due to a negative resistance. Has been done. For example, in the third embodiment corresponding to claim 8 of the reference, the AC / DC converter connected to the secondary winding is merely controlled so as to have a constant resistance characteristic, and the vibration component of the electric power is sufficiently controlled. Cannot be compensated. Further, a capacitor for removing the DC component of the primary winding current and a tertiary winding for canceling the DC magnetic flux component are provided. While reducing the DC component contributes to the miniaturization of the transcore size, there is a problem that the device becomes larger when a capacitor for removing the DC component and a tertiary winding are added.

On the other hand, in the present embodiment, the ripple voltage component Vout ^* of the DC voltage Vb is detected, and the winding voltage of the current transformer 30 is controlled so as to cancel the ^{ripple voltage component Vout *.} Therefore, the compensation characteristic of the ^{ripple voltage component Vout *} can be improved without adding a capacitor or a tertiary winding as in the reference.

(Other embodiments)
(A) As the configuration of the output voltage generation circuit 52 of the control power supply 50, a chopper circuit or the like disclosed in Patent Document 1 may be used in addition to the single-phase inverter. Further, the operation method of the output voltage generation circuit 52 is not limited to the PWM control method based on the duty ratio D, and the output voltage Vout may be generated by any method.

(B) The location where the voltage Vb of the DC power supply 11 is detected is not limited to "between the DC distribution line 12 and the current transformer 30" illustrated in FIG. 1, but "between the current transformer 30 and the power load 70". It may be. That is, the voltage across the power load 70 may be detected.

(C) The configuration of the current sensor 40 is not limited to the magnetic equilibrium type illustrated in FIG. 4, and may be an open-loop Hall element type as shown in FIG. In the open loop type, the output of the Hall element 45 amplified by the amplifier 46 is input to the control power supply 50.

(D) The configuration of the voltage control block 61 of the control circuit 60 is not limited to the configuration in which the output voltage Vout is fed back ^{to the ripple voltage component Vout * obtained by processing the DC voltage Vb with the HPF611.} As shown in FIG. 6, in the feedforward controller 614, the voltage control term V ^* _v may be calculated by ^{multiplying the ripple voltage component Vout * by the gain Kffv.}

(E) In the current control block 62 of the control circuit 60, the LPF 621 may be provided after the deviation calculator 622 with respect to the configuration of FIG. It is equivalent if LPF621 is configured in the current control loop. Further, the averaging process may be performed instead of the filter process by LPF621.

As described above, the present disclosure is not limited to the above-described embodiment, and can be implemented in various forms without departing from the spirit of the present embodiment.

The control circuits and methods thereof described in the present disclosure are realized by a dedicated computer provided by configuring a processor and memory programmed to perform one or more functions embodied by a computer program. May be done. Alternatively, the control circuit and method thereof described in the present disclosure may be realized by a dedicated computer provided by configuring the processor with one or more dedicated hardware logic circuits. Alternatively, the control circuit and method thereof described in the present disclosure may be a combination of a processor and memory programmed to perform one or more functions and a processor composed of one or more hardware logic circuits. It may be realized by one or more dedicated computers configured. Further, the computer program may be stored in a computer-readable non-transitional tangible recording medium as an instruction executed by the computer.

This disclosure has been described in accordance with the embodiments. However, the present disclosure is not limited to such embodiments and structures. The present disclosure also includes various variations and variations within an equal range. Also, various combinations and forms, as well as other combinations and forms that include only one element, more, or less, are within the scope and ideas of the present disclosure.

Claims (2)

1. In a DC system that supplies power from a DC power supply (11) to a power load (70) via a DC distribution line (12), the DC power supply is arranged in series between the DC distribution line and the power load, and the DC power supply is used as described above. A power supply stabilizer that stabilizes the direct current voltage (Vb) applied to the power load.
   A transformer core (33), a primary winding (31) connected in series between the DC distribution line and the power load and penetrating the transformer core, and an N1 turn wound around the transformer core, which is a predetermined number of turns. A current transformer (30) including a secondary winding (32) that magnetically couples with the primary winding via the transformer core.
   The sensor core (43) through which the primary winding penetrates, and the secondary winding (42) for a sensor connected in series with the secondary winding is wound around the sensor core in the same number of N1 turns as the predetermined number of turns. With the current sensor (40)
   A DC electromotive force is generated in the primary winding of the voltage control term (V * _v) for canceling the ripple voltage (Vout *) of the DC voltage and the primary current (Iaux) flowing in the primary winding. The current control term (V * _i) for injecting the canceling current (Ict) into the secondary winding is calculated so as to cancel the direct current component, and the voltage control term and the current control term are added. A control power supply (50) that applies an output voltage (Vout) to the secondary winding by operating the output voltage generation circuit (52) based on the voltage command value (Vinv *) obtained in the above process.
   With
   The current sensor detects the difference between the magnetomotive force due to the primary current flowing through the primary winding and the magnetomotive force due to the canceling current flowing through the secondary winding for the sensor.
   The control power supply is a power supply stabilizing device that calculates the current control term so that the difference in the magnetomotive force approaches zero.
2. The power supply stabilizing device according to claim 1, wherein the control response of the voltage control term is set to be faster than the control response of the current control term.

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