WO2016190819A1

WO2016190819A1 - Circuit and method for mitigating power ripple

Info

Publication number: WO2016190819A1
Application number: PCT/SG2016/050253
Authority: WO
Inventors: Ramprakash KATHIRESAN; Cikai YE; Pritam Das; Sanjib Kumar PANDA; Thomas Guenter REINDL
Original assignee: National University Of Singapore
Priority date: 2015-05-28
Filing date: 2016-05-27
Publication date: 2016-12-01

Abstract

A circuit for mitigating power-ripple in a power supplied between a DC element and a power converter connected to a single-phase AC power grid, systems for mitigating power- ripple and a method for mitigating power-ripple in a power supplied between a DC element and a power converter connected to a single-phase AC power grid is provided. The circuit is capable of being electrically coupled in parallel and between the DC element and the power converter, and may comprise an auxiliary circuit comprising an inductor element for providing energy storage; and a control circuit configured to generate one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the DC element and the power converter.

Description

CIRCUIT AND METHOD FOR MITIGATING POWER RIPPLE

TECHNICAL FIELD

The present disclosure relates broadly to a circuit and method for mitigating power ripple. More particularly, the present disclosure relates to the mitigation of power ripple in a power supplied between a DC element and a power converter connected to a single-phase AC power grid.

BACKGROUND

In recent times, the number of Solar-Photovoltaic system installations is growing exponentially. Such Solar-Photovoltaic system installations are bringing in GWs (Giga Watts) worth of power into the renewable energy mix and they are turning out to be one of the chief renewable energy resources. Grid connected PV (Photovoltaic) systems are among the most common of solar photovoltaic systems, even though off-grid installations are significantly increasing. For grid integration of PV systems, it is necessary to have an inverter, i.e, DC (Direct

Current) to AC (Alternating Current) converter to convert the available DC source into suitable AC form. The grid-connected inverter for PV system is typically available in different forms according to the power handling capacity and functions. They are popularly found as the following forms: Central inverter, String inverter and the AC module microinverter.

In applications where the power level is under several kilowatts, the single-phase connection is used commonly. However, in such a single-phase connection, the power flow to the grid is time-varying while the power extracted from the PV panel needs to be maintained constant for maximizing the energy harvest. Hence, in single-phase AC power system, energy flow includes both an average power portion that delivers useful energy from the energy source (the PV panel) to the load (the inverter) and a double-frequency portion that flows back and forth between the load and the source. The double frequency portion represents undesirable ripple power that can compromise performance of the DC power source. This necessitates the requirement of energy storage elements between the input and output to decouple the unbalance of power. Mathematically, the instantaneous output power 'Ρ₀' of the inverter in a grid connected single phase system consists of:

an average output power,

Pdc = ^ UI; a d a time varying pulsating power or ripple power,

1

P_ac — ~ UI COS(2wt) , which oscillates at twice the line frequency.

2

'IT and T are the amplitudes of the grid voltage and grid current respectively.

While Pdc > Pac, the surplus energy is stored into the energy storage element, and while Pdc < Pac stored energy is released. This process is known as decoupling.

To serve the purpose of decoupling, a capacitor is commonly used as the energy storage element. However, the lifetime of the capacitor is limited. In "A Review of Power Decoupling Techniques for Micro inverters With Three Different

Decoupling Capacitor Locations in PV Systems," Power Electronics, IEEE Transactions on , vol.28, no.6, pp.271 1 ,2726, June 2013, Batarseh et.al.have reviewed decoupling techniques for micro-inverters with different capacitive methods presently available, and have concluded that the lifetime of the capacitor is the limiting factor.

Studies have also shown that presently used electrolytic capacitors only have a lifetime of around 7000 hours at a temperature of 105Ό (C. C. Dubilier, "Type 381 EL 105 °C ultra- long life snap-in, aluminum."). Due to its rapid degradation characteristics, electrolytic capacitors are sometimes replaced by film capacitors with active decoupling schemes that perform better at higher temperatures and with less degradation. While film capacitors have less degradation and higher ripple current density, they come at significantly higher costs of nearly 3 to 4 times compared to electrolytic capacitors ("Advances in Capacitors and Ultracapacitor for power electronics", IEEE Applied Power Electronics Conference, Industrial Session 1 .3, March 2013). Regardless of whether electrolytic or film capacitors are used as the storage element, their lifetimes are still substantially lower than the typical warranty of 20-25 years of a solar module (20-25 years). Therefore, there exists a need for a circuit and method for mitigating power ripple that seek to address at least one of the above problems.

SUMMARY

In accordance with a first aspect of the present disclosure, there is provided circuit for mitigating power-ripple in a power supplied between a DC element and a power converter connected to a single-phase AC power grid, the circuit capable of being electrically coupled in parallel and between the DC element and the power converter, said circuit comprising an auxiliary circuit comprising an inductor element for providing energy storage; and a control circuit configured to generate one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the DC element and the power converter. The auxiliary circuit may further comprises one or more switching means for switching current flow within the auxiliary circuit to charge or discharge the inductor element, based on the one or more control signals.

The auxiliary circuit may comprise a positive and a negative terminal; a first and a second switching element, each one of the first and second switching elements controlled by respective first and second control signals generated by the control circuit; a first and a second diode; wherein the first switching element, second switching element, first diode, second diode and inductor element arranged in a H-bridge configuration across the positive and negative terminals, such that activating the first and second switching elements by the respective control signals allows a positive voltage to be applied across the inductor, with the current flowing from the positive terminal, through the first switching element, the inductive element, and the second switching element, to the negative terminal; deactivating the first switching element and activating the second switching element allows substantially zero voltage to be applied across the inductor, such that the current free-wheels at the negative terminal, with the current flowing through the first diode, the inductor element and the second switching element respectively; activating the first switching element and deactivating the second switching element allows substantially zero voltage to be applied across the inductor, such that the current free-wheels at the positive terminal, with the current flowing through the second diode, the inductor element and the first switching element respectively; and deactivating the first and second switching elements allows a negative voltage to be applied across the inductor, with the current flowing from the negative terminal, through the first diode, the inductive element, and the second diode, to the positive terminal.

The control circuit may be configured to generate control signals to charge the inductor element when there is excess power in the instantaneous power supplied, and to generate control signals to discharge the inductor element when there is insufficient power in the instantaneous power supplied.

The control circuit may comprise a first current sensor for obtaining a first instantaneous current between the DC element and the power converter, said first current representative of the instantaneous power supplied; and a first filter for filtering the first instantaneous current to obtain a first filtered signal comprising an AC component of the first instantaneous current, said first filtered signal providing a basis for the control circuit to generate the first and second control signals to charge or discharge the inductor. The first filter may be a bandpass filter centered at twice the line frequency of the AC grid.

The control circuit may further comprise a phase detector for identifying a mode of operation based on the phase of the first filtered signal.

The phase detector may comprise an operational amplifier for amplifying and inverting the first filtered signal to produce a first phase output signal identifying a first mode of operation; and an inverter coupled to the output of the operational amplifier for inverting the first output signal to produce a second phase output signal identifying a second mode of operation.

The circuit may further comprise a modulation signal generator for generating a modulation signal based on the first filtered signal. The control circuit may further comprise a second current sensor for obtaining a second instantaneous current, said second instantaneous current representative of instantaneous power supplied at the auxiliary circuit; and a second filter for filtering the second instantaneous current to obtain a second filtered signal comprising an AC component of the second instantaneous current, said second filter is a bandpass filter centered at twice the line frequency of the AC grid; the modulation signal generator further comprising a proportional- integral controller for minimizing errors between the peak values of the first and second filtered signals.

The output of the proportional-integral controller may be multiplied with the modulus of a AC grid voltage phase shifted through 45 degrees to obtain the modulation signal.

The polarity of the phase shift is based on the type of DC element and power converter.

The control unit further comprises a modulation circuit for modulating the first and second phase output with the modulation signal to provide a modulated control signal, said modulated control signal providing the control signal for controlling the first switching element.

The control unit may further comprise a pulse generator for generating a pulsed signal, wherein the modulation circuit is configured to further modulate the first and second phase output signals with the pulsed signal.

The pulse generator may be a saw-tooth generator for generating a saw-tooth waveform.

The first phase output signal may identify a "Boost" mode of operation, and the second phase output signal may identify a "Buck" mode of operation.

The first phase output signal may identify a "Buck" mode of operation, and the second phase output signal may identify a "Boost" mode of operation. The modulation circuit may comprises logic gates configured to receive the modulated control signal, saw-tooth waveform, and first and second phase output signals as input, and to output the first control signal; wherein first switching element is activated by the first control signal when the modulated control signal is greater than the saw-tooth waveform, and the "Buck" mode is identified; or when the modulated control signal is less than the saw-tooth waveform, and the "Boost" mode is identified. The first or second phase output signal may provide the second control signal for controlling the second switching element, wherein the second switching element is activated by the second control signal when "Buck" mode is identified. In accordance with a second aspect of the present disclosure, there is provided a system for mitigating power-ripple, the system comprising a DC power source; a DC to AC converter, with a DC-side of the converter coupled to the DC power source and an AC-side of the converter coupled to an AC power grid; and a circuit electrically coupled in parallel and between the DC power source and the DC to AC converter, wherein the circuit comprises an auxiliary circuit comprising an inductor element for providing energy storage; and a control circuit configured to generate one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the DC source and the DC to AC converter. In accordance with a third aspect of the present disclosure, there is provided a system for mitigating power-ripple, the system comprising a DC load; an AC to DC converter, with a DC-side of the converter coupled to the DC load and an AC-side coupled to an AC power grid; and a circuit electrically coupled in parallel and between the DC load and the AC to DC converter, wherein the circuit comprises an auxiliary circuit comprising an inductor element for providing energy storage; and a control circuit configured to generate one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the AC to DC converter and the DC load.

In accordance with a fourth aspect of the present disclosure, there is provided a method for mitigating power-ripple in a power supplied between a DC element and a power converter connected to a single-phase AC power grid, the method comprising electrically coupling a circuit in parallel and between the DC element and the power converter; providing energy storage with an inductor element in an auxiliary circuit; and generating, with a control unit, one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the DC element and the power converter.

The auxiliary circuit may comprise a positive and a negative terminal; a first and a second switching element; each one of the first and second switching elements controlled by respective first and second control signals generated by the control circuit; a first and a second diode, and the method may further comprise arranging the first switching element, second switching element, first diode, second diode and inductor element arranged in a H-bridge configuration across the positive and negative terminals, such that activating the first and second switching elements by the respective control signals allows a positive voltage to be applied across the inductor, with the current flowing from the positive terminal, through the first switching element, the inductive element, and the second switching element, to the negative terminal; deactivating the first switching element and activating the second switching element allows substantially zero voltage to be applied across the inductor, such that the current freewheels at the negative terminal, with the current flowing through the first diode, the inductor element and the second switching element respectively; activating the first switching element and deactivating the second switching element allows substantially zero voltage to be applied across the inductor, such that the current free-wheels at the positive terminal, with the current flowing through the second diode, the inductor element and the first switching element respectively; and deactivating the first and second switching elements allows a negative voltage to be applied across the inductor, with the current flowing from the negative terminal, through the first diode, the inductive element, and the second diode, to the positive terminal.

The method may comprise obtaining, with a first current sensor, a first instantaneous current between the DC element and the power converter, said first instantaneous current representative of the instantaneous power supplied; and filtering, with a bandpass filter, the first instantaneous current to obtain a first filtered signal comprising an AC component of the first instantaneous current, said first filtered signal providing a basis for the control circuit to generate the first and second control signals to charge or discharge the inductor.

BRIEF DESCRIPTION OF THE DRAWINGS

Exemplary embodiments of the invention will be better understood and readily apparent to one of ordinary skill in the art from the following written description, by way of example only, and in conjunction with the drawings, in which:

FIG. 1 is schematic view of a grid-connected photovoltaic system in an exemplary embodiment.

FIG. 2 is schematic view of an auxiliary circuit in an exemplary embodiment. FIG. 3 show exemplary waveforms of "Buck" and "Boost" modes of operation in an example embodiment.

FIG. 4 shows a schematic view of a control circuit in an exemplary embodiment.

FIG. 5 is a schematic view of a filter in an exemplary embodiment.

FIG. 6 is a schematic view of a phase detector circuit in an exemplary embodiment. FIG. 7 is a schematic view of a modulation signal generator circuit in an exemplary embodiment.

FIG. 8 is a schematic view of a pulse generator in an exemplary embodiment. FIG. 9 is a schematic view of a modulation circuit in an exemplary embodiment.

FIG. 10 show simulated output current and voltage of a PV panel in an exemplary embodiment. FIG. 1 1 show simulated signal waveforms of the auxiliary inductor current, the 2^nd harmonic AC current of the DC-AC converter, the Boost and the Buck signals, in the example embodiment.

FIG. 12 is schematic view of a grid-connected system in an exemplary embodiment.

FIG. 13 is schematic view of an auxiliary circuit in an exemplary embodiment.

FIG. 14 shows a schematic view of a control circuit in an exemplary embodiment. FIG. 15 is a schematic view of a filter in an exemplary embodiment.

FIG. 16 is a schematic view of a phase detector circuit in an exemplary embodiment.

FIG. 17 is a schematic view of a modulation signal generator circuit in an exemplary embodiment. FIG. 18 is a schematic view of a pulse generator in an exemplary embodiment.

FIG. 19 is a schematic view of a modulation circuit in an exemplary embodiment. FIG. 20 is a simulated waveform showing the rise and fall in inductor current in an exemplary embodiment.

FIG. 21 shows the simulated waveforms for control signals in an exemplary embodiment.

FIG. 22 shows the simulated waveforms for a modulation signal and an output waveform.

FIG. 23 is a schematic flowchart for illustrating a method of mitigating a power ripple in a power supplied between a DC element and a power converter connected to a single- phase AC power grid, in an exemplary embodiment.

DETAILED DESCRIPTION

Exemplary embodiments described herein are applicable to the field of electrical power systems and may provide an electrical circuit for mitigating power-ripple in a power supplied between a DC element and a power converter connected to a single-phase AC power grid. The exemplary circuits provide for an inductor, with associated controls, to be used as an energy storage element to take care of the AC power ripple at the DC side of the DC to AC converter. Using the inductor as the storage element can dramatically increase the lifetime of the system where power-ripple mitigation is required. In particular, using the inductor instead of the capacitor can improve the degradation performance of an overall system, such as e.g., a Photovoltaic source (or other DC Source) providing power to a DC-AC converter (inverter) interfacing an AC grid.

In an example embodiment, the DC element may be a DC source (such as a Photovoltaic source) providing power to the power converter, where the power converter is in the form of a DC to AC converter, connected to a single-phase AC power grid. In an alternative embodiment, the DC element may be a DC load receiving power from the power converter, where the power converter is in the form of an AC to DC converter, connected to a single- phase AC power grid.

FIG. 1 is schematic view of a grid-connected photovoltaic system 100 in an exemplary embodiment. In the exemplary embodiment, the system 100 comprises a DC source 102 in the form of the photovoltaic panel which is electrically connected to an inverter, i.e., the DC to AC Converter 104 interfacing with an AC utility grid 106. The source 102 is connected to the inverter 104 via a positive rail 108 and negative rail 1 10. It will be appreciated that the electrical connection 108 experiences a second harmonic power ripple (i.e., a power ripple at twice the frequency of the grid 106) as a result of the connection with the grid 106. Further, while the example embodiment shows a photovoltaic panel as the DC source, it will be appreciated that any other form of DC source is similarly applicable.

FIG. 2 is schematic view of an auxiliary circuit 200 in an exemplary embodiment. In the example embodiment, the auxiliary circuit 200 is electrically coupled or connected between the DC source 102 and the DC to AC converter 104 (FIG. 1 ). The auxiliary circuit 200 comprises an inductor element 202 for providing energy storage. The auxiliary circuit 200 further comprises switching elements or means for switching current flow within the auxiliary circuit to charge or discharge the inductor element 202, based on one or more input control signals.

In the example embodiment, the auxiliary circuit comprises first and second active MOSFET switches S1 and S2 (numerals 204 and 206 in FIG. 2), first and second diodes D1 and D2 (numerals 208 and 210 in FIG. 2). The first switch 204 is controlled by a first input control signal (Gup) 212 and the second switch 206 is controlled by a second input control signal (Buck) 214. In FIG. 2, the inductor element 210 is represented with a series resistance Ri_, and an inductive reactance Xi_ (where Xi_ = 2^*pi^*L and L is the inductance).

In the example embodiment, the auxiliary circuit 200 also comprises a positive terminal 216 and a negative terminal 218 for connecting the auxiliary circuit 200 to the positive and negative rails 108, 1 10 of the system 100 (See FIG. 1 ) respectively.

In the example embodiment, the switches 204, 206, diodes 208, 208 and inductor element 202 are connected or arranged in an H-bridge configuration. Based on a charging or discharging mode determined by a control unit (not shown), the control unit is configured to generate the input control signals 212, 214 accordingly to control (i.e., activate or deactivate) the switches 204, 206 respectively, such that current flow is controlled or directed within the auxiliary circuit 200 to charge or discharge the inductor element 202.

The operation of the auxiliary circuit 200 in the example embodiment is divided into a "Buck" mode and a "Boost" mode. In the "Buck" mode, excess ripple energy is determined based on the instantaneous power supplied between the panel 102 and the inverter 104 at the positive rail 108 (FIG. 1 ) and the excess energy is to be stored inside the inductor element 202. In the "Boost" mode, insufficient energy is determined based on the instantaneous power supplied between the panel 102 and the inverter 104 at the positive rail 108 (FIG. 1 ) and energy stored in the inductor element 202 is to be extracted from the inductor to boost the power supplied in the positive rail 108 (FIG. 1 ).

In the example embodiment, during "Buck" (or energy storage) mode, the second switch 206 is always turned on. During "Buck" mode operation, operation of the auxiliary circuit is further subdivided into two further modes, "Buck mode 1 " and "Buck mode 2", based on the state of the first switch 204.

During "Buck mode 1 ", both the first switch 204 and the second switch 206 are turned on (i.e., activated) according to the control signals 'Gup' and 'Buck' respectively described in more detail below. By doing so, a positive voltage V_dc at the output of the PV panel source 102 (FIG. 1 ) is reflected across the inductor 202 (FIG. 2) so that the current through the inductor will rise, thus storing energy in the inductor 202. The storing of energy in the inductor 202 is also referred to as charging the inductor 202. During "Buck mode 2", the first switch 204 is turned off (i.e., deactivated), but the second switch 206 remains on (i.e., activated). By doing so, the current in the inductor 202 freewheels through the first diode 208 and the second switch 206, with ideally substantially zero voltage across the inductor 202. The current through the inductor 202 remains constant during this mode. "Buck mode 2" ends when the first switch 204 is turned on again, with the circuit returning to "Buck mode 1 ", signifying the completion of one switching cycle in the "Buck" mode.

In the example embodiment, during "Boost" (or energy release) mode, the second switch 206 is always turned off. The "Boost" mode is further subdivided into two further modes, "Boost mode 1 " and "Boost mode 2", based on the state of the first switch 204. During "Boost mode 1 ", the first switch 204 is turned on (i.e., activated) and the second switch 206 is turned off (i.e., deactivated) again according to the control signals 'Gup' and 'Buck' respectively. By doing so, the current in the inductor 202 freewheels through the second diode 210 and the first switch 204, with ideally substantially zero voltage across the inductor 202. The current through the inductor 202 remains constant during this mode.

During "Boost mode 2", the first switch 204 is turned off (i.e., deactivated) with the second switch 206 remaining turned off (i.e., deactivated). By doing so, the current in the inductor 202 flows through the second diode 210 and the first diode 208 so that ideally -V_dc is incident across the inductor, such that the current through the inductor will ramp down and the energy stored in the inductor 202 is released. The release of energy from the inductor 202 is also be referred to as discharging the inductor 202.

"Boost mode 2" ends when the first switch 204 is turned on again, with the circuit returning to "Boost mode 1 ", signifying the completion of one switching cycle in the "Boost" mode.

FIG. 3 show exemplary waveforms of "Buck" and "Boost" modes of operation in the example embodiment. In FIG. 3, "Buck mode 1 " is represented in the period between t0< t < t1 , while "Buck mode 2" is represented in the period between t1 < t < t2. "Boost mode 1 " is represented in the period between t'0< t < t'1 , while "Buck mode 2" is represented in the period between t'1 < t < t'2.

As mentioned above, the input control signals 212, 214 for controlling the switches 204, 206 such that current flow is controlled or directed within the auxiliary circuit 200 to charge or discharge the inductor element 202 (FIG. 2), is generated by a control circuit.

FIG. 4 shows a schematic block diagram of a control circuit 400 in an exemplary embodiment. The control circuit 400 comprises a first and a second current sensor 402, 404, a filter 406, a phase detector circuit 408, a modulation signal generator circuit 410, a pulse generator 412, and a modulation circuit 414. The control circuit 400 is configured to generate first and second input control signals for controlling control switches 204, 206 (FIG. 2) based on instantaneous currents obtained by the first and second current sensors 402, 404.

The first current sensor 402 is configured to obtain an instantaneous current l_ref at the positive rail 108 (FIG. 1 ), between the DC source 102 and the DC to AC converter 104. The instantaneous current is representative or indicative of the instantaneous power supplied between the DC source 102 and the DC to AC converter. The second current sensor 404 is configured to obtain an instantaneous current x flowing through the auxiliary circuit 200 at the positive terminal 216 of the auxiliary circuit 200. The placement of the current sensors 402 and 404 are also shown in FIG. 1 and FIG. 2 respectively.

FIG. 5 is a schematic view of the filter 406 in an exemplary embodiment. In the example embodiment, the control circuit 400 (FIG. 4) is configured to ensure that the second harmonic current component required by the DC-AC converter 104 (FIG. 1 ) is provided for by the auxiliary circuit 200 (FIG. 2). Thus, the filter 406 comprises bandpass filters 502 and 504, each respectively providing bandpass filtering at a center frequency which matches the 2^nd harmonic frequency of the utility AC grid 106 (FIG. 1 ). The bandpass filters filter the instantaneous current l_ref and x to provide filtered signals l_ref² and l_aUx² respectively. The filtered signals l_ref² and l_aUx² consist of the AC components of the instantaneous current l_ref and laux respectively. The first and second filtered signals l_ref² and x² provides a basis for the control circuit 400 to generate the first and second control signals 212, 214 (FIG. 2) to charge or discharge the inductor 202 (FIG. 2).

FIG. 6 is a schematic view of the phase detector circuit 408 in an exemplary embodiment. In the example embodiment, the phase detector circuit 408 is configured to determine and control the required mode of operation of the auxiliary circuit 200, based on the filtered instantaneous current l_ref². The phase detector circuit 408 comprises an operational amplifier 602 for amplifying and inverting the filtered instantaneous current l_ref² to produce a first phase output signal 606. The phase detector circuit 408 further comprises an inverter 604 coupled to the output of the operational amplifier 602 to produce a second phase output signal 608. The first phase output signal 606 identifies a "Boost" operation mode, while the second phase output signal 608 identifies a "Buck" operation mode, as previously described.

In the example embodiment, when p_ac>Pdc, i.e., when the DC to AC converter 104 (FIG. 1 ) requires more energy than what is provided by the DC source 102 (FIG. 1 ), the inductor 202 (FIG. 2) is configured to discharge to help meet the load demand, hence the "Boost" operation mode. When p_ac < Pdc, i.e., when the DC to AC converter 104 (FIG. 1 ) requires less energy than what is provided by the DC source 102 (FIG. 1 ), the inductor 202 (FIG. 2) is configured to charge and store the excess energy, hence the "Buck" operation mode. FIG. 7 is a schematic view the modulation signal generator circuit 410 in an exemplary embodiment. In the example embodiment, the modulation signal generator circuit 410 comprises a pair of peak current converters 702, a controller 704, and a multiplication circuit 706. In the example embodiment, the controller 704 may preferably be a proportional integral (PI) controller although alternative controllers may also be used. The peak current converters 702 obtain the peak values of the instantaneous currents x² and l_ref² as l_aUx^2p and l_ref^2p respectively. Thereafter, the PI controller 704 is used to minimise the error between the peak values l_aUx^2p and l_ref^2p and to provide for closed loop operations. The output 708 of the PI controller is multiplied with the modulus of the grid voltage, V_grid , phase-shifted by a value of (-) 45 degrees to obtain the modulation signal "Ctrl" (reference numeral 710). The phase-shift value is not limited to (-) 45 degrees. In alternative embodiments, the output 708 of the PI controller may be multiplied with the modulus of the grid voltage, V_grid, phase-shifted by alternative values, preferably between (-) 45 degrees to (-) 55 degrees. FIG. 8 is a schematic view of the pulse generator 412 in an exemplary embodiment. In the example embodiment, the pulse or carrier wave generator 412 is preferably a saw tooth pattern generator for generating a saw tooth waveform "ST" (reference numeral 802) having a constant amplitude and constant frequency as an output. FIG. 9 is a schematic view of the modulation circuit 414 in an exemplary embodiment.

In the example embodiment, the modulation circuit 414 comprises a plurality of logic gates to output a logic signal "Gup" (reference numeral 902). The modulation circuit 414 receives, as input signals, the first and second phase output signals "Boost" 606 and "Buck" 608 from the phase detector circuit 408 (FIG. 6), the modulation signal "Ctrl" 710 from the modulation signal generator circuit 410 (FIG. 7), and the saw-tooth waveform "ST" 802 from the pulse generator 412 (FIG. 8). Based on these input signals the logic gates generates the output logic signal "Gup" 902 as a gating pulse, when:

- the modulation signal "Ctrl" 710 is greater than the saw-tooth waveform "ST" 802, and the "Buck" 608 mode is identified; OR

- the modulation signal "Ctrl" is less than the saw-tooth waveform "ST" 802, and the

"Boost" 606 mode is identified.

The output logic signal "Gup" 902 is coupled to and provides for the first control signal 212 (FIG. 2) which controls the first switch 204 (FIG. 2), while the second phase output logic signal "Buck" 608 provides the second control signal 214 (FIG. 2) which controls the switch 206 (FIG. 2). The modulation logic for "Gup" 212 (Fig. 2) is required for controlling the amount of charging and discharging of the inductor 202 (FIG. 2) while in the "Buck" and "Boost" modes respectively.

Simulation

An example implementation was simulated on a 250W microinverter powered by a PV panel with 45V, 6.25A at the maximum power point tracking (MPPT) voltage and current. The value of the inductor used is 6mH. FIG. 10 shows the simulated output current and voltage of the PV panel in the example embodiment.

FIG. 1 1 shows the simulated signal waveforms of the auxiliary inductor current, the 2^nd harmonic AC current of the DC-AC converter, the Boost and the Buck signals, in the example embodiment. It will be appreciated that both the Boost and Buck modes are respectively dependent on the status of the ripple. According to the phase status of the ripple inherited by the "Ctrl" signal, the switch S1 (e.g., reference numeral 204 in FIG. 2) is modulated with a sawtooth carrier for controlling the charging and discharging of the inductor. As mentioned above, a need for power electronics components with longer lifetime is the primary concern in solar PV installations. In particular, PV installations are required to last for 20-25 years. However, presently used electrolytic capacitors have a life time of only around 1000 - 7000 hours at a temperature of 105 degrees Celsius. While film capacitors degrade less to high ripple current density than electrolytic capacitors, they are however nearly 3 to 4 times as costly as electrolytic capacitors.

Example embodiments of the present application provide a purely inductive decoupling scheme, which seeks to address reliability concerns and lifetime degradation issues of a solar PV system. The example embodiments of the present application can advantageously improve the reliability of the DC-AC converter without incurring significant cost and complexity, when compared with existing means of active power decoupling schemes.

Moreover, it will be appreciated that the inductive storage elements in the example embodiments may advantageously be more compact than electrolytic capacitors and more cost effective than film capacitors. As the design of the inductor may be dependent on the power level and the line frequency for optimum performance, an optimised design approach and appropriate material selection may be required to achieve increased compactness and cost-effectiveness. It will be appreciated that as the example embodiments are easily scalable, they may be advantageously applicable in a variety of inverters including, but not limited to, central inverters, string inverters and AC module Microinverters.

It will be appreciated that while the example embodiments described above recite PV sources as the DC source, the present disclosure is not limited to such sources. The example embodiments are similarly applicable to other DC sources where appropriate.

While the example embodiments define a particular control circuit comprising e.g., the use of a saw-tooth carrier, mode identification methods, etc., it will be appreciated that alternative circuit arrangements may be implanted to achieve the same effect of charging and discharging an inductive element according to the instantaneous power supplied between the source and the inverter, as disclosed above. As an example, the diodes in the auxiliary circuit as presented above may be replaceable by active switches in the form of e.g., MOSFETs, with additional circuitry which may be easily implemented by a person skilled in the art, in view of the present disclosure.

MITIGATION OF POWER RIPPLE IN AC INPUT AND DC OUTPUT ENVIRONMENT

The example embodiments described above are directed at mitigating power-ripple in a power supplied from a DC supply to a DC to AC converter. It has been further recognised that a similar concept for power-ripple mitigation may also be applied to a system where an AC to DC converter supplies power to a DC load, but with a minor variation.

As previously mentioned, mathematically, the instantaneous output power 'Ρ₀' of the inverter/converter in a grid connected single phase system consists of:

an average output power, P_dc = ^ UI; and

a time varying pulsating power, P_ac = ui cos(2wt) , which oscillates at twice the line frequency,

where 'U' and T are the amplitudes of the grid voltage and grid current respectively. Previously, for a DC source connected to a DC to AC converter interfacing the AC grid, when Pdc > P_ac, the surplus energy is stored into the energy storage element. This stored energy is released when Pdc < Pac- However, for an embodiment where the system comprises a DC load connected to a AC to DC converter interfacing the AC grid, it will be appreciated that surplus energy exists when Pdc < Pac- Thus, surplus energy is stored in the energy storage when Pdc < Paci and stored energy is released when Pdc > Pac- FIG. 12 is schematic view of a grid-connected system 1200 in an exemplary embodiment. In the exemplary embodiment, the system 1200 comprises an AC to DC Converter 1202 for interfacing an AC utility grid 1204, and a DC Load 1206 electrically connected to the AC to DC Converter 1202. The load 1206 is connected to the AC to DC converter 1202 via a positive rail 1208 and negative rail 1210. It will be appreciated that the electrical connection 1208 experiences a second harmonic power ripple (i.e., a power ripple at twice the frequency of the grid 1204) as a result of the connection with the grid 1204.

FIG. 13 is schematic view of an auxiliary circuit 1300 in an exemplary embodiment. In the example embodiment, the auxiliary circuit 1300 is electrically coupled or connected between the DC load 1206 and the AC to DC converter 1202 (FIG. 12). The auxiliary circuit 1300 comprises an inductor element 1302 for providing energy storage. The auxiliary circuit 1300 further comprises switching elements or means for switching current flow within the auxiliary circuit to charge or discharge the inductor element 1302, based on one or more input control signals.

In the example embodiment, the auxiliary circuit comprises first and second active MOSFET switches S1 and S2 (numerals 1304 and 1306 in FIG. 13), first and second diodes D1 and D2 (numerals 1308 and 1310 in FIG. 13). The first switch 1304 is controlled by a first input control signal 1312 and the second switch 1306 is controlled by a second input control signal 1314. In FIG. 13, the inductor element 1310 is represented with a series resistance Ri_, and an inductive reactance Xi_ (where Xi_ = 2^*pi^*L and L is the inductance).

In the example embodiment, the auxiliary circuit 1300 also comprises a positive terminal 1316 and a negative terminal 1318 for connecting the auxiliary circuit 1300 to the positive and negative rails 1208, 1210 of the system 1200 (See FIG. 12) respectively. In the example embodiment, the switches 1304, 1306, diodes 1308, 1308 and inductor element 1302 are connected or arranged in an H-bridge configuration. Based on a charging or discharging mode determined by a control unit (e.g., numeral 1400 in Fig. 14), the control unit is configured to generate input control signals 1312, 1314 accordingly to control the switches 1304, 1306 such that current flow is controlled or directed within the auxiliary circuit to charge or discharge the inductor element 1302.

The operation of the auxiliary circuit 1300 in the example embodiment is divided into a "Buck" mode and a "Boost" mode. In the "Buck" mode, excess ripple energy is determined based on the instantaneous power supplied between the AC to DC converter 1202 and the load 1206 at the positive rail 1208 (FIG. 12) and the excess energy is to be stored inside the inductor element 1302. In the "Boost" mode, insufficient energy is determined based on the instantaneous power supplied between the AC to DC converter 1202 and the load 1206 at the positive rail 1208 (FIG. 12) and energy stored in the inductor element 1302 is to be extracted from the inductor to boost the power supplied in the positive rail 1208 (FIG. 12).

In the example embodiment, during "Buck" mode, the second switch 1306 is always turned on. During "Buck" mode operation, operation of the auxiliary circuit is further subdivided into two further modes, "Buck mode 1 " and "Buck mode 2", based on the state of the first switch 1304.

During "Buck mode 1 ", both the first switch 1304 and second switch 1306 are turned on (i.e., activated). By doing so, a positive voltage V_dc at the output of the AC to DC converter 1202 (FIG. 12) is reflected across the inductor 1302 (FIG. 13) so that the current through the inductor will rise, thus storing energy in the inductor 1302. The storing of energy in the inductor 1302 is also referred to as charging the inductor 1302.

During "Buck mode 2", the first switch 1304 is turned off (i.e., deactivated), but the second switch 1306 remains on (i.e., activated). By doing so, the current in the inductor 1302 freewheels through the first diode 1308 and the second switch 1306, with ideally substantially zero voltage across the inductor 1302. The current through the inductor 1302 remains constant during this mode. "Buck mode 2" ends when the first switch 1304 is turned on again, with the circuit returning to "Buck mode 1 ", signifying the completion of one switching cycle in the "Buck" mode. In the example embodiment, during "Boost" (or energy release) mode, the second switch 1306 is always turned off. The "Boost" mode is further subdivided into two further modes, "Boost mode 1 " and "Boost mode 2", based on the state of the first switch 1304. During "Boost mode 1 ", the first switch 1304 is turned on (i.e., activated) and the second switch 1306 is turned off (i.e., deactivated). By doing so, the current in the inductor 1302 freewheels through the second diode 1310 and the first switch 1304, with ideally substantially zero voltage across the inductor 1302. The current through the inductor remains constant during this mode 1302.

During "Boost mode 2", the first switch 1304 is turned off (i.e., deactivated) with the second switch 1306 remaining off (i.e., deactivated). By doing so, the current in the inductor 1302 flows through the second diode 1310 and the first diode 1308 so that ideally -V_dc is incident across the inductor, such that the current through the inductor will ramp down and the energy stored in the inductor is released.

"Boost mode 2" ends when the first switch 1304 is turned on again, with the circuit returning to "Boost mode 1 ", signifying the completion of one switching cycle in the "Boost" mode.

As mentioned above, the input control signals 1312, 1314 for controlling the switches 1304, 1306 such that current flow is controlled within the auxiliary circuit 1300 to charge or discharge the inductor element 1302 (FIG. 13), is generated by a control circuit. FIG. 14 shows a schematic block diagram of a control circuit 1400 in an exemplary embodiment. The control circuit 1400 comprises a first and a second current sensor 1402, 1404, a filter 1406, a phase detector circuit 1408, a modulation signal generator circuit 1410, a pulse generator 1412, and modulation circuit 1414. The control circuit 1400 is configured to generate first and second input control signals for controlling control switches 1304, 1306 (FIG. 13) based on instantaneous currents obtained by the first and second current sensors 1402, 1404.

The first current sensor 1402 is configured to obtain an instantaneous current l_ref at the positive rail 1208 (FIG. 12), between the AC to DC converter 1202 and the DC load 1206. The instantaneous current is representative or indicative of the instantaneous power supplied between the AC to DC Converter 1202 and the load 1206. The second current sensor 1404 is configured to obtain an instantaneous current x flowing through the auxiliary circuit 1300 at the positive terminal 1316 of the auxiliary circuit 1300. The placement of the current sensors 1402 and 1404 are also shown in FIG. 12 and FIG. 13 respectively. FIG. 15 is a schematic view of the filter 1406 in an exemplary embodiment. In the example embodiment, the control circuit 1400 (FIG. 14) is configured to ensure that the second harmonic current required by the AC-DC converter 1304 (FIG. 13) is provided for by the auxiliary circuit 1302 (FIG. 13). Thus, the filter 1406 comprises bandpass filters 1502 and 1504, each respectively providing bandpass filtering at a center frequency which matches the 2^nd harmonic frequency of the utility AC grid 1206 (FIG. 12). The bandpass filters filter the instantaneous current l_ref and x to provide filtered signals l_ref² and l_aUx² respectively. The filtered signals l_ref² and l_aUx² consist of the AC components of the instantaneous current l_ref and laux respectively. The first and second filtered signals l_ref² and x² provides a basis for the control circuit 1400 to generate the first and second control signals 1312, 1314 (FIG. 13) to charge or discharge the inductor 1302 (FIG. 13).

FIG. 16 is a schematic view of the phase detector circuit 1408 in an exemplary embodiment. In the example embodiment, the phase detector circuit 1408 is configured to determine and control the required mode of operation of the auxiliary circuit 1300, based on the filtered instantaneous current l_ref². The phase detector circuit 1408 comprises an operational amplifier 1602 for amplifying and inverting the filtered instantaneous current l_ref²to produce a first phase output signal 1606. The phase detector circuit 1408 further comprises an inverter 1604 coupled to the output of the operational amplifier 1602 to produce a second phase output signal 1608. The first phase output signal 1606 identifies a "Buck" operation mode, while the second phase output signal 1608 identifies a "Boost" operation mode, as previously described.

In the example embodiment, when p_ac<Pdc, i.e., when the AC to DC converter 1202 (FIG. 12) provides less energy than what is required by the DC load 1206 (FIG. 12), the inductor 1302 (FIG. 13) is configured to discharge to help meet the load demand, hence the "Boost" operation mode. When p_ac > Pdc, i.e., when the AC to DC converter 1202 (FIG. 12) provides more energy than what is required by the DC load 1206 (FIG. 12), the inductor 1302 (FIG. 13) is configured to charge and store the excess energy, hence the "Buck" operation mode. FIG. 17 is a schematic view a modulation signal generator circuit 1410 in an exemplary embodiment. In the example embodiment, the modulation signal generator circuit 1410 comprises a pair of peak current converters 1702, a controller 1704, and a multiplication circuit 1706. In the example embodiment, the controller 1704 may preferably be a proportional integral (PI) controller although alternative controllers may also be used. The peak current converters 1702 obtain the peak values of the instantaneous currents l_aUx² and l_ref² as l_aUx^2p and l_ref^2p respectively. Thereafter, the PI controller 1704 is used to minimise the error between the peak values l_aUx^2p and l_ref^2p and to provide for closed loop operations. The output 1708 of the PI controller is multiplied with the modulus of the grid voltage, V_grid, phase-shifted by a value of (+) 45 degrees to obtain the modulation signal "Ctrl" (reference numeral 1710). The phase-shift value is not limited to (+) 45 degrees. In alternative embodiments, the output 1708 of the PI controller may be multiplied with the modulus of the grid voltage, V_grid, phase-shifted by alternative values, preferably between (+) 45 degrees to (+) 55 degrees.

FIG. 18 is a schematic view of a pulse generator 1412 in an exemplary embodiment. In the example embodiment, the pulse generator 1412 is a saw tooth generator for generating a saw tooth waveform "ST" (reference numeral 1802) having a constant amplitude and constant frequency as an output.

FIG. 19 is a schematic view of the modulation circuit 1414 in an exemplary embodiment. In the example embodiment, the modulation circuit 1414 comprises a plurality of logic gates to output a logic signal "Gup" (reference numeral 1902). The modulation circuit 1414 receives, as input signals, the first and second phase output signals "Buck" 1606 and "Boost" 608 from the phase detector circuit 1408 (FIG. 16), the modulation signal "Ctrl" 1710 from the modulation signal generator circuit 1410 (FIG. 17), and the saw-tooth waveform "ST" 1802 from the pulse generator 1412 (FIG. 18). Based on these input signals the logic gates generates the output logic signal "Gup" 1902 as a gating pulse, when:

- the modulation signal "Ctrl" 1710 is greater than the saw-tooth waveform "ST" 1802, and the "Buck" 1606 mode is identified; OR

the modulation signal "Ctrl" is less than the saw-tooth waveform "ST" 1802, and the "Boost" 1608 mode is identified.

The output logic signal "Gup" 1902 is coupled to and provides the first control signal 1312 (FIG. 13) which controls the switch 1304 (FIG. 13), while the first phase output logic signal "Buck" 1606 is coupled to and provides the second control signal 1314 (FIG. 13) which controls the switch 1306 (FIG. 13). The modulation logic for "Gup" 1312 (FIG. 13) is required for controlling the amount of charging and discharging of the inductor 1302 (FIG. 13) while in the "Buck" and "Boost" modes respectively. An example implementation was simulated to provide 120W output power from the AC to DC converter to a DC load. FIG. 20 is a simulated waveform showing the rise and fall in inductor current in the exemplary embodiment. The charging and discharging of the inductor for ripple mitigation is illustrated. FIG. 21 shows the simulated waveforms for control signals, Iref2, Iaux2, Ctrl, Buck and

Boost, in the exemplary embodiment. FIG. 22 shows the simulated waveforms for the modulation signal ST and the output waveform, Gup.

FIG. 23 is a schematic flowchart 2300 for illustrating a method for mitigating power-ripple in a power supplied between a DC element and a power converter connected to a single- phase AC power grid, in an example embodiment. At step 2302, a circuit is electrically coupled in parallel and between the DC element and the power converter. At step 2304, energy storage is provided with an inductor element in an auxiliary circuit. At step 2306, one or more control signals to charge or discharge the inductor element is generated with a control unit, based on an instantaneous power supplied between the DC element and the power converter.

Example embodiments of the present application can also advantageously improve the reliability and lifetimes of AC-DC converter systems powering a DC load. As mentioned above, presently used electrolytic capacitors have a life time of only around 1000 - 7000 hours at a temperature of 105 degrees Celsius. Also, while film capacitors degrade less to high ripple current density than electrolytic capacitors, they are however nearly 3 to 4 times as costly as electrolytic capacitors.

It will be appreciated that the inductive storage elements in the example embodiments may advantageously be more compact than electrolytic capacitors and more cost effective than film capacitors. However, because the design of the inductor may be dependent on the power level and the line frequency for optimum performance, an optimised design approach and appropriate material selection may be required to achieve increased compactness and cost-effectiveness. It will be appreciated that as the example embodiments are scalable, they may be advantageously applicable in a variety of converters including AC-DC converters. These converters can be applied in devices which work in high temperature environments such as in e.g., Street lighting purposes.

While the example embodiments define a particular control circuit comprising e.g., the use of a saw-tooth carrier, mode identification methods, etc., it will be appreciated that alternative circuit arrangements may be implanted to achieve the same logic as disclosed above. As an example, the diodes in the auxiliary circuit as presented above may be replaceable by active switches in the form of e.g., MOSFETs, with additional circuitry which may be easily implemented by a person skilled in the art, in view of the present disclosure.

The terms "coupled" or "connected" as used in this description are intended to cover both directly connected or connected through one or more intermediate means, unless otherwise stated.

Additionally, when describing some embodiments, the disclosure may have disclosed a method and/or process as a particular sequence of steps. However, unless otherwise required, it will be appreciated the method or process should not be limited to the particular sequence of steps disclosed. Other sequences of steps may be possible. The particular order of the steps disclosed herein should not be construed as undue limitations. Unless otherwise required, a method and/or process disclosed herein should not be limited to the steps being carried out in the order written. The sequence of steps may be varied and still remain within the scope of the disclosure.

Further, in the description herein, the word "substantially" whenever used is understood to include, but not restricted to, "entirely" or "completely" and the like. In addition, terms such as "comprising", "comprise", and the like whenever used, are intended to be non-restricting descriptive language in that they broadly include elements/components recited after such terms, in addition to other components not explicitly recited. Further, terms such as "about", "approximately" and the like whenever used, typically means a reasonable variation, for example a variation of +/- 5% of the disclosed value, or a variance of 4% of the disclosed value, or a variance of 3% of the disclosed value, a variance of 2% of the disclosed value or a variance of 1 % of the disclosed value. Furthermore, in the description herein, certain values may be disclosed in a range. The values showing the end points of a range are intended to illustrate a preferred range. Whenever a range has been described, it is intended that the range covers and teaches all possible sub-ranges as well as individual numerical values within that range. That is, the end points of a range should not be interpreted as inflexible limitations. For example, a description of a range of 1 % to 5% is intended to have specifically disclosed sub-ranges 1 % to 2%, 1 % to 3%, 1 % to 4%, 2% to 3% etc., as well as individually, values within that range such as 1 %, 2%, 3%, 4% and 5%. The intention of the above specific disclosure is applicable to any depth/breadth of a range.

It will be appreciated by a person skilled in the art that other variations and/or modifications may be made to the specific embodiments without departing from the scope of the invention as broadly described. The present embodiments are, therefore, to be considered in all respects to be illustrative and not restrictive.

Claims

1 . A circuit for mitigating power-ripple in a power supplied between a DC element and a power converter connected to a single-phase AC power grid, the circuit capable of being electrically coupled in parallel and between the DC element and the power converter, said circuit comprising

an auxiliary circuit comprising an inductor element for providing energy storage; and a control circuit configured to generate one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the DC element and the power converter.

2. The circuit as claimed in claim 1 , wherein the auxiliary circuit further comprises one or more switching means for switching current flow within the auxiliary circuit to charge or discharge the inductor element, based on the one or more control signals.

3. The circuit as claimed in claim 1 or 2, wherein the auxiliary circuit comprises

a positive and a negative terminal;

a first and a second switching element, each one of the first and second switching elements controlled by respective first and second control signals generated by the control circuit;

a first and a second diode;

wherein the first switching element, second switching element, first diode, second diode and inductor element arranged in a H-bridge configuration across the positive and negative terminals, such that

activating the first and second switching elements by the respective control signals allows a positive voltage to be applied across the inductor, with the current flowing from the positive terminal, through the first switching element, the inductive element, and the second switching element, to the negative terminal;

deactivating the first switching element and activating the second switching element allows substantially zero voltage to be applied across the inductor, such that the current freewheels at the negative terminal, with the current flowing through the first diode, the inductor element and the second switching element respectively;

activating the first switching element and deactivating the second switching element allows substantially zero voltage to be applied across the inductor, such that the current free- wheels at the positive terminal, with the current flowing through the second diode, the inductor element and the first switching element respectively; and

deactivating the first and second switching elements allows a negative voltage to be applied across the inductor, with the current flowing from the negative terminal, through the first diode, the inductive element, and the second diode, to the positive terminal.

4. The circuit as claimed in any one of the preceding claims, wherein the control circuit is configured to generate control signals to charge the inductor element when there is excess power in the instantaneous power supplied, and to generate control signals to discharge the inductor element when there is insufficient power in the instantaneous power supplied.

5. The circuit as claimed in claims 3 or 4, the control circuit comprising

a first current sensor for obtaining a first instantaneous current between the DC element and the power converter, said first current representative of the instantaneous power supplied; and

a first filter for filtering the first instantaneous current to obtain a first filtered signal comprising an AC component of the first instantaneous current, said first filtered signal providing a basis for the control circuit to generate the first and second control signals to charge or discharge the inductor.

6. The circuit as claimed in claim 5, wherein the first filter is a bandpass filter centered at twice the line frequency of the AC grid.

7. The circuit as claimed in claim 5 or 6, the control circuit further comprising a phase detector for identifying a mode of operation based on the phase of the first filtered signal.

8. The circuit as claimed in claim 7, wherein the phase detector comprises

an operational amplifier for amplifying and inverting the first filtered signal to produce a first phase output signal identifying a first mode of operation; and

an inverter coupled to the output of the operational amplifier for inverting the first output signal to produce a second phase output signal identifying a second mode of operation.

9. The circuit as claimed in claim 8, further comprising a modulation signal generator for generating a modulation signal based on the first filtered signal.

10. The circuit as claimed in claim 9, wherein the control circuit further comprises a second current sensor for obtaining a second instantaneous current, said second instantaneous current representative of instantaneous power supplied at the auxiliary circuit; and

a second filter for filtering the second instantaneous current to obtain a second filtered signal comprising an AC component of the second instantaneous current, said second filter is a bandpass filter centered at twice the line frequency of the AC grid;

the modulation signal generator further comprising

a proportional-integral controller for minimizing errors between the peak values of the first and second filtered signals.

1 1 . The circuit as claimed in claim 10, wherein the output of the proportional-integral controller is multiplied with a modulus of the AC grid voltage phase shifted through 45 degrees to obtain the modulation signal.

12. The circuit as claimed in claim 1 1 , wherein the polarity of the phase shift is based on the type of DC element and power converter.

13. The circuit as claimed in claim 12, wherein the control unit further comprises a modulation circuit for modulating the first and second phase output with the modulation signal to provide a modulated control signal, said modulated control signal providing the control signal for controlling the first switching element.

14. The circuit as claimed in claim 13, wherein the control unit further comprises a pulse generator for generating a pulsed signal,

wherein the modulation circuit is configured to further modulate the first and second phase output signals with the pulsed signal.

15. The circuit as claimed in claim 14, wherein the pulse generator is a saw-tooth generator for generating a saw-tooth waveform.

16. The circuit as claimed in claim 15, wherein the first phase output signal identifies a "Boost" mode of operation, and the second phase output signal identifies a "Buck" mode of operation.

17. The circuit as claimed in claim 15, wherein the first phase output signal identifies a "Buck" mode of operation, and the second phase output signal identifies a "Boost" mode of operation.

18. The circuit as claimed in claim 16 or 17, wherein the modulation circuit comprises logic gates configured to receive the modulated control signal, saw-tooth waveform, and first and second phase output signals as input, and to output the first control signal;

wherein first switching element is activated by the first control signal

when the modulated control signal is greater than the saw-tooth waveform, and the "Buck" mode is identified; or

when the modulated control signal is less than the saw-tooth waveform, and the "Boost" mode is identified.

19. The circuit as claimed in claim 18, wherein the first or second phase output signal provides the second control signal for controlling the second switching element, wherein the second switching element is activated by the second control signal when "Buck" mode is identified.

20. A system for mitigating power-ripple, the system comprising

a DC power source;

a DC to AC converter, with a DC-side of the converter coupled to the DC power source and an AC-side of the converter coupled to an AC power grid; and

a circuit electrically coupled in parallel and between the DC power source and the DC to AC converter, wherein the circuit comprises

an auxiliary circuit comprising an inductor element for providing energy storage; and

a control circuit configured to generate one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the DC source and the DC to AC converter.

21 . A system for mitigating power-ripple, the system comprising

a DC load;

an AC to DC converter, with a DC-side of the converter coupled to the DC load and an AC-side coupled to an AC power grid; and

a circuit electrically coupled in parallel and between the DC load and the AC to DC converter, wherein the circuit comprises an auxiliary circuit comprising an inductor element for providing energy storage; and

a control circuit configured to generate one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the AC to DC converter and the DC load.

22. A method for mitigating power-ripple in a power supplied between a DC element and a power converter connected to a single-phase AC power grid, the method comprising

electrically coupling a circuit in parallel and between the DC element and the power converter;

providing energy storage with an inductor element in an auxiliary circuit; and generating, with a control unit, one or more control signals to charge or discharge the inductor element, based on an instantaneous power supplied between the DC element and the power converter.

23. The method as claimed in claim 22, wherein the auxiliary circuit comprises a positive and a negative terminal; a first and a second switching element; each one of the first and second switching elements controlled by respective first and second control signals generated by the control circuit; a first and a second diode; the method further comprising

arranging the first switching element, second switching element, first diode, second diode and inductor element arranged in a H-bridge configuration across the positive and negative terminals, such that

activating the first switching element and deactivating the second switching element allows substantially zero voltage to be applied across the inductor, such that the current freewheels at the positive terminal, with the current flowing through the second diode, the inductor element and the first switching element respectively; and deactivating the first and second switching elements allows a negative voltage to be applied across the inductor, with the current flowing from the negative terminal, through the first diode, the inductive element, and the second diode, to the positive terminal.

24. The method as claimed in claim 23, the method comprising

obtaining, with a first current sensor, a first instantaneous current between the DC element and the power converter, said first instantaneous current representative of the instantaneous power supplied; and

filtering, with a bandpass filter, the first instantaneous current to obtain a first filtered signal comprising an AC component of the first instantaneous current, said first filtered signal providing a basis for the control circuit to generate the first and second control signals to charge or discharge the inductor.