WO2012086095A1

WO2012086095A1 - Motor-driving apparatus for driving three-phase motor of variable speed type

Info

Publication number: WO2012086095A1
Application number: PCT/JP2010/073883
Authority: WO
Inventors: Shouichi Tanaka
Original assignee: Three Eye Co., Ltd.
Priority date: 2010-12-24
Filing date: 2010-12-24
Publication date: 2012-06-28

Abstract

A motor-driving apparatus can increase a motor torque of a variable-speed-type three-phase motor in a wide speed range. Switching power loss of the motor-driving apparatus can be decreased, and an amplitude of an output voltage applied to a variable-speed-type three-phase motor can be increased. The motor-driving apparatus has a nine-switch inverter driving the first and the second three-phase windings of one motor. Turns of a stator winding of the one motor is changed by means of switching the nine-switch inverter. Preferably, the motor-driving apparatus has a three-phase inverter and a boost DC/DC converter, which are operated the single-phase-switching method. The three-phase inverter consists of a nine-switch inverter applying the three-phase voltage to two three-phase windings of the motor. The single-phase-switching method reduces a switching power loss of the nine switch inverter largely. The boost DC/DC converter boosting a battery voltage reduces a copper loss of the inverter and the motor, too. Preferably, ripple of the DC link voltage is absorbed by the surge-absorbing circuit having two capacitors capable of changing series-connected and parallel-connection.

Description

DESCRIPTIOM

MOTOR-DRIVING APPARATUS FOR DRIVING THREE-PHASE MOTOR OF

VARIABLE SPEED TYPE

BACKGROUND OF THE INVENTION

[0001]

1. Field of the Invention

The present invention relates to a motor-driving apparatus for driving a three-phase motor of variable speed type, in particular to a motor-driving apparatus for driving a three-phase traction motor of a vehicle .

[0002]

2. Description of the Related Art

[0003]

A three-phase induction motor is preferable for a variable-speed motor, for example a traction motor of a vehicle. However, the traction motor needs a very wide speed range, for example from zero to more than 15000 rpm. The induction motor has lower torque in a low-speed range than a permanent magnet synchronous motor.

[0004]

It is well-known that a torque-speed characteristic of a motor can be changed, when a turn number of a stator winding is changed. Generally, the turn number is decreased in the high speed range, and increased in the low speed range. A series-parallel-changing-method is popular for the turn-number-changing. The series-parallel-changing-method changes series-connection to parallel-connection. However, the series-parallel-changing-method of a three-phase motor-driving apparatus needs many AC power switches, which normally consist of mechanical relays. The mechanical relays are not preferable for frequent switching.

[0005] A partial-short-circuiting-method is known for the turn-number-changing. The partial-short-circuiting-method short-circuits a part of phase windings, when reduction of the turn number is required. However, the partial-short-circuiting-method of a three-phase motor-driving apparatus needs many AC power switches, which normally consist of mechanical relays, too. It is capable of employ a plurality of one-directional power switches instead of mechanical relays or bi-directional transistors. However, the partial-short-circuiting-method or the series-parallel-changing-method is very complicated if one-directional power switches are employed. Moreover, many power switches of the partial-short-circuiting-method or the series-parallel-changing-method increase power consumption of the motor-driving circuit. After all, it is not easy to employ the series-parallel-changing-method or the partial-short-circuiting-method for the vehicle motor, which is sensitive for a production cost and an increasing power loss.

[0006]

The document with the title, "A Novel Nine-Switch Inverter For Independent Control of Two Three-Phase Loads, Industry Applications Conference, 2007. 2346-2350 pp . 42nd IAS Annual Meeting, Conference Record of the 2007 IEEE, Issue Date : 23-27 Sept. 2007" proposes a two-motor-driving method by a nine-switch-inverter. However, the document does not describe about one-motor-driving method capable of changing turns of windings by means of employing the nine-switch inverter.

[0007]

The document with the title, "Switching Loss Reduction Using a Single-Phase PWM Control Method, 4-029, IEE JAPAN, 2009" proposes a single-phase-switching-method of voltage type. However, the above document does not describe to adopt the single-phase-switching-method in order to drive a variable-speed three-phase motor. Because, a current of the motor, especially the variable-speed motor changes in a wide range. When the motor current is small, the inverter applies small amplitude of three-phase voltage. The single-phase-switching-method with the three-phase inverter and the boost DC/DC converter can not apply such small voltage. Furthermore, the above single-phase-switching-method reducing a switching power loss of the inverter requires the PWM" switched boost DC/DC converter.

[0008]

It is not reasonable to employ the DC/DC converter for reducing the switching power loss of the inverter. Because, the sum of the switching power losses of the inverter and the boost DC/DC converter is mostly equal to a conventional three-phase inverter driven with the spontaneous space vector method. Both have two PWM-switched half bridges. The motor-driving apparatus with the single-phase-switching-method requires more production cost, because the single-phase-switching-method requires the expensive boost DC/DC converter. Consequently, the idea about the motor-driving apparatus drive with the single-phase-switching-method was not reasonable on ordinary applications using motors.

Citation List

Non-patent Literature

[0009]

NPTL l : A Novel Nine-Switch Inverter For Independent Control of Two Three-Phase Loads, Industry Applications Conference, 2007. 2346-2350 pp . 42nd IAS Annual Meeting, Conference Record of the 2007 IEEE, Issue Date : 23-27 Sept. 2007

[0010]

NPTL 2: Switching Loss Reduction Using a Single-Phase PWM Control Method, 4-029, IEE JAPAN, 2009 SUMMARY OF THE INVENTION

[0011]

One object of the invention is to provide a motor-driving apparatus for increasing a motor torque of a variable-speed-type three-phase motor in a wide speed range. Another object of the invention is to provide a motor-driving apparatus capable of decreasing a switching power loss and increasing amplitude of an output voltage applied to a variable- speed-type three-phase motor. Another object of the invention is to provide a motor-driving apparatus capable of decreasing a capacitance of a smoothing capacitor for reducing a switching surge voltage generated by the motor-driving apparatus.

[0011]

According to a first aspect of the invention, the motor-driving apparatus for driving a three-phase motor of variable speed type has a nine-switch inverter (4) driving two three-phase windings (4A and 4B) of one induction motor. The nine-switch inverter (4) with three legs ( l "3) has three upper switches (11, 2 1 and 31), three-lower switches (12, 22 and 32) and three middle switches ( 13, 23 and 33).

[0012]

The first three-phase winding (4A) is connected between three upper switches (11 , 21 and 31) and three middle switches (13, 23 and 33) . The second three-phase winding (4B) is connected between three middle switches ( 13, 23 and 33) and three lower switches (12, 22 and 32) .

[0013]

Turns of a stator winding consisting of the first three-phase winding (4A) and the second three-phase winding (4B) are changed by means of selecting one of a series connection mode and a parallel connection mode. According to the first aspect, the motor torque of a variable-speed-type three-phase motor is increased in a wide speed range easily with the simple circuit topology mentioned above.

[0014]

According to a preferred embodiment, the controller selects the series connection mode in a low speed range, and selects the parallel connection mode in a high speed range. Therefore, the turns of the three-phase winding is increased in the low speed range, and decreased in the high speed range.

[0015]

According to another preferred embodiment, the series connection mode makes double poles of the stator poles in comparison with the parallel mode. Therefore, the motor can have a rich torque in the low-speed range, and have the wide rotating range. According to another preferred embodiment, the six phase windings (U l , V2, Wl, U2, VI and W2) have equal turns to each other. Therefore, the torque ripple of the motor is reduced.

[0016]

According to a second aspect of the invention, a boost DC/DC converter applies a biggest inter-phase voltage (Vx) between two legs of a three-phase nine-switch inverter. The PWM-switched other one leg of the three-phase nine-switch inverter outputs a smaller inter-phase voltage (Vy). The biggest inter-phase voltage (Vx) is larger than the smaller inter-phase voltage (Vx) . The nine-switch inverter applies at least one three-phase sinusoidal voltage to two three-phase windings. The two three-phase windings are connected to series or connected in parallel. After all, it can be realized with less switching power loss to increase a motor torque at a high speed range, by means of controlling of the nine switches of the inverter.

[0017]

According to a preferred embodiment, the boost DC/DC converter boosts a battery voltage applied from a vehicle battery and applies the boosted biggest inter-phase voltage (Vx) to the inverter driving the motor being a traction motor of a vehicle. The inverter driving the traction motor having a wide rotating speed range reduces own switching power loss. Furthermore, it is possible to reduce the battery voltage of the vehicle, because the boost DC/DC converter applies the boosted voltage to the motor via the inverter. Accordingly, the car maintenance becomes easy and safety, because the battery voltage can be decreased. In the other words, the vehicle traction apparatus, for example a hybrid car, generally employs the boost DC/DC converter for increasing the DC-link voltage applied to the inverter in order to avoid to increase the battery voltage. Consequently, the single-phase-switching method which needs the boost DC/DC converter can be realized without addition of the boost DC/DC converter.

[0018]

According to the second aspect of the invention, a boost DC/DC converter applies a biggest inter-phase voltage (Vx) between two legs of a three-phase inverter. The PWM- switched other one leg of the three-phase inverter outputs a smaller inter-phase voltage (Vy) . The biggest inter-phase voltage (Vx) is larger than the smaller inter-phase voltage (Vx). The inverter applies a three-phase voltage to a three-phase winding. The boost DC/DC converter controls amplitude and a waveform of the biggest inter-phase voltage (Vx) in accordance with a value of a motor current and a value of a rotor angle of the motor. As the result, it is realized largely to save a switching power loss of the inverter driving the variable-speed three-phase motor. In the other words, the variable-speed motor is driven with the single-phase-switching mode (SPSM) with the three -phase inverter and the boost DC/DC converter. As the result, the switching power loss of the inverter and a copper loss of the variable-speed three-phase motor can be reduced largely.

[0019] The biggest inter-phase voltage Vx is changed every 60 degrees in order. Only one leg of the three-phase inverter is switched with the PWM method. The PWM-switched leg is changed every 60 degrees of the electric angle in order. The other two fixed legs, which are half-bridges each, are not switched with the PWM method. A duty-increasing mode and a duty-decreasing mode are operated each 60 degrees of the electric angle alternatively for controlling of the PWM-switched leg. In the duty-increasing mode, the PWM duty ratio increases from 0% of to 100% successively. In a duty-decreasing mode, the PWM duty ratio decreases from 100% of to 0% successively. A pair of two legs for outputting the biggest inter-phase voltage Vx is changed every 60 degrees of the electric angle in order.

[0020]

According to a preferred embodiment, the biggest inter-phase voltage (Vx) has a three -phase -full-wave -rectified wave form. The inverter outputs a three -phase sinusoidal voltage of which a frequency is changed in accordance with the rotation speed of the motor. Accordingly, vibration and noise of the noise are reduced, because the motor is driven with the three-phase sinusoidal voltage.

[0021]

According to another preferred embodiment, the boost DC/DC converter consists of a chopper type DC/DC converter. The controller (9) changes a PWM duty ratio of the boost DC/DC converter (8) in the single-leg-switching mode in accordance with a received torque instruction value (Tr) and a detected rotation speed (ω) of the motor (6) . As the result, the motor can have a preferable operation. Furthermore, the chopper-type boost DC/DC converter can charge the vehicle battery when the rotation speed is decreased.

[0022]

According to another preferred embodiment, the controller has a table keeping a relation among the biggest inter-phase voltage (Vx) , a rotor angle (Θ), the torque instruction value (Tr) and the rotation speed (ω ) . The controller decides the biggest inter-phase voltage (Vx) in accordance with the received values of the detected values of the rotor angle (Θ), the torque instruction value (Tr) and the rotation spee d (ω) . As the result, the variable speed motor employing the SPSM can generates a required torque value at a required rotation sp eed smoothly.

[0023]

According to another preferred embodiment, the boost DC/DC converter boosts a battery voltage applied from a vehicle battery and applies the boosted biggest inter-phase voltage (Vx) to the inverter driving the motor being a traction motor of a vehicle . Accordingly, the switching power loss of the inverter and the copper loss of the motor can be re d uced largely by means of the boosting of the battery voltage and employing of the SPSM. Furthermore, the vehicle traction apparatus, for example a hybrid car, generally employs the boost DC/DC converter in order to reduce th e battery voltage, As the result, the motor-driving apparatus does not need to add the boost DC/DC converter in order to realize the SPSM.

[0024]

According to another preferred embodiment, the motor has permanent magnets fixed to a rotor of the motor. The boost DC/DC converter applies the biggest inter-phase voltage (Vx) , which is larger than a generation voltage of the motor (6), in the single-leg-switching mode. The motor current is reduced by an induced motor voltage increased at a high rotation speed range . The reduction of the motor current is large for the motor with permanent rotor. Accordingly, the boost ratio must be increased at the high rotation speed. However, an average motor current supp lied by the inverter is reduced, because outputting periods of the boost DC/DC conve rter of the chopper type decreases. As the result, the motor torque is reduced at the high rotation speed.

[0025]

It was found that the boost ratio of the boost DC/DC converter can be largely reduced in comparison with the conventional motor-driving apparatus with the boost DC/DC converter. As the result, a power loss of the boost DC/DC converter, which is proportional to the boost voltage of the boost DC/DC converter, is reduced. Furthermore, the inverter and the converter can be made easy, because the voltage is applied to the inverter and the converter without reduction of the motor torque .

[0026]

The reduction reason of the voltage is explained. The boost converter applies a biggest inter-phase voltage Vx to the three -phase inverter. The inter-phase voltage means a voltage between two phase voltages of the three-phase voltage . The conventional boost DC/DC converter must apply two times value of the biggest amplitude of one phase voltage .

[0027]

The biggest inter-phase voltage Vx is smaller than two times value of the biggest amplitude of one phase voltage . Accordingly, the boost ratio of the boost DC/DC converter can be reduced about 15-20%. Furthermore, the voltage-outputting period of the boost converter increases, because the boost ratio is decreased. The motor torque of the variable-speed motor is proportional to a current supplied to the motor. Consequently, the variable speed motor can generate a strong torque at the high rotation speed range by employing the SPSM.

[0028]

According to another preferred embodiment, the controller further has a plural-leg-switching mode for controlling the converter and the inverter. At least two legs are switched with a PWM method in the plural-leg-switching mode. The single-leg-switching mode is selected when the biggest inter-phase voltage (Vx) is larger than the battery voltage . The plural-leg-switching mode is selected when the biggest inter-phase voltage (Vx) is smaller than the battery voltage. The plural-leg-switching mode is essentially same as the conventional PWM-switched operation of the three-phase motor.

[0029]

For example, one of the plural-leg- switching modes is the conventional three-phase PWM switching operation switching the three legs. Another one of the plural'leg-switching mode is a known two-phase modulation method or a known spontaneous space vector method. The two legs are PWM-switched. Consequently, the motor-driving apparatus can drive even though the motor voltage must be smaller than the power source voltage by means of employing the known plural-leg-switching mode. As the result, the motor-driving apparatus with the SPSM can drive the motor at the low rotation speed with the small motor torque .

[0030]

According to another preferred embodiment, the PWM-switched leg is PWM-switched for outputting the smaller inter-phase voltage (Vy) while the boost converter outputs the biggest inter-phase voltage (Vx) . The turning-on period of the upper switching element of the PWM-switched leg is shorter than the outputtin period of the boost converter outputting the biggest inter-phase voltage (Vx) . The PWM-switched leg does not turned on when the potential of the high potential bus line falls down. As the result, voltage ripples of the smaller inter-phase voltage (Vy) are reduced even though the capacitance of the smoothing capacitor is not large .

[0031]

According to another preferred embodiment, a surge-absorbing circuit as a smoothing circuit absorbs a surge voltage generated by the inverter or the converter. The surge-absorbing circuit has two capacitors which are charged independently one another. The capacitors are discharged when the capacitors are connected to series. Switching elements charging and discharging of the two capacitors are controlled by a high frequency component of a voltage . When the voltage drops rapidly, the capacitors connected to series are discharged. The capacitors are charged independently, when the voltage rises rapidly. As the result, the two capacitors absorb the surge energy.

[0032]

The above-explained surge-absorbing circuit needs a small capacitance value in comparison with a conventional smoothing capacitor. Accordingly, the inverter-converter with the single-phase switching mode can reduce the capacitance of the smoothing capacitor by employing the surge-absorbing circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

[0033]

Figure 1 is a circuit diagram showing a motor-driving circuit of the first embodiment.

Figure 2(A) is a circuit diagram showing a nine-switch three-phase inverter with a series connection mode.

Figure 2(B) is a circuit diagram showing the nine-switch three-phase inverter with the series connection mode, too.

Figure 3(A) is a circuit diagram showing the nine-switch three-phase inverter with a parallel connection mode.

Figure 3(B) is a circuit diagram showing the nine -switch three-phase inverter with the parallel connection mode shown, too.

Figure 4(A) is a schematic development showing a series connection mode of a stator of a three-phase induction motor.

Figure 4(B) is a schematic development showing a parallel connection mode of the stator of the three-phase induction motor.

Figure 5 is a flow chart showing of mode-changing operation.

Figure 6 is a vector diagram showing the three-phase flux at the series connection mode .

Figure 7 is a vector diagram showing the three-phase flux at the parallel connection mode .

Figure 8 is a block circuit diagram showing a motor-driving apparatus driving a variable speed three-phase motor.

Figure 9 is a schematic connection diagram showing six switching states of the three-phase inverter shown in Figure 8.

Figure 10 is a diagram showing a six gate voltage patterns in six stages of the three-phase inverter shown in Figures 8 and 9.

Figure 11 is a timing chart showing one PWM-carrier period of the three-phase inverter shown in Figures 8" 10.

Figure 12 is a wave form of the three-phase voltage applied to the motor by the three-phase inverter shown in Figures 8" 10.

Figure 13 is a timing chart showing the biggest inter-phase voltage applied to the inverter by the converter shown in Figure 8.

Figure 14 is a block diagram of a controller controlling the three-phase inverter-converter shown in Figure 8.

Figure 15 is a vector diagram showing the biggest inter-phase voltage and a smaller inter-phase voltage .

Figure 16 is a circuit diagram showing the stage when the converter outputs the biggest inter-phase voltage and the switched phase leg of the three-phase inverter is turned off.

Figure 17 is a circuit diagram showing the stage when the converter outputs the biggest inter-phase voltage and the switched phase leg of the three-phase inverter is turned on.

Figure 18 is a circuit diagram showing the stage when the converter does not output the biggest inter-phase voltage and the switched phase leg of the three -phase inverter is turned off.

Figure 19 is a timing chart of the converter and the switched phase leg.

Figure 20 is a timing chart of the converter and the switched phase leg.

Figure 21 is another timing chart of the converter and the switched phase leg.

Figure 22 is a timing chart showing an error-following PWM method as the one of the PWM method.

Figure 23 is a circuit diagram operating the error-following PWM method.

Figure 24 is a flow chart showing to control the operation of the variable speed motor.

Figure 25 is a timing chart showing wave patterns from the converter shown in Figure 8.

Figure 26 is a circuit diagram showing three states of a nine-switch inverter driving a motor with two three-phase windings connected in parallel.

Figure 27 is a circuit diagram showing three states of the nine-switch inverter driving a motor with two three-phase windings connected in parallel.

Figure 28 is a circuit diagram showing four states of the nine-switch inverter driving a motor with two three-phase windings connected to series.

Figure 29 is a circuit diagram showing four states of the nine-switch inverter driving a motor with two three-phase windings connected to series.

Figure 30 is a circuit diagram showing four states of the nine-switch inverter driving a motor with two three-phase windings connected to series.

Figure 31 is an equivalent circuit diagram showing the nine-switch inverter driving two three-phase windings connected in parallel. Figure 32 is an equivalent circuit diagram showing the nine-switch inverter driving two three-phase windings connected to series.

Figure 33 is a schematic development showing a stator of a three-phase induction motor.

Figure 34 is a schematic development showing a stator of a three-phase induction motor.

Figure 35 is a schematic development showing a stator of a three-phase induction motor.

Figure 36 is a schematic development showing a stator of a three -phase induction motor.

Figure 37 is a circuit diagram showing four states of the nine-switch inverter driving two series-connected three-phase windings by the spontaneous space vector method.

Figure 38 is a circuit diagram showing two states of the nine-switch inverter driving two series-connected three-phase windings by the spontaneous space vector method.

Figure 39 is a timing chart showing one PWM-carrier period of the nine -switch inverter.

Figure 40 is a circuit diagram showing a surge -absorbing circuit as a smoothing capacitor.

PREFERRED EMBODIMENTS OF THE INVENTION

[0034]

A motor-driving apparatus of the embodiment operating the turn-number-changing-method (TNCM) is explained. Furthermore, a motor-driving apparatus of the single -phase-switching-method (SPSM) is explained. Moreover, a motor-driving apparatus with the surge-absorbing method is explained. It is considerable that these three methods are operated together, even though these three methods can be operated independently. [0035]

(The first embodiment)

Figure 1 shows a circuit topology of a motor-driving circuit having a three-phase inverter 4, which drives a three-phase induction motor having a stator winding 6. The stator winding 6 consists of a first three-phase winding 4A and a second three-phase winding 4B. The star-connected first three-phase winding 4A consists of a U-phase winding U l , a V-phase winding VI and a W-phase winding W l .

[0036]

The star-connected second three-phase winding 4B consists of a U-phase winding U2, a V-phase winding V2 and a W-phase winding W2. However, the first three-phase winding 4A and the second three-phase winding 4B can have a delta-connection instead of the star-connection. The phase windings U l, VI, Wl, U2, V2 and W2 can be wound with a concentrated (centralized) winding or a distributed winding. The inverter 4 consists of a U-phase leg 1 , a V-phase leg 2 and a W-phase leg 3, which are connected in parallel to each other between a high potential bus line 100 and a low potential bus line 101. A DC-link voltage VX is applied between two bus lines 100 and 101.

[0037]

The U-phase leg 1 consists of an upper switch 11, a middle switch 13 and a lower switch 12, which are connected in series to each other. The V-phase leg 2 consists of an upper switch 21, a middle switch 23 and a lower switch 22, which are connected in series to each other. The W-phase leg 3 consists of an upper switch 31, a middle switch 33 and a lower switch 32, which are connected in series to each other. An upper-switch group X consists of switches 11, 21 and 31. A middle-switch group Y consists of switches 13, 23 and 33. A lower-switch group Z consists of switches 12, 22 and 32. The first three-phase winding 4A is connected between the upper-switch group X and the middle - switch group Y. The second three-phase winding 4B is connected between the middle-switch group Y and the lower-switch group Z.

[0038]

Figure 2(A) and 2(B) show a series connection mode. Figure 2(A) shows the series connection mode, when amplitude of the U-phase current IU 1 (=IU2) is larger than amplitudes of the other two phase currents IV1 (=IV2) and IW1 (=IW2) in a positive area. Figure 2(B) shows the series connection mode, when amplitude of the U-phase current IU 1 (=IU2) is larger than amplitudes of the other two phase currents IV1 (=IV2) and IW1 (=IW2) in a negative area.

[0039]

In Figure 2(A), the phase currents IU 1 (=IU2), IV1 (=IV2) and IW1 (=IW2) flow, when the switches 11, 12, 23 and 33 are turned on. In Figure 2(B), the phase currents IU 1 (=IU2) , IV 1 (=IV2) and IW1 (=IW2) flow, when the switches 13, 21, 22, 31 and 32 are turned on.

[0040]

Consequently, the middle switches 13, 23 and 33 become lower switches of the conventional six-switch-inverter for the first three-phase winding 4A in the series connection mode. Similarly, the middle switches 13, 23 and 33 becomes a upper switches of the conventional six-switch type three-phase inverter for the second three-phase winding 4B in the series connection mode.

[0041]

Figure 3(A) and 3(B) show a parallel mode. Figure 3(A) shows the parallel mode, when amplitude of the U-phase current IU 1 (=IU2) is larger than amplitudes of the other two phase currents IV1 (=IV2) and IW1 (=IW2) in a positive area. Figure 3(B) shows the parallel connection mode, when amplitude of the U-phase current IU 1 (=IU2) is larger than amplitudes of the other two phase currents IV1 (=IV2) and IW1 (=IW2) in a negative area.

[0042] In the parallel connection mode, three middle switches 13, 23 and 33 are always turned on. As the result, the inverter 4 becomes the conventional six-switch type three-phase inverter for the first and the second three-phase windings 4A and 4B. Consequently, the DC link voltage Vx is applied to both of the first three-phase winding 4A and the second three-phase winding 4B, which are connected in parallel to each other.

[0043]

The inverter 4 can be switched by any switching methods for driving a conventional three-phase inverter. For example, a three -phase PWM moderation method, a two-phase PWM modulation method, a space vector PWM method and one pulse method can be employed. Moreover, the single -phase-switching-method (SPSM) explained later can be employed, too. Each turns of the phase winding U l, VI , Wl , U2, V2 and W2 is equal to each other.

[0044]

Figure 4(A) shows a schematic development of six teeth 1001U 1, 1001V2, 1001W1 , 1001U2, 1001V 1 , 1001W2 of the three -phase induction motor in the series connection mode. Phase windings U l , V2, Wl, U2, VI and W2 are wound on six teeth 1001U 1 , 1001V2, 1001W1, 1001U2, 1001V1, 1001W2 arranged in turn respectively with the concentrated-winding method. The distributed-winding method can be employed instead of the concentrated-winding method. Phase currents Iu l, Iv2, Iw l, Iu2, Iv l and Iw2 are supplied to the phase winding U l, V2, Wl , U2, VI and W2 respectively.

[0045]

In Figure 4(A), three phase windings U l , VI and Wl of the first three-phase winding 4A and three phase windings U2, V2 and W2 of the second three-phase winding 4B are connected in series to each other. Phase current Iu l excites the U-phase flux Wu l in the teeth 1001U 1. Phase current Iv2 excites the V-phase flux Ψν2 in the teeth 1001V2. Phase current Iw l excites the W-phase flux Ww l in the teeth 1001W1. Phase current Iu2 excites the U-phase flux Τη2 in the teeth 1001U2. Phase current Iv l excites the V-phase flux Ψν ΐ in the teeth 1001V1. Phase current Iw2 excites the W-phase flux in the teeth 1001W2.

[0046]

Each of flux vectors is shown in Figure 6. These vectors are produced by means of employing the series connection mode shown in Figure 2. The series connection mode is employed in the low rotation speed range. As the result, the induction motor has two three-phase windings 4A and 4B with double turns per phase, when each phase winding has ten turns. Furthermore, six teeth being adjacent to each other constitute an angle of 360 degrees.

[0047]

Figure 4(B) shows a schematic development of six teeth 1001U 1, 1001V2, 1001W1, 1001U2, 1001V 1 , 1001W2 of the three -phase induction motor in the parallel connection mode . The six teeth 1001U 1, 1001V2, 1001W1, 1001U2, 1001V1, 1001W2 are arranged in an electrical angle of 720 degrees, because the phase current Iu2, Iv2 and Iw2 have an opposite direction each. In Figure 4(B), three phase windings U l, VI and Wl of the first three-phase winding 4A and three phase windings U2, V2 and W2 of the second three-phase winding 4B are connected in parallel to each other. Accordingly, directions of phase currents Iu2, Iv2 and Iw2 become opposite in comparison with them of the series connection mode shown in Figure 4(A). As the result, U-phase flux ^Pu l has the equal direction to U-phase flux ^TPu2. V-phase flux Φν ΐ has the equal direction to V-phase flux Ψν2. W-phase flux Ww l has the equal direction to W-phase flux Fw2.

[0048]

Each of flux vectors is shown in Figure 7. These flux vectors are produced by means of employing the parallel connection mode. The parallel connection mode is employed in the high rotation speed range. As the result, the induction motor has two three-phase windings 4A and 4B of which two phase windings of each phase are connected in parallel to each other.

[0049]

One control method of the above induction motor is explained referring to Figure 5. Firstly, it is judged at the step S 100 whether or not the series connection mode should be executed. For example, it is judged whether or not the present operation mode is the parallel connection mode, and whether or not the rotation speed is lower than the predetermined rotor speed. The series connection mode is selected at the step 102, when the rotation speed is lower.

[0050]

Next, it is judged at the step S 104 whether or not the parallel connection mode should be executed. For example, it is judged whether or not the present operation mode is the series connection mode, and whether or not the rotation speed is higher than the predetermined rotor speed. The parallel connection mode is selected at the step 106, when the rotation speed is higher.

[0051]

(The second embodiment)

The turn-number-changing-method (TNCM) described on the above embodiments increases a low-speed torque and enlarges a rotation speed range, because both of the stator poles and the turns are changed. However, the power consumption of the motor-driving apparatus is increased by the added middle switches. The power consumption of the motor-driving apparatus is reduced by employing the single-phase switching method (SPSM) explained hereinafter. The SPSM is explained referring to Figures 8-25. Figure 8 is a circuit diagram showing the motor-driving apparatus operated with the SPSM. [0052]

A three-phase motor consisting of a synchronous motor or an induction motor is driven by the motor-driving apparatus, which has a three-phase inverter 4, a smoothing capacitor 5, a chopper type boost DC/DC converter 8, and a controller 9. A battery voltage Vb of a battery 7 is applied to the converter 8. The inverter 4 outputs a three-phase voltage to motor 6. In the embodiment, motor 6 has a star-connected three-phase stator winding consisting of a U-phase winding 6U, a V-phase winding 6V and a W-phase winding 6W.

[0053]

Three-phase inverter 4 has a U-phase leg 1, a V-phase leg 2 and a W-phase leg 3. Each of the legs 1 - 3 consists of a half-bridge . U-phase leg 1 has an upper switch 11 and a lower switch 12 connected to series. V-phase leg 12 has an upper switch 21 and a lower switch 22 connected to series. W-phase leg 3 has an upper switch 31 and a lower switch 32 connected to series. Each switch consists of a power transistor and a free -wheeling diode connected in parallel to each other. Inverter 4 drives three-phase motor 6.

[0054]

Boost DC/DC converter 8 changes the supplied battery voltage Vb to a periodically-changed DC-link voltage Vx, which is the biggest inter-phase voltage explained later. The converter 8 outputs the DC-link voltage Vx to inverter 4 through a high potential bus line 100 and a low potential bus line 101. Converter 8 periodically changes amplitude of DC-link voltage Vx applied to inverter 4.

[0055]

The SPSM operation of inverter 4 is explained referring to Figure 9. Figures 9 shows stages A-F. An electric angle of 360 degrees is divided to six stages A"F having the electric angle of 60 degrees each. One of the stages A-F is selected in accordance with a detected rotor angle in turn. [0056]

In each of stages A-F, four switches of two half-bridges keep a constant state. In the other words, two fixed legs of the three legs 1 -3 are not PWM-switched. In each of stages A-F, only two switches of one half-bridge are PWM-switched in order to produce a smaller inter-phase voltage (Vy) with the sinusoidal waveform. In the other words, one switched leg is PWM-switched. The PWM-switched half bridge is called the switched leg. The other two half-bridges of which four switches are not PWM-switched is called the fixed legs.

[0057]

As shown in Figure 9, V-phase leg 2 is the switched leg in the stages A and D. W-phase leg 3 is the switched leg in the stages B and E. U-phase leg 1 is the switched leg in the stages C and F. The upper switch and the lower switch of the switched leg are PWM-switched. Upper switch 11 of U-phase leg 1 and lower switch 32 of W-phase leg 3 are turned-on in the stage A. The biggest inter-phase voltage Vx is applied to U-phase winding 6U and W-phase winding 6W. In the stage A, U-phase leg 1 and W-phase leg 3 become the fixed legs.

[0058]

In Figure 9, U-phase leg 1 outputs U-phase voltage Vu. V-phase leg 2 outputs V-phase voltage Vv. W-phase leg 3 outputs W-phase voltage Vw. A voltage between two phase voltages selected among three phase voltages Vu, Vv and Vw is called the inter-phase voltage . The inter-phase voltage having the biggest amplitude is called as the biggest inter-phase voltage Vx.

[0059]

Figure 10 shows six states of six switches 11, 12, 21, 22, 31 and 32 of inverter 4 in the six stages A-F. A gate voltage UU is applied to the switch 11. A gate voltage UL is applied to the switch 12. A gate voltage VU is applied to the switch 21. A gate voltage VL is applied to the switch 22. A gate voltage WU is applied to the switch 31. A gate voltage WL is applied to the switch 32. Each of the six switches of inverter 4 is PWM- switched for an electric angle of 60 degrees. In the next electric angle of 120 degrees, each of the six switches are turned off or turned on continuously and radiated. Accordingly, it is suppressed to over-heat the switches.

[0060]

Figure 11 shows wave forms of the gate voltages applied to the six switches of inverter 4 for one PWM-carrier period TP of the stage A. In one PWM-carrier-period TP, the switches 11 and 32 are turned on and the switches 12 and 31 are turned off. The switches 21 and 22 of V-phase leg are PWM-switched.

[0061]

One of two switches of the switched leg has the duty ratio changing from 0% to 100% in the period of 60 degrees excessively. The other one of two switches of the PWM phase has the duty ratio changing from 100% to 0% in the period of the above 60 degrees excessively.

[0062]

The SPSM operation of the converter 8 is explained referring to Figure 8 and Figure 12. Figure 12 shows a three-phase sinusoidal wave voltage applied to motor 6. The converter 8 outputs DC-link voltage Vx, which is the biggest inter-phase voltage Vx, to the inverter 4. The boost operation of converter 8 is well known. By turning-on of the lower switch 8F, the reactor 8C accumulates the magnetic energy. By turning-off of the lower switch 8F, the boost voltage is applied to the high potential bus line 100 through the switch 8E.

[0063]

Smoothing capacitor 5 connects the high potential bus line 100 to a positive terminal of the battery 7. Smoothing capacitor 5 absorbs the surge energy when upper switches 11, 21 and 31 are turned off. Furthermore, smoothing capacitor 5 reduces the voltage ripple of high potential bus line 100. However, a large capacitance of the smoothing capacitor 5 prevents to change the biggest inter-phase voltage Vx.

[0064]

Controller 9 calculates a duty ratio Dx of the converter 8 in order to output the biggest inter-phase voltage Vx with the sinusoidal waveform in accordance with the detected rotor angle and an instruction value of a motor torque. Controller 9 can control converter 8 with the well-known PWM feedback control method. Furthermore, controller 9 calculates the duty ratio Dy of the switched leg of inverter 4 in order to output the smaller inter-phase voltage Vy with the sinusoidal waveform in accordance with the detected rotor angle and the instruction value of the motor torque. The biggest inter-phase voltage Vx and the smaller inter-phase voltage Vy are shown in Figure 22.

[0065]

The biggest inter-phase voltage Vx is changed as shown in Figure 12. In the stage A from the 30 degree to the 90 degree, the biggest inter-phase voltage Vx is the inter-phase voltage Vu-Vw. In the stage B from the 90 degree to the 150 degree, the biggest inter-phase voltage Vx is the inter-phase voltage Vu-Vv. In the stage C from the 150 degree to the 210 degree, the biggest inter-phase voltage Vx is the inter-phase voltage Vw-Vv. In the stage D from the 210 degree to the 270 degree, the biggest inter-phase voltage Vx is the inter-phase voltage Vw-Vu. In the stage E from the 270 degree to the 330 degree, the biggest inter-phase voltage Vx is the inter-phase voltage VvVu. In the stage F from the 330 degree to the 30 degree, the biggest inter-phase voltage Vx is the inter-phase voltage VvVw.

[0066]

The biggest inter-phase voltage Vx has a waveform shown in Figure 13. The waveform of the biggest inter-phase voltage Vx is equal to a full-wave -rectified three-phase voltage. The value of the biggest inter-phase voltage Vx is 1.5- 1.73, when the biggest value of one phase-voltage is 1.

[0067]

A boost ratio of converter 8 becomes in a range from 75% to 86.5% of the boost ratio of the conventional motor-driving apparatus with a converter and an inverter. As the result, the upper switch 8E of converter 8 can have higher duty ratio than the conventional motor-driving apparatus. For example, the converter of the conventional motor-driving apparatus outputs a boost voltage of 700 V, when a battery voltage Vb is 250V. The boost ratio becomes 2.8. On the other hand, the converter of the motor-driving apparatus of the embodiment outputs the boost voltage of only 525-605V Both of the apparatuses can apply an equal biggest inter-phase voltage Vx to the inverter 4. Accordingly, the converters of the both apparatuses have equal value of the output current. The boost voltage of the converter of the embodiment is reduced largely.

[0068]

The biggest inter-phase voltage Vx has a part of sinusoidal waveform in each stage A"F. As the result, only one leg is PWM-switched in order to output the three-phase sinusoidal waveforms as shown in Figure 12. A value of the smaller inter-phase voltage Vy alternately changes from 0% to 100% and from 100% to 0% of a value of the biggest inter-phase voltage Vx.

[0069]

A memory in the controller 9 has a map keeping a relation between the rotor angle and the relative duty ratio Dz in order to decide the smaller inter-phase voltage Vy. Relative duty ratio Dz, which is equal to Dy/Dx, shows a relative amplitude ratio between smaller inter-phase voltage Vy and the biggest inter-phase voltage Vx. The half-bridge consisting of the switched leg is PWM-switched with PWM duty ratio Dz. [0070]

Controller 9 reads the relative duty ratio Dz from a map in accordance with the detected rotor angle Θ. The map memorizes each relative duty ratio Dz in each rotor angle . The upper arm switches 11, 21 and 31 of the switched leg are PWM- switched with the relative duty ratio Dz. The lower arm switches 12, 22 and 32 of the switched leg are complimentary PWM-switched with the relative duty ratio 1-Dz. Three phase voltages Vu, Vv and Vw are hereby decided by only PWM-switching of one phase leg of three-phase inverter 4.

[0071]

Figure 14 shows a part of a block diagram of controller 9. Controller 9 has a stage-decision circuit 10A, wave-generation circuits 10B and 10D and PWM-signal generation circuits IOC and 10E. The stage-decision circuit 10A decides the present stage in accordance with the detected rotor angle Θ. The map shown in Figure 10 in the memory is used for the decision. The wave-generation circuit 10B generates a PWM-signal with relative duty ratio Dz for the switched leg in accordance with the rotor angle Θ. The PWM-signal generation circuit IOC generates PWM-gate voltages UU, UL, VU, VL, WU and WL in accordance with the decided stage, the decided PWM-signal of the switched leg in each period of 60 degrees.

[0072]

The wave - generation circuit 10D generates a PWM-signal with duty ratio Dx for DC/DC converter 8B . The wave signal of the biggest inter-phase Vx is changed as shown in Figure 10. The biggest inter-phase Vx is decided in accordance with the detected rotor angle Θ and the torque instruction value Ti. The PWM-signal generation circuit 10E generates the PWM-gate voltage for DC/DC converter 8B.

[0073]

Vectors of voltage Vx and Vy are shown in Figure 15. Vu = Vm sin ω t

Vv = Vm (sin ω t"2n/3)

Vw = Vm (sin ω t+2n/3)

Vx = Vu - Vw = 1.73 * Vm * sin (<a t- 2n/3) = 1.73 * Vm * Dx

Vy = Vv - Vw = 1.73 * Vm * sin (ω t-n/2) = 1.73 * Vm * Dy

The PWM ratio Dx of the biggest inter-phase voltage Vx shows the sinusoidal waveform function of sin (cot-2n/3) . The PWM ratio Dy of the smaller inter-phase voltage Vy shows the sinusoidal waveform function of sin (cot-n/2) . Consequently, the relative duty ratio Dz, = Dy/Dx, can be obtained by calculating the following equation.

Dz = sin (ω t-n/2) / sin (ω t-2n/3)

[0074]

The pre-calculated duty ratio Dx and the pre -calculated relative duty ratio Dz are described on the map in the memory. Accordingly, the duty ratio Dx and the duty ratio Dy are searched from the map by using the detected rotor angle Θ, which is cot. The instruction value of the biggest amplitude of the phase voltage Vx, = 1.73 * Vm, is calculated in accordance with the instruction value of the motor torque . The calculated instruction value of the biggest inter-phase voltage Vx is compared with the detected value of the DC-link voltage Vx.

[0075]

The Duty ratio of converter 8 can be feedback-controlled in accordance with result of the comparison. Furthermore, the upper arm switches 11, 21 and 31 of the switched legs are switched by the PWM method with the relative duty ratio Dz. Each of the PWM-switched lower arm switches 12, 22 and 32 are complimentary switched with the relative duty ratio which is l "Dz.

[0076]

One arranged embodiment of the motor-driving apparatus with the SPSM is explained referring to Figures 16- 19. Figures 16- 19 show a circuit diagram of the motor-driving circuit for driving the three-phase motor each. The apparatus has three-phase inverter 4 and converter 8. Three-phase inverter 4 and boost converter 8 shown in Figure 16 are same as the inverter 4 and the converter 8 shown in Figure 8.

[0077]

Converter 8 has the upper switch 8E and the lower switch 8F connected to series. A connecting point between two switches 8E and 8F is connected a positive terminal of battery 8A through reactor 8C. Smoothing capacitor 8D connects between two DC link lines 100 and 101. The well-known chopper-type converter 8 is a bi-directional boost/down type DC/DC converter, which outputs a boost voltage Vx to three-phase inverter 4 and outputs a step -down voltage Vb to battery 8A.

[0078]

Operation of the above motor-driving circuit is explained as bellows. Figures 16- 18 show the operation in the stage A. U-phase leg 1 and W-phase leg 3 are the fixed legs. V-phase leg 2 is the switched leg. In Figure 10, the switches 8E, 11 , 22 and 32 are turned on. The boost voltage Vx supplies current I to the switch 11. Current I is equal to U-phase current Iu. The switch 22 supplies V-phase current Iv being the free-wheeling current. In Figure 17, the switches 8E, 11, 21 and 32 are turned on. The boost voltage Vx supplies the current I to the switches 11 and 21. Accordingly, Current I is equal to the sum of U-phase current Iu and V-phase current Iv.

[0079]

In Figure 18, the switches 8F, 11 , 22 and 32 are turned on. Reactor 8C accumulates the magnetic energy. Smoothing capacitor 8D supplies U-phase current Iu. However, V-phase upper switch 21 is turned off, when the switch 8F is turned on. Smoothing capacitor 8D does not need to supply V-phase current Iv. Accordingly, a voltage drop of smoothing capacitor 8D is reduced.

[0080]

Figure 19 shows a timing chart showing an operation of converter 8 and the switched leg of the inverter 4. The upper switch 21 of the switched leg is turned on in the period from t l to t2 in the output period of converter 8 from t3 to t2. Accordingly, the smoothing capacitor 8D can become small. Furthermore, the upper switch 21 of the switched leg is turned off at the same time when the upper switch 8E of the converter 8 is turned off. Accordingly, the voltage ripple is reduced.

[0081]

Another arranged embodiment of the motor-driving apparatus with the SPSM is explained referring to Figure 20. In this embodiment, the upper switch 21 of the switched leg is turned-on in the period when upper switch 8E is turned on. Consequently, the turning-on periods of the upper switch 21 of the switched leg is overlapped with the turning-on periods of upper switch 8E partially.

[0082]

In Figure 20, turning-on periods of the upper switch 21 of the switched leg is overlapped with the odd turning-on periods of upper switch 8E. The gate pulse P3' is further cancelled, if upper switch 21 of the switched leg has a small duty-ratio. The gate pulse voltage VU overlapped with even gate pulse voltage CU is further added if upper switch 21 of the switched leg has larger duty-ratio.

[0083]

Another arranged embodiment of the motor-driving apparatus with the SPSM is explained referring to Figures 21. Figure 21 shows a timing chart showing one relative-timing-relation between the gate voltages of inverter 4 and converter 8. The gate voltage CU is applied to the upper switch 8E . The gate voltage CL is applied to the lower switch 8F. The biggest inter-phase voltage Vx is changed by the gate voltage CL. The upper switch 21 employs one of two gate voltages VU 1 and VU2.

[0084]

The gate voltages VU 1 rises up at the essentially same timing as the falling-down timing of the gate voltage CL. As the result, the voltage ripple of the high potential bus line 100 is reduced. Because, the increasing voltage Vx of the line 100 by means of the turning-off of the lower switch 8F is reduced by means of the turning-on of the upper switch 21. The gate voltages VU2 fall down at the essentially same timing as the rising-up timing of the gate voltage CL. As the result, the voltage ripple of the line 100 is reduced, because the decreasing voltage Vx of the line 100 by means of the turning-on of the upper switch 8F is reduced by means of the turning-off of the upper switch 21.

[0085]

Consequently, the upper switches 11 , 21 and 31 of inverter 4 are turned on at the essentially same timing as the turning-off timing of the lower switch 8F in this arranged embodiment. In the other embodiment, the upper switches 11, 21 and 31 of the inverter 4 are turned off at the essentially same timing as the turning-on timing of the lower switch 8F. As the result, the noise of the line 100 and the switching loss of the upper switches of the inverter 4 are reduced.

[0086]

Another arranged embodiment of the motor-driving apparatus with the SPSM is explained referring to Figures 22 and 23. Figure 22 shows the principle of the error-following PWM method, which is one kind of the PWM method. This error-following PWM method can be employed to generate the biggest inter-phase voltage Vx and smaller inter-phase voltage Vy instead of the conventional PWM method having the PWM carrier signal with a constant frequency. [0087]

In Figure 22, a broken line shows an instruction value of the biggest inter-phase voltage Vx. Two real lines Vx + AV and Vx - AV are formed at both side of broken line Vx. DC/DC converter 8 outputs the biggest inter-phase voltage Vx within the two voltages Vx + AV and Vx AV.

[0088]

Figure 23 shows a comparator circuit of the error-following PWM method. Detected value Vxd of the biggest inter-phase voltage Vx is compared with Vx + AV and Vx - AV by the comparators 91 and 92. The AND gate 93 controls the upper switch 8E shown in Figure 26. Similarly, the detected value Vyd of the smaller inter-phase voltage Vy is compared with Vy + AV and Vy - AV by the comparators 94 and 95. The AND gate 96 controls the upper switch 21 of the switched leg 2 shown in Figure 16.

[0089]

Another arranged embodiment of the motor-driving apparatus with the SPSM is explained referring to Figure 24 and 25. Boost converter 8 outputs the biggest inter-phase voltage Vx, which is higher than battery voltage Vb. However, boost converter 8 can not output the biggest inter-phase voltage Vx, which is lower than battery voltage Vb . It means that the SPSM can not be used for the motor-driving apparatus, when the motor torque or the rotation speed is larger than a predetermined value .

[0090]

In other words, the biggest inter-phase voltage Vx is a function value, which is changed by the motor torque instruction value Tr and the rotation speed ω. The biggest inter-phase voltage Vx is almost proportional to the motor torque instruction value Tr and the rotation speed ω. As the result, the SPSM can be not operated, if the calculated biggest inter-phase voltage Vx is smaller than a predetermined value. Consequently, the motor-driving apparatus, which outputs a small torque at a low rotation speed, can not be operated by the

SPSM. The solution is explained hereinafter.

[0091]

Figure 24 shows a flowchart showing a control operation of the motor-driving apparatus with the SPSM. In Figure 24, the torque instruction value Tr, the rotor angle Θ and the rotation speed ω are detected at step S 100. Next, the biggest inter-phase voltage Vx is calculated in accordance with the torque instruction value Tr, the rotor angle Θ and the rotation speed ω at step S 102.

[0092]

The controller 9 has a table showing a relation among the biggest inter-phase voltage Vx, the torque instruction value Tr, the rotor angle Θ and the rotation speed o. Furthermore, it is judged whether or not the biggest inter-phase voltage Vx is larger than the battery voltage Vb at step S 102. A plural-leg-switching mode is selected, when the biggest inter-phase voltage Vx is not larger than the battery voltage Vb. In the plural-leg-switching mode, the conventional PWM-switching method, of which two legs or three legs are PWM switched, is executed at step S 104.

[0093]

In step S 102, a single-leg- switching mode is selected, when the biggest inter-phase voltage Vx is larger than the battery voltage Vb. The control of the single-leg-switching mode is executed at step S 106 and S 108. At step S 106, one of the stages A-F shown in Figure 12 is selected in accordance with the detected rotor angle Θ. The controller 9 has a table showing a relation between the stages A"F and the detected rotor angle Θ.

[0094]

Next, at step S 108, the gate signals S i and S2 are calculated in accordance with the torque instruction value Tr, the rotation speed ω and the rotor angle Θ. The gate signals S I shows a duty ratio of the converter 8. The gate signals S2 shows a duty ratio of the switched leg of inverter 4. The controller keeps a relation among the gate signals S i and S2, torque instruction value Tr, the rotation speed ω and the rotor angle Θ.

[0095]

The duty ratio of the switched-leg of inverter 4 is further explained. The switched leg outputs the smaller inter-phase voltage Vy. The relative duty ratio Dz of the switched leg is calculated in accordance with the detected rotor angle Θ and the relation between the rotor angle Θ and relative duty ratio Dz in order to output the smaller inter-phase voltage Vy. The relative duty ratio Dz is proportional to the ratio between the value Vy and the value Vx.

[0096]

The decided relative duty ratio Dz of the switched leg is given to the upper switch of the switched leg. The decided relative duty ratio 1 -Dz of the switched leg is given to the lower switch of the switched leg. The lower switch of the switched leg has the opposite motion to the upper switch of the switched leg.

[0097]

Figure 25 shows several waveforms of the biggest inter-phase voltage Vx. In Figure 25, the biggest inter-phase voltages Vx l, Vx2, Vx3, Vx4, Vx5 and Vx6 have different amplitudes to each other. These waveforms of the biggest inter-phase voltages are essentially equal to the waveform of the biggest inter-phase voltage Vx shown in Figure 13. However, the amplitudes of the voltages Vx l , Vx2, Vx3, Vx4, Vx5 and Vx6 are different to each other, because the instruction values of the motor torque are different to each other. The period Ty is a period when the voltage Vx l is higher than battery voltage Vb. The period Tx is a period when the voltage Vx l is lower than battery voltage Vb. The SPSM is operated in the period Ty. The conventional PWM-switching method is operated in the period Tx.

[0098]

For example, the SPSM is operated, when the biggest inter-phase voltage Vx is higher value, which is 231V-700V, than battery voltage Vb, which is 230V, As the result, the motor-driving apparatus of the embodiment can control the motor, even though the torque and the rotation speed are small.

[0099]

(The third embodiment)

Figure 26 shows a circuit diagram showing the SPSM-operated motor-driving apparatus driven by the series-parallel-changing method (SPCM) . The SPCM is one embodiment of the turn-number-changing method (TNCM) explained above. Figure 26 shows a nine- switch inverter 4 driving a three-phase motor of variable-speed type, but illustration of the boost DC/DC converter explained in the third embodiment is abbreviated. The biggest inter-phase voltage Vx as the DC-link voltage is applied to the nine-switch inverter 4. The three -phase stator winding 6 of the motor 6 has two three-phase windings 6A and 6B . The star-connected three-phase winding 6A has a U-phase winding A, a V-phase winding B and a W-phase winding C. The star-connected three-phase winding 6B has a U-phase winding D, a V-phase winding E and a W-phase winding F.

[0100]

The nine-switch inverter 4 shown in Figure 26 consists of three legs 1 - 3 connected in parallel to each other. Each of the legs 1- 3 has an upper switch X, a middle switch Y and a lower switch Z, which are connected in series to each other. In the other words, the U-phase leg 1 consists of the upper switch UX, the middle switch UY and the lower switch UZ. The V-phase leg 2 consists of the upper switch VX, the middle switch VY and the lower switch VZ. The W-phase leg 3 consists of the upper switch WX, the middle switch WY and the lower switch WZ. Three upper switches X and three lower switches Z are connected to the DC/DC converter through the high potential line 100 and the low potential line 101. The three-phase winding 6A is connected between the three upper switches X and the three middle switches Y. The three-phase winding 6B is connected between the three middle switches Y and the lower switches Z.

[0101]

The parallel operation of nine - switch inverter 4 is explained referring to Figures 26 and 27. The parallel operation has six periods TA-TF. Figure 26 illustrates three states of inverter 4 in the periods TA, TB and TC. Figure 27 illustrates three states of the inverter 4 in the periods TD, TE and TF. Each of six periods TA-TF is a period having the electric angle of 60 degrees (n/3). Two three -phase voltages applied to the windings 6A and 6B have the sinusoidal waveform shown in Figure 12. In the periods TA-TF, three middle switches Y are turning on in the parallel operation. The upper switch X and the lower switch Z of the same phase have an opposite state to each other. For example, the switch UX is turned on, when the switch UZ is turned off. The switch UX is turned off, when the switch UZ is turned on. By this motion, short-circuiting currents are protected.

[0102]

The period TA is a period from 0 degree to 60 degree . The switch WZ is always turned on. In a first half of the period TA (0 degree- 30 degree) , the switch VX is always turned on. In a second half of the period TA (30 degree-60 degree), the switch UX is always turned on. The period TB is a period from 60 degree to 120 degree . The switch UX is always turned on. In a first half of the period TB (60 degree-90 degree), the switch WZ is always turned on. In a second half of the period TB (90 degree- 120 degree), the switch VZ is always turned on . The period TC is a period from 120 degree to 180 degree. The switch VZ is always turned on. In a first half of the period TC (120 degree- 150 degree), the switch UX is always turned on. In a second half of the period TC (150 degree- 180 degree), the switch WX is always turned on.

[0103]

The period TD is from 180 degree to 240 degree. The switch WX is always turned on. In a first half of the period TD (180 degree-210 degree), the switch VZ is always turned on. In a second half of the period TD (210 degree-240 degree), the switch UZ is always turned on. The period TE is a period from 240 degree to 300 degree. The switch UZ is always turned on. In a first half of the period TE (240 degree-270 degree), the switch WX is always turned on. In a second half of the period TE (270 degree- 300 degree), the switch VX is always turned on. The period TF is a period from 300 degree to 360 degree. The switch VX is always turned on. In a first half of the period TF (300 degree-330 degree), the switch UZ is always turned on. In a second half of the period TF (330 degree- 360 degree), the switch WZ is always turned on.

[0104]

Two three -phase currents supplied to two three-phase windings 6A and 6B are changed. The changing timings are shown in Figure 10. In the other words, three upper switches X and the three lower switches Z are driven with the timing scheme shown in Figures 9 and 10. As the result, only two switches of nine -switch inverter 4 are PWM-switched at a time so as to generate the three-phase sinusoidal wave -form in the parallel operation. Real lines with arrows in inverter 4 shown in Figures 26 and 27 show free-wheeling currents in the first half of the periods TA'TF. Broken lines with an arrow in inverter 4 shown in Figures 26 and 27 show free-wheeling currents in the second half of the periods TA-TF. The inverter 4 supplies the same three-phase voltage to both three -phase windings 6A and 6B.

[0105] The series operation of nine-switch inverter 4 is explained referring to Figures 28- 30. Figure 28 illustrates four states of inverter 4 in the periods TA', TB', TC and TD'. Figure 29 illustrates four states of inverter 4 in the periods TE\ TF\ TG' and TH'. Figure 40 illustrates four states of inverter 4 in the periods ΤΓ, TJ\ TK' and TL'. Each of twelve periods TA'-TL' is a period having the electric angle of 30 degrees (n/6) . The three -phase voltages applied to the windings 6A and 6B are shown in Figure 22. In the periods TA'-TL', one middle switch Y is always turned on. The leg with the turned-on middle switch Y is called the fixed leg. Another one leg of three legs 1 " 3 has the turned-on upper switch X and the turned-on lower switch Z.

[0106]

The leg with the turned-on upper switch X and the turned-on lower switch Z is called the fixed leg, too. The other one leg of three legs is PWM-switched. Combination of the two fixed legs and the one switched leg is changed in turn. In the period TL' and TA', U-phase leg 1 is PWM- switched. The switches VX, VZ and WY are turned on. In the periods TB' and TC, V-phase leg 2 is PWM- switched. The switches UX, UZ and WY are turned on. In the periods TD' and ΤΕ', W-phase leg 3 is PWM-switched. The switches UX, UZ and VY are turned on. In the periods TF' and TG', U-phase leg 1 is PWM-switched. The switches WX, WZ and VY are turned on. In the periods TH' and ΤΓ, V-phase leg 2 is PWM-switched. The switches WX, WZ and UY are turned on. In the periods TJ' and TK', W-phase leg 3 is PWM-switched. The switches VX, VZ and UY are turned on.

[0107]

As the result, inverter 4 connects two three-phase windings 6A and 6B in series to each other. Three-phase windings 6A and 6B are driven with the SPSM explained in the third embodiment. Inverter 4 supplies the same three-phase current to both of three -phase windings 6A and 6B connected in series to each other. Each of real lines with an arrow in inverter 4 shown in Figures 28-30 shows each of free -wheeling currents in the periods TA'-TL'.

[0108]

The SPCM (series-parallel-changing method) with nine-switch inverter 4 has simple structure. Furthermore, the nine-switch inverter 4 driven with both of the SPCM and the SPSM reduces a number of the PWM-switched transistors switched at the same period, even though inverter 4 has nine transistors. Nine -switch inverter 4 shown in Figures 28- 30 can employ the other known switching method. For example, inverter 4 can be switched with the three -phase PWM method or the spontaneous sp ace vector method or the two-phase modulation method. In the parallel operation, all middle switches are turned on. Three upper switches X and three lower switches Z are switched with these known switching methods adopted for the conventional six-switch three -phase inverter. In the series operation, at least one middle switch is turned off. At least another one middle switch is turned on. The current of nine-switch inverter 4 flows through the second three-phase winding 6B after flowing through the first three-phase winding 6A.

[0109]

(A turn-number-changing method (TNCM))

The important aspect of the above series-parallel-changing method (SPCM) of nine -switch inverter 4 is explained. The three-phase current of three-phase winding 6B in the series-connection shown in Figures 28- 30 flows oppositely in comparison with the three-phase current of three-phase winding 6B in the parallel-connection shown in Figures 26 and 27. In this embodiment, the IJ-phase windings A and D are wound on the same stator poles. Furthermore, a turn number of U-phase winding A is different from a turn number of U-phase winding D. Current directions of U-phase windings A and D connected in parallel as shown in Figures 26 and 27 are same . [0110]

Accordingly, current directions of U-phase windings A and D connected to series as shown in Figures 28- 30 are opposite to each other. Relation between V-phase windings B and E are same as the above U-phase windings A and D. Relation between the W-phase windings C and F are same as the above U-phase windings A and D, too. For example, U-phase winding A has 300 turns and the U-phase winding D has 200 turns. In Figures 26 and 27, the parallel-connected windings A and D are mostly equal to a U-phase winding with 250 (= (300-200)/2) turns.

[0111]

In Figures 28'30, the series-connected U-phase winding A and D are mostly equal to a U-phase winding with 100 (=300-200) turns. After all, it is considered that the turn number of the winding is changed by nine- switch inverter 4 employing the SPCM (series-parallel-changing-method), when the two phase windings with same phase have different turn number to each other.

[0112]

(A pole-number-changing method)

Another method for solving the current-direction-changing problem of the second inverter 6B is explained referring to Figures 31 - 36. This method is called the pole-number-changing method. Figures 31 is an equivalent circuit view showing the parallel connection of two three-phase windings 6A and 6B shown in Figures 26 and 27 for the period TB. Figures 35 is an equivalent circuit view showing the series connection of two three-phase windings 6A and 6B shown in Figures 28-30 for the periods TC and TD'.

[0113]

In the pole-number-changing method, each one of six phase windings A-F having an equal turn number each are wound around each one of stator poles (stator teeth) respectively as shown in Figures 33- 36. In Figures 33- 34, each one of three phase windings A- C of the three-phase winding 6A is wound around each one of odd stator poles. Similarly, each one of three phase windings D -F of the three-phase winding 6B is wound around each one of even stator poles.

[0114]

In Figures 33- 36, a stator core 1000 has teeth 1001, which are stator poles, connected with a back core 1002 to each other. The arrow lines A-F illustrated on the teeth 1001 shows six phase windings. The single arrow shows the direction of the current with smaller amplitude. The dual arrow shows the direction of the current with larger amplitude. Figure 33 shows six phase currents in the period TB from 60 degree to 120 degree. Figure 34 shows six phase currents in the period TC from 120 degree to 180 degree. In Figures 33 and 34, it is considered that the electrical angle of 360 degrees is equal to six stator teeth pitches.

[0115]

Figure 35 shows six phase currents in the periods TC and TD' from 60 degree to 120 degree . Figure 36 shows six phase currents in the periods TE' and TF ' from 120 degree to 180 degree. In Figures 35 and 36, it is considered that the electrical angle of 360 degrees is equal to three stator teeth pitches, because the flow directions of three phase currents of the three-phase winding 6B in the series connection are opposite in comparison with them in the parallel connection.

[0116]

As the result, the pole number of the stator 1000 is doubled by means of changing the connection from the parallel to the series. In the other words, by employing the series connection shown in Figures 28- 30 and Figures 35"36, both of the turn number and the stator pole number of the stator winding are doubled. The above SPCM (stator-pole-changing method) is preferably employed for the induction motor. The above SPCM can be employed, when the synchronous motor has a rotor which is capable to change a rotor pole number. The above pole -number-changing method can change the motor torque largely. The series connection can be adopted at the low rotating speed range preferably. The parallel connection can be adopted at the high rotating speed range preferably.

[0117]

The parallel operation of the nine-switch inverter driven with the SSVM is shown in Figures 26 and 27. The series operation of the nine-switch inverter driven with the SSVM is shown in Figures 37 and 38. Figure 39 shows one PWM carrier period Tp of the SSVM. The nine wave forms of the gate-voltages applied to the nine switches of the inverter 4 are shown in Figure 39.

[0118]

(The fourth embodiment)

The single-phase-switching method reducing the switching power loss of the inverter 4 needs the small smoothing capacitor 8D shown in Figure 16. However, the switching surge noise voltage generated by the upper switches 11, 21 and 31 shown in Figure 16 can not be suppressed by the small smoothing capacitor 8D. It causes increased surge noise voltage on the high potential bus line 100, because the line inductance of the high potential bus line 100 generates the surge noise voltage, when the upper switches 11, 21 and 31 are turned off. A surge-absorbing circuit 400 shown in Figure 40 can reduce the surge noise voltage on the high potential bus line 100. The surge -absorbing circuit 400 absorbs high-frequency components of the DC link voltage Vx, which is the biggest inter-phase voltage between two points A and B.

[0119] The surge -absorbing circuit 400 consists of the capacitor-chargers 4001-4002, gate drivers 4003-4005 and a series-connecting switch 403. The capacitor-charger 4001 consists of a capacitor 401 and a P-itype MOS transistor 404 connected to series. The source electrode of the transistor 404 is connected to the high-potential terminal A. The collector electrode of transistor 404 is connected to the low potential terminal B through the capacitor 401.

[0120]

Capacitor-charger 4002 consists of a capacitor 402 and an N-type MOS transistor 405 connected to series. The source electrode of the transistor 405 is connected to the low potential terminal B. The collector electrode of the transistor 405 is connected to the high potential terminal A through the capacitor 402.

[0121]

Gate driver 4003 consists of a capacitor 406 and a high-impedance element 407. Preferably, the high-impedance element 407 consists of an inductance element or a resistor element. The high-impedance element 407 is connected to capacitor 406 to series. One end of capacitor 406 is connected to the low potential terminal B. A voltage Vx between terminals A and B is applied to gate driver 4003. The contact point X of gate driver 4003 is connected to the gate electrode of the P-type MOS transistor 404.

[0122]

Gate driver 4004 consists of a capacitor 409 and a high-impedance element 410. Preferably, the high-impedance element 410 consists of an inductance element or a resistor element. The high-impedance element 410 is connected to capacitor 409 to series. One end of the capacitor 409 is connected to the high-potential terminal A. The voltage Vx is applied to gate driver 4004. The contact point Y of the gate driver 4004 is connected to the gate electrode of the N-type MOS transistor 405.

[0123]

The contact point S of the capacitor-charger 4001 and the contact point T of the capacitor-charger 4002 are connected by series-connecting switch 403. The series-connecting switch 403 consists of N-type MOS transistor, of which the source electrode is connected to the contact point T of the capacitor-charger 4002. The gate driver 4005 consists of a capacitor 412 and a high-impedance element 413. Preferably, the high-impedance element 413 consists of an inductance element or a resistor element.

[0124]

The high-impedance element 413 is connected to capacitor 412 to series. One end of capacitor 412 is connected to the low potential terminal B . A voltage V3 of the contact point T is applied to gate driver 4005. The contact point Z of gate driver 4005 is connected to the gate electrode of N-type MOS transistor 403.

[0125]

In the other words, the surge-absorbing circuit 400 has two capacitors 401 and 402 and the series-connecting switch 403. Two capacitors 401 and 402 are charged through two transistors 404 and 405 separately. A potential of the high potential terminal A falls down by charging of capacitors 401 and 402. When the series-connecting switch 403 connects two capacitors 401 and 402, two capacitors 401 and 402 are discharged to the high potential terminal A. By means of discharging of capacitors 401 and 402, a potential of the high potential terminal A increases.

[0126]

It is considerable to two conditions are required. The first condition is that two transistors 404 and 405 must turn on and transistor 403 must turn off, when the potential of the high potential terminal A rises up rapidly. The second condition is that two transistors 404 and 405 must turn off and transistor 403 must turn on, when the potential of the high potential terminal A falls down rapidly. Gate drivers 4003-4005 controls transistors 403-405 separately.

[0127]

Motions of gate drivers 4003-4005 are explained. High-impedance elements 407, 410 and 413 and capacitors 406, 409 and 410 connected to series one another consist of a low-pass filter each. When the voltage Vx does not change rapidly, the gate voltages VI is almost equal to the voltage Vx, the gate voltages V4 is almost equal to zero V and the gate voltages V5 is almost equal to the voltage V3. Firstly, the motion when the voltage Vx rises up rapidly is explained. The gate potential VI is almost constant but the source potential of P-type MOS transistor 404 rises up rapidly. As the result, P-type MOS transistor 404 turns on.

[0128]

The capacitor 401 absorbs the charging current from the high-potential terminal A. Similarly, by rapidly-rising-up of the gate potential V4, the N-type MOS transistor 405 turns on. The capacitor 402 absorbs the charging current from the high-potential terminal A. By rapidlyrising-up of the source potential V3, the N-type MOS transistor 403 turns off. As the result, the rapidlyrising-up of the voltage Vx is suppressed.

[0129]

Next, motions when the voltage Vx falls down rapidly are explained. The gate potential VI is almost constant but the source potential of the P-type MOS transistor 404 falls down rapidly. As the result, P-type MOS transistor 404 turns off. The capacitor 401 is cut off from the high-potential terminal A. Similarly, by rapidly-falling-down of the gate potential V4, the N-type MOS transistor 405 turns off. The capacitor 402 is cut off from the low potential terminal B .

[0130]

By rapidly-falling-down of the source potential V3, the N-type MOS transistor 403 turns on. Capacitors 401 and 402 connected to series through the N-type MOS transistor 403 are discharged. As the result, the rapidly-falling-down of the voltage Vx is suppressed.

[0131]

For example, an average value of the voltage Vx is 650 V and the turning-on-threshold voltages of transistors 403-405 are 2V. The positive peak of the surge voltage is 32.5V. If the surge-absorbing circuit 400 is not connected to the high potential terminal A, the voltage Vx becomes 682.5V. If the surge-absorbing circuit 400 is connected to the high potential terminal A, the voltage Vx becomes less than 660V. Similarly, the negative peak of the surge voltage is - 32.5V. If the surge -absorbing circuit 400 is not connected to the high potential terminal A, the voltage Vx becomes 617.5V. If the surge -absorbing circuit 400 is connected to the high-potential terminal A, the voltage Vx becomes more than 640V.

[0132]

The surge-absorbing circuit 400 permits smaller capacitors 401-402 for charging and discharging of the high-potential terminal A than the conventional smoothing capacitor. Because, voltage changes of the capacitors 401 -402 are very large . Consequently, the small surge absorber is realized. The other advantage of the surge-absorbing circuit 400 is good reliability and less DC power loss. Surge -absorbing circuit 400 shown in Figure 20 does not have DC current paths. All currents flow from the high potential terminal A to the low potential terminal B through capacitors 401, 402, 406, 409, 410 and 412.

[0133] Consequently, the voltage ripples generated by the PWM-switching with the high frequency is absorbed by the surge-absorbing circuit 400. However, changing of the biggest inter-phase voltage Vx with a low frequency is not prevented by the surge -absorbing circuit 400.

[0134]

The above-explained gate drivers 4003-4005 are changed to the other known gate driver circuit. For example , a comparator including a window comparator can be adopted instead of gate drivers 4003-4005. The gate driver circuit compares the voltage Vx of the high-potential terminal A, and a DC voltage component of the voltage Vx. The comparator controls the MOS transistors 403-405 in accordance with the result.

Claims

Claim 1

A motor-driving apparatus for driving a three-phase motor of variable speed type comprising:

an inverter (4) applying a three-phase voltage to a stator winding (6) of the three-phase motor consisting of an induction motor;

the stator winding has a first three-phase winding (4A) and a second three-phase winding (4B);

the first three -phase winding (4A) has a U-phase winding (U l), a V-phase winding (Vl) and a W-phase winding (W l) ;

the second three-phase winding (4B) has a TJ-phase winding (U2), a V-phase winding (V2) and a W-phase winding (W2);

a controller of controlling the inverter (4) having a U-phase leg (l), a V-phase leg (2) and a W-phase leg (3), which are connected in parallel to each other between a high potential bus line ( 100) and a low potential bus line ( 101); the U-phase leg ( l) has an upper switch (ll), a middle switch ( 13) and a lower switch ( 12) connected in series to each other;

the V-phase leg (2) has an upper switch (21), a middle switch (23) and a lower switch (22) connected in series to each other;

the W-phase leg (3) has an upper switch (31), a middle switch (33) and a lower switch (32) connected in series to each other;

the first three -phase winding (4A) is connected between an upper group of the upper switches (11, 21 and 31) and a middle group of the middle switches (13, 23 and 33) ; and

the second three-phase winding (4B) is connected between an the middle group of the middle switches ( 13, 23 and 33) and a lower group of the lower switches ( 12, 22 and 32) ;

wherein the first three -phase winding (4A) and the second three-phase winding (4B) are wound on a stator core of the one motor,^'

the controller changes turns of a stator winding consisting of the first three-phase winding (4A) and the second three-phase winding (4B) by means of selecting one of a series connection mode and a parallel connection mode both of same phase windings of the first three -phase winding (4A) and the second three -phase winding (4B) are connected in series to each other in the series connection mode,' and

both of same phase windings of the first three-phase winding (4A) and the second three-phase winding (4B) are connected in parallel to each other in the parallel connection mode.

Claim 2

The motor-driving apparatus according to claim 1, wherein the controller selects the series connection mode in a low speed range, and selects the parallel connection mode in a high speed range.

Claim 3

The motor-driving apparatus according to claim 3, wherein the six phase windings (U l, V2, W l, U2, VI and W2) arranged in turn in an electrical angle 4nin the series connection;

the six phase windings (U l , -V2, W l, -U2, V I and -W2) arranged in turn in an electrical angle 2nin the parallel connection; and

the series connection mode makes double poles of the stator poles in comparison with the parallel mode .

Claim 4

The motor-driving apparatus according to claim 3, wherein the six phase windings (U l , V2, W l, U2, VI and W2) has equal turns to each other.

Claim 5

A motor-driving apparatus for driving a three-phase motor of variable speed type comprising^ an inverter (4) having three legs ( 1 - 3) connected in parallel in order to apply a three-phase voltage to the motor (6) ;

a boost DC/DC converter (8) applying a boost voltage to the inverter (4) via a high potential bus line (100) and a low potential bus line (101),' and a controller (9) controlling the inverter (4) and the converter (8);

wherein the inverter (4) has one switched leg, a first fixed leg and a second fixed leg;

each of the legs (1 - 3) has an upper switch (11 , 21, and 31) and a lower switch (12, 22, and 32) connected to series_^

the boost DC/DC converter (8) applies a biggest inter-phase voltage (Vx) of the three-phase voltage to the motor (6) via the two fixed legs!

the switched leg applies a smaller inter-phase voltage (Vy) which is smaller than the biggest inter-phase voltage (Vx) by means of switching of the upper switch ( 11 , 2 1 , 31) and the lower switch ( 12, 22, 32) ;

the controller (9) has a single-leg-switching mode changing the switched leg and the two fixed legs in order,'

the first fixed leg has the turned-on upper switch ( 11, 21, and 31) and the turned-off lower switch ( 12, 22, and 32) in the single-leg-switching mode! the second fixed leg has the turned-off upper switch ( 11, 21, 31) and the turned-on lower switch (12, 22, 32) in the single-leg-switching mode; and the controller (9) controls amplitude and a waveform of the biggest inter-phase voltage (Vx) in accordance with a value of a motor current and a value of a rotor angle of the motor (6) .

Claim 6

The motor-driving apparatus according to claim 5;

wherein the controller (9) further has a plural-leg-switching mode for controlling the boost DC/DC converter (8) and the inverter (4);

at least two legs ( 1 -3) of the inverter (4) are switched with a PWM method in the plural-leg-switching mode !

the controller (9) selects the single-leg-switching mode when the biggest inter-phase voltage (Vx) is larger than a voltage of the power supply apparatus (7) ; and

the controller (9) selects the plural-leg-switching mode when the biggest inter-phase voltage (Vx) is smaller than the voltage of the power supply apparatus (7) .

Claim 7

The motor-driving apparatus according to claim 5!

wherein the biggest inter-phase voltage (Vx) has a three -phase-full-wave -rectified wave form; and

the inverter (4) outputs a three -phase sinusoidal voltage of which a frequency is changed in accordance with the rotation speed of the motor (6) .

Claim 8

The motor-driving apparatus according to claim 5;

wherein the boost DC/DC converter (8) consists of a chopper type DC/DC converter (8) with a reactor (8C) and a half bridge of which an upper switch (8E) and a lower switch (8F) are connected to series,' and

the controller (9) changes a PWM duty ratio of the boost DC/DC converter (8) in the single-leg-switching mode in accordance with a received torque instruction value (Tr) and a detected rotation speed (ω) of the motor (6) . Claim 9

The motor-driving apparatus according to claim 5;

the high potential bus line (100) and the low potential bus line ( 101) are connected by a surge-absorbing circuit (400);

the surge-absorbing circuit (400) has capacitor-chargers (4001 -4002), gate drivers (4003-4005) and a series-connecting transistor (403);

the capacitor-charger (4001) has a capacitor (401) and a transistor (404) connected to series;

the capacitor-charger (4002) has a capacitor (402) and a transistor (405) connected to series;

the series-connecting transistor (403) connects a contact point (S) between the capacitor (401) and the transistor (404) to a contact point (T) between the capacitor (402) and the transistor (405) ;

the capacitor-chargers (4001-4002) are connected between two terminals (A and B) each;

the gate drivers (4003-4005) turn on the transistors (404-405) and turn off the transistor (403), when an voltage (Vx) between two terminals (A and B) is increased; and

the gate drivers (4003-4005) turn off the transistors (404-405) and turn on the transistor (403), when the voltage (Vx) between two terminals (A and B) is decreased.