US20010001535A1 - Multi-mode power converters incorporating balancer circuits and methods of operation thereof - Google Patents
Multi-mode power converters incorporating balancer circuits and methods of operation thereof Download PDFInfo
- Publication number
- US20010001535A1 US20010001535A1 US09/756,505 US75650501A US2001001535A1 US 20010001535 A1 US20010001535 A1 US 20010001535A1 US 75650501 A US75650501 A US 75650501A US 2001001535 A1 US2001001535 A1 US 2001001535A1
- Authority
- US
- United States
- Prior art keywords
- voltage
- switches
- bus
- switch
- approximately
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Images
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J9/00—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting
- H02J9/04—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source
- H02J9/06—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems
- H02J9/062—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems for AC powered loads
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J9/00—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting
- H02J9/04—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source
- H02J9/06—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems
- H02J9/061—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems for DC powered loads
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
- H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
- H02M5/42—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters
- H02M5/44—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac
- H02M5/453—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal
- H02M5/458—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M5/4585—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only having a rectifier with controlled elements
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J9/00—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting
- H02J9/04—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source
- H02J9/06—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems
- H02J9/062—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems for AC powered loads
- H02J9/063—Common neutral, e.g. AC input neutral line connected to AC output neutral line and DC middle point
Definitions
- the present invention relates to electrical power devices and methods of operation thereof, and more particularly, to power conversion devices and methods of operation thereof.
- UPSs Uninterruptible power supplies
- UPSs are power conversion devices that are commonly used to provide conditioned, reliable power for computer networks, telecommunications networks, medical equipment and the like. UPSs are widely used with computers and similar computing devices, including but not limited to personal computers, workstations, mini computers, network servers, disk arrays and mainframe computers, to insure that valuable data is not lost and that the device can continue to operate notwithstanding temporary loss of an AC utility source. UPSs typically provide power to such electronic equipment from a secondary source, such as a battery, in the event that a primary alternating current (AC) utility source drops out (blackout) or fails to provide a proper voltage (brownout).
- AC primary alternating current
- a typical off-line UPS disconnects a load from a primary AC source 10 when the primary AC source fails or is operating in a degraded manner, allowing the load to be served from a secondary source such as a battery.
- the AC power source 10 is connected in series with a switch S 1 , producing an AC voltage across a load 20 when the switch S 1 is closed, Energy storage is typically provided in the form of a storage capacitor C S .
- the secondary power source here a battery B, is connected to the load 20 via a low voltage converter 30 and a transformer T.
- the switch S 1 is opened, causing the load to draw power from the battery B.
- the low voltage converter 30 typically is an inverter that produces a quasi-square wave or sine wave voltage on a first winding L 1 of the transformer T from a DC voltage produced by the battery B.
- the first winding L 1 is coupled to a second winding L 2 of the transformer T connected across the load 20 .
- the battery B may be charged using the low-voltage converter 30 or a separate battery charger circuit (not shown).
- FIG. 2 A line interactive (LIA) UPS topology is illustrated in FIG. 2.
- the transformer T has a third winding L 3 that may be connected in series with the load 20 using switches S 2 , S 3 to “buck” or “boost” the voltage applied to the load 20 .
- the switch S 1 can be opened to allow the load 20 to run off the battery B.
- a typical on-line UPS includes a rectifier 40 that receives an AC voltage from an AC power source 10 , producing a DC voltage across a storage capacitor C S at an intermediate node 45 .
- An inverter 50 is connected between the intermediate node 45 , and is operative to produce an AC voltage across a load 20 from the DC voltage.
- a battery B is connected to the intermediate node 45 via a DC/DC converter 60 , supplying auxiliary power.
- the DC/DC converter can be eliminated and a high-voltage battery (not shown) connected directly to the intermediate node 45 .
- Each of these topologies may have disadvantages.
- typical conventional on-line and LIA UPSs for 60 Hz applications use 60 Hz magnetic components (e.g., transformers and inductors) that are sized for such frequencies, and thus may be large, heavy and expensive.
- LIA UPSs often exhibit step voltage changes that can affect the performance of the load.
- Conventional off-line, LIA and on-line UPSs often use large storage capacitors, which tend to be bulky and expensive, in order to maintain an acceptable output voltage under heavy loading conditions.
- UPSs are typically designed to operate in only one of the above-described off-line, LIA or on-line modes
- sellers of UPSs may be required to maintain large inventories including several different types of UPSs in order to meet a variety of different customer applications.
- a rectifier circuit produces first and second voltages (e.g., ⁇ DC voltages) on first and second voltage busses from an AC input voltage produced by an AC power source
- an inverter circuit produces an AC output voltage from the first and second voltages
- a balancer circuit controls the relative magnitudes of the first and second voltages responsive to respective first and second rates at which the balancer circuit couples the first and second voltage busses to a neutral bus of the AC power source.
- the rectifier circuit includes first and second switches that selectively couple the first and second voltage busses to a phase bus of the AC power source through a first inductance
- the inverter circuit includes third and fourth switches that selectively couple the first and second voltage busses to a load through a second inductance
- the balancer circuit includes fifth and sixth switches that selectively couple the first and second voltage bussed to the neutral bus through a third inductance, such as an inductor or transformer winding.
- the balancer circuit enables energy transfer between first and second storage capacitors connected between the neutral bus and the first and second voltage busses, respectively, thus allowing the storage capacitors to be smaller than the storage capacitors typically used in conventional power converters with comparable power ratings.
- the switches in the rectifier, inductor and balancer can be controlled such that the power converter can be operated in a number of different power transfer modes.
- a secondary power source such as a battery, may also be coupled to the power converter via a winding of a transformer that also serves as an inductance for the balancer circuit.
- this coupling may be achieved through a combination battery converter/battery charger circuit that can also charge the battery when the converter is running off an AC power source.
- switches in the balancer circuit can be operated at varying duty cycles in positive and negative half-cycles of the AC input voltage, which can allow the power converter to be operated in a more efficient manner.
- a power converter includes first and second voltage busses and a neutral bus.
- a first switching circuit e.g., a rectifier circuit, is operative to selectively couple an input node thereof to the first and second voltage busses.
- a balancer circuit is operative to selectively couple the neutral bus to the first and second voltage busses such that relative magnitudes of respective first and second voltages on the first and second voltage busses are controlled responsive to respective first and second rates at which the balancer circuit couples the first and second voltage busses to the neutral bus.
- a second switching circuit e.g., an inverter circuit, is operative to selectively couple the first and second voltage busses to a load at an output node thereof.
- the balancer circuit preferably includes first and second switches operative to selectively couple respective ones of the first and second voltage busses to the neutral bus through an inductance, such that the relative magnitudes of the first and second voltages are controlled responsive to respective first and second duty cycles of the first and second switches.
- the balancer circuit preferably is responsive to an AC input voltage applied to the input node to vary the respective duty cycles at which the first and second switches operate.
- a battery converter circuit may be switchably coupled to at least one of the first and second voltage busses.
- the battery converter circuit may include an inductor configured to be connected in series with a battery.
- a first switch is operative to selectively couple one end of a series combination of a battery and the inductor to one of the first or second voltage busses.
- a second switch is operative to selectively couple the one end of the series combination of a battery and an inductor to another end of the series combination of a battery and the inductor.
- the balancer circuit includes a transformer including a first winding having a first tap coupled to the first and second switches and a second tap coupled to the neutral bus.
- the first and second switches selectively couple the first tap of the transformer to the first and second voltage busses.
- the power converter may further include first and second diodes, the first diode having a cathode coupled to the first voltage bus, the second diode having a cathode coupled to an anode of the first diode and an anode coupled to the second voltage bus.
- the first winding of the transformer may have a first end tap coupled to the first and second switches, a second end tap coupled to the anode of the first diode and the cathode of the second diode, and a center tap coupled to the neutral bus.
- the converter may further include third and fourth switches.
- the first winding of the transformer may have a first end tap coupled to the first and second switches, a second end tap coupled to the third and fourth switches, and a center tap coupled to the neutral bus, wherein the third switch is operative to couple and decouple the second end tap of the first winding and the first voltage bus and the fourth switch is operative to couple and decouple the second end tap of the first winding and the second voltage bus.
- a second winding of the transformer is inductively coupled to the first winding.
- An AC voltage generating circuit is coupled to the second winding of the transformer and operative to apply an AC voltage thereto.
- the AC voltage generating circuit may include a battery converter circuit operative to generate an AC voltage on the second winding of the transformer from a DC voltage produced by a battery coupled to the battery converter circuit.
- the AC voltage generating circuit may include a combined battery converter/battery charger circuit that is operative to generate an AC voltage on the second winding of the transformer from a DC voltage produced by the battery, and to produce a DC voltage across the battery from an AC voltage induced on the second winding of the transformer.
- a power converter includes a rectifier circuit configured to connect to an AC power source and operative to produce first and second DC voltages at first and second voltage busses, respectively, by selectively coupling the first and second voltage busses to the AC power source through a first inductance.
- First and second capacitors couple the first and second voltage busses, respectively, to a neutral bus.
- An inverter circuit is configured to connect to a load and operative to selectively couple the first and second voltage busses to the load through a second inductance.
- a balancer circuit is operative to selectively couple the first and second voltage busses to the neutral bus through a third inductance such that relative magnitudes of the first and second DC voltages are controlled responsive to respective first and second rates at which the balancer circuit couples the first and second voltage busses to the neutral bus.
- the power converter includes a switch control circuit.
- the rectifier circuit includes a first inductor having a first terminal configured to receive an AC input voltage, a first switch responsive to the switch control circuit to couple and decouple a second terminal of the first inductor and the first voltage bus, and a second switch responsive to the switch control circuit to couple and decouple the second terminal of the first inductor and the second voltage bus.
- the inverter circuit includes a second inductor having a first terminal configured to connect to a load, a third switch responsive to the switch control circuit to couple and decouple a second terminal of the second inductor and the first voltage bus, and a fourth switch responsive to the switch control circuit to couple and decouple the second terminal of the second inductor and the second voltage bus.
- the balancer circuit may include a third inductor having a first terminal coupled to the neutral bus, a fifth switch responsive to the switch control circuit to couple and decouple a second terminal of the third inductor and the first voltage bus, and a sixth switch responsive to the switch control circuit to couple and decouple the second terminal of the third inductor and the second voltage bus.
- First and second voltage busses are selectively coupled to the phase bus through a first inductance to produce first and second DC voltages at first and second voltage busses, respectively.
- the first and second voltage busses are selectively coupled to the load through a second inductance.
- the first and second voltage busses are selectively coupled to the neutral bus through a third inductance such that relative magnitudes of the first and second DC voltages are controlled responsive to respective first and second rates at which the first and second voltage busses are coupled to the neutral bus.
- the first and second voltage busses are selectively coupled to phase bus by switching a first switch to couple and decouple the phase bus and the first voltage bus through a first inductor and by switching a second switch to couple and decouple the phase bus and the second voltage bus through the first inductor.
- the first and second voltage busses are selectively coupled to the load by switching a third switch to couple and decouple the first voltage bus and the load through a second inductor and by switching a fourth switch to couple and decouple the second voltage bus and the load bus through the second inductor.
- the switching is preferably performed responsive to the AC input voltage.
- the respective duty cycles at which the first, second, third and fourth switches are operated are varied responsive to the AC input voltage.
- a substantially continuous low impedance connection between the phase bus and the load is provided through selected combinations of the first, second, third and fourth switches when the AC input voltage is approximately at a nominal level.
- the first and second switches are switched to boost the magnitudes of the first and second DC voltages while providing respective substantially continuous low impedance connections between the load and respective ones of the first and second voltages busses through respective ones of the third and fourth switches during respective positive and negative half-cycles of the AC input voltage.
- respective substantially continuous low-impedance connections between the AC power source and respective ones the first and second voltage busses are provided through respective ones of the first and second switches during respective positive and negative half-cycles of the AC input voltage, while bucking a voltage generated at the load from the first and second DC voltages.
- fifth and sixth switches that couple and decouple respective ones of the first and second voltage busses and the neutral bus through a third inductor are switched such that the magnitude of the first DC voltage is substantially greater than the magnitude of the second DC voltage during a positive half-cycle of the AC input voltage and such that the magnitude of the second DC voltage is substantially greater than the magnitude of the first DC voltage during a negative half-cycle of the AC input voltage.
- a battery may be selectively coupled to at least one of the first inductance, the first voltage bus, or the second voltage bus to enable power transfer between the battery and the first and second voltage busses.
- FIGS. 1 - 3 are schematic diagrams of power conversion circuit topologies used in typical conventional uninterruptible power supplies (UPSs).
- UPSs uninterruptible power supplies
- FIGS. 4 - 5 are schematic diagrams illustrating power converters according to embodiments of the present invention.
- FIGS. 6 - 8 are waveform diagrams illustrating exemplary operations of a power converter according to the embodiment of FIG. 4.
- FIGS. 9 - 10 are schematic diagrams illustrating power converters according to other embodiments of the present invention.
- FIGS. 11 - 12 are waveform diagrams illustrating exemplary operations for a power converter of FIG. 10.
- FIG. 13 is a schematic diagram illustrating a power converter according to another embodiment of the present invention.
- FIG. 14 is a waveform diagram illustrating exemplary operations for a power converter of FIG. 12.
- FIG. 15 is a schematic diagram illustrating a power converter according to another embodiment of the present invention.
- FIG. 16 is a schematic diagram illustrating an AC voltage generating circuit according to an embodiment of the present invention.
- FIG. 4 is a schematic diagram illustrating a power converter 400 according to an embodiment of the present invention.
- the power converter 400 includes a rectifier circuit 410 that is coupled to an AC power source 10 , producing first and second DC voltages V 1 , V 2 on respective first and second voltage busses 402 a, 402 b that are coupled to a neutral bus N by first and second capacitors C 1 , C 2 .
- the rectifier circuit 410 includes first and second switches 411 , 412 that selectively couple a phase bus 401 of the AC power source 10 to the first and second power busses 402 a, 402 b through a first inductor L 1 , responsive to a switch control circuit 440 .
- the power converter 400 includes an inverter circuit 420 that produces an AC output voltage V out across a load 20 (here shown as including capacitance C L and generalized impedance Z L ) at an output 403 from the first and second DC voltages at the first and second voltage busses 402 a, 402 b.
- the inverter circuit includes third and fourth switches 421 , 422 that selectively couple the first and second voltage busses 402 a, 402 b, respectively, to the load 20 through a second inductor L 2 , responsive to the switch control circuit 440 .
- the power converter 400 also includes a balancer circuit 430 that is operative to control the relative magnitudes of the first and second DC voltages V 1 , V 2 on the first and second voltage busses 402 a, 402 b by controlling respective first and second rates (e.g., duty cycles) at which the neutral bus N is connected to respective ones of the first and second voltage busses 402 a, 402 b through a third inductor L 3 .
- a “balancer” circuit is a circuit that is capable of effecting a desired “balance” between voltages on different busses, such as the first and second voltage busses 402 a, 402 b of FIG. 4.
- this capability can allow power converters according to embodiments of the present invention, among other things, to operate more efficiently and/or to utilize smaller components, e.g., storage capacitors, than those used in many conventional devices.
- this control is achieved by switching fifth and sixth switches 431 , 432 responsive to the switch control circuit 440 .
- the switches 431 , 432 preferably operate in a substantially complementary fashion (one on and one off at a given time), to constrain the voltages across the capacitors C 1 , C 2 . For example, if the switches 431 , 432 are each operated at 50% duty cycle, the voltages V 1 , V 2 at the first and second voltage busses 402 a, 402 b are constrained to be approximately equal.
- the inverter circuit 420 can allow current flow from the first and second voltage busses 402 a, 402 b to the load 20 , or vice versa, thus making the inverter circuit 420 a four-quadrant converter.
- the rectifier circuit 410 preferably has similar characteristics, except that different pulse-width modulation (PWM) patterns preferably are applied to the switches 411 , 412 of the rectifier circuit 410 than those applied to the switches 421 , 422 of the inverter circuit 420 .
- PWM patterns employed for the inverter circuit 420 preferably produce a voltage controlled, current limited output voltage, while the PWM patterns employed for the rectifier circuit 410 preferably provide a controlled current to and from the AC power source 10 .
- the rectifier circuit 410 can be operated such that a current is produced that causes power flow into the converter 400 from the AC power source 10 , or such that a current is produced that causes power flow into the AC power source 10 from the power converter 400 .
- FIG. 5 illustrates a power converter 500 according to another embodiment of the present invention. Portions of the converter 500 of FIG. 5 that are the same as those illustrated in FIG. 4 are denoted with like reference numerals, and further detailed discussion of their operations will not be provided in light of the preceding discussion of FIG. 4.
- diode protected transistor switches Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 6 are used for the switches 411 , 412 , 421 , 422 , 431 , 432 of FIG. 4.
- the transistors Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 6 operate under control of a switch control circuit 440 that includes a microcontroller 442 and a driver circuit 444 .
- the switch control circuit 440 may control the transistors Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 6 responsive to sensed AC input voltage V in,sensed and/or sensed AC output voltage V out,sensed .
- the switch control circuit 440 may operate using different types of control techniques.
- the switch control circuit 440 may utilize an “open loop” control technique wherein the operations of the transistors Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 6 are controlled based on a sensed AC input voltage V in,sensed , without reference to the output voltage V out produced by the converter 500 .
- the switch control circuit 500 may also operate on a “closed loop” basis, using a sensed AC output voltage V out,sensed to guide control of the operations of the transistors Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 6 .
- the sensed AC input voltage V in,sensed and sensed AC output voltage V out,sensed may be provided to the switch control circuit 440 in a number of different forms including, but not limited to, analog or digital representations of the input and output voltages V in , V out , or quantities related to and/or derived from the input and output voltages V in , V out . If battery-powered or battery-boosted operation is provided, as described below, the switch control circuit 440 may also operate based on a sensed battery voltage (not shown), which may include an analog or digital representation of an actual battery voltage, or a quantity related to and/or derived from such a battery voltage.
- FIG. 5 represents an exemplary implementation, and that other circuit implementations fall within the scope of the present invention.
- the switching functions of the transistors Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 6 may be provided by a variety of switching devices including, but not limited to, bipolar transistors, field-effect transistors (FETs), metal oxide semiconductor FETs (MOSFETs), gate turn-on devices (GTOs), and the like.
- the driver circuit 444 may include variety of different components as well, and preferably includes components suitable for controlling the particular type(s) of switching devices used.
- switching control circuit 440 may be implemented in a number of different ways within the scope of the present invention.
- functions of the microcontroller 442 may be implemented using discrete logic circuits or programmable logic circuits such as programmable logic devices (PLDs) instead or in conjunction with a microcontroller, microprocessor or similar device.
- functions of the microcontroller 442 and the driver circuit 444 may also be combined in one or more devices, such as an application-specific integrated circuit (ASIC) or a hybrid microcircuit.
- ASIC application-specific integrated circuit
- FIGS. 6 - 8 , 11 - 12 and 14 are waveform diagrams illustrating exemplary operations of power converters according to embodiments of the present invention.
- operation of the power converters of FIGS. 4, 10 and 13 will be described in terms of the control of the functions of switches 411 , 412 , 421 , 422 , 431 , 432 of the rectifier, inverter and balancer circuits 410 , 420 , 430 of the power converter 400 of FIG. 4 (and corresponding devices in the embodiments of FIGS.
- duty cycle generally refers to a percentage of time during a switching cycle period (corresponding to the switching frequency described above) that a switch is in a “closed,” i.e., conductive, state.
- a switch that is maintained at a 50% duty cycle is intermittently “on” for half of a switching cycle, while a switch approaching a 100% duty cycle, e.g., a 99% duty cycle, is on for nearly all of the switching cycle.
- the switches discussed may also operate at a 100% duty cycle, i.e., be maintained in an “on” state throughout one or more switching cycles, or at a 0% duty cycle, i.e., be maintained in an “off” state throughout one or more switching cycles.
- duty cycle control may be achieved, for example, by application of appropriate control signals to appropriate switching components.
- duty cycle control of the switching transistors Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 6 may be achieved by the microcontroller 442 and driver circuit 444 applying appropriate base drive signals to the switching transistors Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 6 . It will be understood, however, that such duty cycle control may be achieved using any of a number of other switching devices and control signal generating circuits.
- each of the first, second, third, fourth, fifth and sixth switches 411 , 412 , 421 , 422 , 431 , 432 of the rectifier, inverter and balancer circuits 410 , 420 , 430 of FIG. 4, and corresponding components of the embodiments of FIGS. 5 , 9 - 10 , 13 , and 15 are operated at one or more switching frequencies that are relatively high with respect to the frequency of the AC input voltage V in supplied to the converter 400 , more preferably, at one or more switching frequencies that each are at least 10 times higher than the frequency of the AC input voltage V in .
- the magnetic components e.g., the inductors L 1 , L 2 , L 3
- the magnetic components e.g., the inductors L 1 , L 2 , L 3
- a common switching frequency also may be used.
- Low pass filtering to produce a smoothed output voltage V out may be achieved by the combination of the output inductor L 2 and the capacitance C L .
- the output capacitance C L may be provided by the load 20 , as illustrated in FIG. 4, or may be incorporated in the converter 400 .
- the three switch pairs comprising the first and second switches 411 , 412 , the third and fourth switches 421 , 422 , and the fifth and sixth switches 431 , 432 , (and corresponding switching transistor pairs in the embodiments of FIGS. 5 , 9 - 10 , 13 , and 15 ) are operated in a “complementary” fashion.
- the second switch 412 of FIG. 4 is preferably generally constrained to be “off” when the first switch 411 is “on,” and vice versa.
- practical circuit implementations within the scope of the present invention may be used in which the “complementary” switches are operated in an approximately or substantially complementary fashion.
- a switch pair may be operated in a “break before make” fashion, such that one of the switches in the pair is turned off slightly before the other switch in the pair is turned on.
- a slight amount of overlap of “on” periods of switches of a complementary pair may be allowed, e.g., a “make before break” mode of operation.
- FIG. 6 illustrates exemplary operations of the converter 400 of FIG. 4 when the input voltage V in is at or near a desired level for the output voltage V out .
- the first switch 411 is operated at a duty cycle of approximately 100% (i.e., approaching a steady state “closed” state), while the second switch 412 is operated a complementary duty cycle of approximately 0% (i.e., approaching a steady state “open” state).
- the third switch 421 and the fourth switch 422 also operate at complementary duty cycles of approximately 100% and approximately 0%, respectively.
- the first DC voltage V 1 and the output voltage V out essentially track the input voltage V in .
- the balancer circuit 430 can be left in an inactive state, i.e., both the fifth and sixth switches can be left in an “off” state, the fifth switch 431 preferably is operated at a duty cycle of approximately 0% until the first DC voltage V 1 falls below a first threshold voltage V T1 while the sixth switch 432 is operated at a complementary duty cycle of approximately 100%.
- the duty cycles of the first, second, third and fourth switches 411 , 412 , 421 , 422 are changed.
- the first and second switches 411 , 412 are operated at complementary duty cycles of approximately 0% and approximately 100%, respectively, while the third and fourth switches 431 , 432 operate at complementary duty cycles of approximately 0% and approximately 100%, respectively.
- This provides a substantially continuous low-impedance connection between the input node 401 and the output node 403 via the second voltage bus 402 b , such that the second DC voltage V 2 and the output voltage V out essentially track the input voltage V in .
- the fifth switch 431 preferably is operated at a duty cycle of approximately 100% while the second DC voltage V 2 is less than second threshold voltage V T2 , with the sixth switch 432 being operating at a complementary duty cycle of approximately 0%.
- the switches 431 , 432 in the balancer circuit 430 can be used to smooth transition of the output voltage V out near zero volts, by varying the duty cycles of these switches between the threshold voltages V T1 , V T2 .
- the balancer circuit 430 begins to increase the duty cycle of the fifth switch 431 while decreasing the duty cycle of the sixth switch 432 in a complementary fashion, thus driving the second DC voltage V 2 negative before the actual zero crossing of the input voltage V in .
- the duty cycles of the fifth and sixth switches 431 , 432 are increased and decreased, respectively, such that by the time the second DC voltage V 2 becomes more negative than the second threshold voltage V T2 , the fifth and sixth switches are switching at duty cycles of approximately 100% and approximately 0%, respectively.
- This anticipatory generation of the second DC voltage V 2 allows the inverter circuit 420 to be switched such that a relatively smooth transition of the output voltage V out through zero volts can be achieved.
- Similar zero-crossing control can be achieved as the AC input voltage V in approaches zero volts during the negative half cycle 620 by varying the duty cycles of the fifth and sixth switches 431 , 432 in a complementary manner. As the second DC voltage V 2 increases above the second threshold voltage V T2 , the duty cycle of the fifth switch 431 is decreased while the duty cycle of the sixth switch 432 is increased, driving the first DC voltage V 1 in a positive direction before the zero crossing of the input voltage V in .
- the duty cycles of the fifth and sixth switches 431 , 432 are decreased and increased, respectively, such that by the time the first DC voltage V 1 exceeds the first threshold voltage V T1 , the fifth and sixth switches 431 , 432 are switching at complementary duty cycles of approximately 0% and approximately 100%, respectively.
- FIG. 7 illustrates exemplary operations of the converter 400 of FIG. 4 when the input voltage V in falls below a desired level for the output voltage V out .
- the first and second switches 411 , 412 are operated at complementary duty cycles sufficiently less than 100% and sufficiently greater than 0%, respectively, such that the action of the switches 411 , 412 and the first inductor L 1 boosts the first DC voltage V 1 above the input voltage V in .
- the third switch 421 and the fourth switch 422 operate at complementary duty cycles of approximately 100% and approximately 0%, respectively, providing a substantially continuous low-impedance connection between the first voltage bus 402 a and the output node 403 .
- the fifth and sixth switches 431 , 432 of the balancer circuit 430 are operated at complementary duty cycles of approximately 0% and approximately 100%, respectively.
- the duty cycles of the fifth and sixth switches 431 , 432 of the balancer circuit 430 can be increased and decreased, respectively, while the third and fourth switches 421 , 422 of the inverter 420 are modulated to produce a smoothed zero crossing for the output voltage V out .
- the duty cycles of the first, second, third and fourth switches 411 , 412 , 421 , 422 are changed.
- the first and second switches 411 , 412 are operated at complementary duty cycles sufficiently greater than 0% and sufficiently less than 100%, respectively, such that action of the first and second switches 411 , 412 and the first inductor L 1 drive the second DC voltage V 2 more negative than the input voltage V in .
- the third and fourth switches 421 , 422 operate at complementary duty cycles of approximately 0% and approximately 100%, respectively, providing a substantially continuous low-impedance connection between the second voltage bus 402 b and the output node 403 .
- the fifth and sixth switches 431 , 432 of the balancer circuit 430 are operated at complementary duty cycles of approximately 100% and approximately 0%, respectively, allowing the output voltage V out to essentially track the second DC voltage V 2 .
- the duty cycles of the fifth and sixth switches 431 , 432 of the balancer circuit 430 may be decreased and increased, respectively, while the third and fourth switches 421 , 422 of the inverter 420 are modulated to produce a smoothed zero crossing for the output voltage V out .
- FIG. 8 illustrates exemplary operations of the converter 400 of FIG. 4 when the input voltage V in exceeds a desired level for the output voltage V out .
- the first switch 411 is operated at a duty cycle of approximately 100%
- the second switch 412 is operated at a complementary duty cycle of approximately 0%, thus providing a substantially continuous low impedance connection between the input node 401 to the first voltage bus 402 a .
- the third and fourth switches 421 , 422 are switched at complementary duty cycles sufficiently less than 100% and sufficiently greater than 0%, respectively, such the action of the third and fourth switches 421 , 422 and the second inductor L 2 reduce the magnitude of the output voltage V out applied to the load 20 .
- the fifth and sixth switches 431 , 432 of the balancer circuit 430 are preferably operated at complementary duty cycles of approximately 0% and approximately 100%, respectively, which can provide an advantageous energy transfer between the capacitors C 1 , C 2 , as described above.
- the duty cycles of the fifth and sixth switches of the balancer circuit 430 can be increased and decreased, respectively, while the third and fourth switches 421 , 422 of the inverter 420 are modulated to produce a smoothed zero crossing for the output voltage V out .
- the duty cycles of the first, second, third, and fourth switches 411 , 412 , 421 , 422 are changed.
- the first and second switches 411 , 412 operate at complementary duty cycles of approximately 0% and approximately 100%, respectively, thus providing a substantially continuous low impedance connection between the input node 401 and the second voltage bus 402 b .
- the third and fourth switches 421 , 422 are operated at complementary duty cycles sufficiently greater than 0% and sufficiently less than 100%, respectively, such that the action of the switches 421 , 422 and the second inductor L 2 decrease the magnitude of the output voltage V out applied to the load 20 .
- the fifth and sixth switches 431 , 432 of the balancer circuit 430 are operated at complementary duty cycles of approximately 100% and approximately 0%, respectively.
- the duty cycles of the fifth and sixth switches of the balancer circuit 430 can be decreased and increased, respectively, while he third and fourth switches 421 , 422 of the inverter 420 are modulated to produce a smoothed zero crossing for the output voltage V out .
- FIG. 9 illustrates a power converter 600 according to another embodiment of the present invention, in which a secondary power source, e.g., a battery 460 , may be provided to generate the output voltage V out when the AC power source 10 fails.
- a secondary power source e.g., a battery 460
- either the AC power source 10 or the battery 460 is connected to the input inductor L 1 by a transfer switch S T .
- Techniques for controlling operation of such a transfer switch are known to those skilled in the art, and will not be discussed in greater detail herein.
- the rectifier circuit 410 is operated as a battery boost circuit.
- the first transistor Q 1 is first switched at a duty cycle of approximately 100% and the second transistor Q 2 is switched at a complementary duty cycle of approximately 0%, when the battery voltage V B is greater than the desired output voltage V out .
- the balancer circuit 430 can be left inactive, but preferably is operated such that the fifth transistor Q 5 is switched at a duty cycle of approximately 0% while the sixth transistor Q 6 is switched at a complementary duty cycle of approximately 100%.
- the duty cycles of the transistors Q 3 , Q 4 of the inverter 420 are varied such that the voltage V 1 at the first voltage bus 402 a is bucked to produce the desired output voltage V out .
- the duty cycles of the first and second transistors Q 1 , Q 2 of the rectifier circuit 410 are varied to provide an appropriate boost to the voltage V 1 at the first voltage bus 402 a needed to track the desired output voltage V out .
- the duty cycles of the transistors Q 5 , Q 6 of the balancer circuit 430 are changed to operate at duty cycles of approximately 0% and approximately 100%, respectively.
- the transistors Q 3 , Q 4 of the inverter circuit 420 are operated at complementary duty cycles of approximately 100% and approximately 0%, respectively.
- the rectifier transistors Q 1 , Q 2 are operated again at complementary duty cycles of approximately 100% and approximately 0%, respectively.
- the balancer circuit 430 can be left inactive, but preferably is operated such that the fifth transistor Q 5 is switched at a duty cycle of approximately 0% and the sixth transistor Q 6 is switched at a complementary duty cycle of approximately 100%.
- the transistors Q 3 , Q 4 of the inverter 420 are switched such that the voltage Y 1 at the first voltage bus 402 a is bucked to provide the desired output voltage V out .
- the transistors Q 1 , Q 2 of the rectifier 410 are operated at complementary duty cycles of approximately 100% and approximately 0%, respectively, making the voltage V 1 at the first voltage bus 402 a approximately the same as the battery voltage V B .
- the duty cycles of the transistors Q 5 , Q 6 of the balancer circuit 430 are varied such that the voltage V 2 at the second voltage bus 402 b approximately tracks the desired output voltage V out .
- the transistors Q 3 , Q 4 of the inverter 420 are operated at duty cycles of approximately 0% and approximately 100%, respectively, as the voltage V 2 at the second voltage bus 402 b is approximately the desired output voltage V out .
- the above-described operations can be used to generate a wider variety of output voltage waveforms.
- the power converter 600 of FIG. 9 can produce sinusoidal, quasi-sinusoidal, square wave, quasi-square wave and a variety of other output waveforms.
- FIG. 10 illustrates a power converter 700 according to another embodiment of the present invention, which resembles the embodiment of FIG. 5 with the addition of a battery coupling circuit 470 including seventh and eight switching transistors Q 7 , Q 8 that selectively couple a battery 460 to the first voltage bus 402 a and the neutral bus N via a switch S B and an inductor L 4 .
- the switch S B which may be a mechanical or other switching device, is not required, but can reduce energy losses if opened when the battery 460 does not require charging.
- the first DC voltage V 1 preferably is controlled such that it never falls below the battery voltage V B .
- the switching actions of the first and second switching transistors Q 1 , Q 2 of the rectifier circuit 410 and/or the fifth and sixth switching transistors Q 5 , Q 6 of the balancer circuit 430 can control the first DC voltage V 1 .
- the seventh switching transistor Q 7 can be modulated (switched) to produce a current that charges the battery 460 .
- FIG. 11 is a waveform diagram illustrating exemplary operations for the embodiment of FIG. 10 when the battery 460 is charging.
- the seventh and eighth switching transistors Q 7 , Q 8 are switched to allow current to flow from the first voltage bus 402 a to the battery 460 .
- the fifth and sixth switching transistors Q 5 , Q 6 of the balancer circuit 430 are each operated at 50% duty cycles for the time the magnitude of the input voltage V in is greater than the battery voltage V B .
- the seventh and eighth transistors Q 7 , Q 8 can be operated such that current does not flow between the battery 460 and the first voltage bus 402 a during a positive half-cycle 1210 of the input voltage V in .
- the switch S B remains open, the operations of the balancer circuit 430 during the negative half-cycle 1220 of the input voltage V in can be changed from the operations described above with reference to FIG. 10 to provide more efficient operation while maintaining the first DC voltage V 1 at or above the battery voltage V B and prevent current flow therebetween. If the switch S B is opened, further improvement in efficiency can be gained, as the first DC voltage V 1 need no longer be constrained to be greater than or equal to the battery voltage V B .
- FIG. 13 illustrates a power converter 800 according to yet another embodiment of the present invention, wherein the inductance through which the fifth and sixth switching transistors Q 5 , Q 6 couple the neutral bus N to the first and second voltage busses 402 a, 402 b is provided by a first winding L a of a transformer T 1 .
- the fifth and sixth switching transistors Q 5 , Q 6 selectively couple the first and second voltage busses 402 a, 402 b to a first end tap of the winding L a , a center tap of the winding L a being coupled to the neutral bus N.
- a second end tap of the winding L a is coupled to a node between a serial-connected pair of diodes D 1 , D 2 coupled between the first and second voltage busses 402 a, 402 b.
- a secondary power source here a battery 460
- a secondary power source here a battery 460
- a battery 460 may be coupled to a second winding L b of the transformer T 1 via a switch S B and an AC voltage generating circuit 450 (e.g. a battery converter circuit).
- the switch S B is closed and the AC voltage generating circuit 450 produces an AC voltage (e.g., a square wave, quasi-square wave, sine wave, quasi sine wave, or other periodic or quasi-periodic voltage) across the second winding L b from a DC voltage produced by the battery 460 . This induces a corresponding AC voltage across the first winding L a .
- the fifth and sixth transistors Q 5 , Q 6 in conjunction with the diodes D 1 , D 2 , produce DC voltages V 1 , V 2 on the first and second voltage busses 402 a, 402 b from the AC voltage induced across the first winding L a .
- These DC voltages V 1 , V 2 may be inverted by the inverter 420 to produce an AC voltage at the output bus 403 .
- the battery 460 may be decoupled by opening the switch S B , as might be done when the converter 800 is operated as an offline UPS.
- the AC voltage generating circuit 450 may act as a combined battery converter/battery charger circuit. In a battery charging mode, the AC voltage generating circuit 450 may act as a rectifier, producing a DC voltage across the battery 460 from an AC voltage induced on the second winding L b , thus allowing the battery 460 to be charged.
- An exemplary implementation for such dual purpose AC voltage generating circuit 450 is described with reference to FIG. 15, below.
- FIG. 14 is waveform diagram illustrating exemplary operations for the converter 800 of FIG. 13 for a case in which the input voltage V in is boosted to produce a desired output voltage V out .
- the fifth and sixth switching transistors Q 5 , Q 6 are preferably both operated at 50% duty cycles in both positive and negative half-cycles 1410 , 1420 of the input voltage V in .
- the first and second DC voltages V 1 , V 2 are constrained to have substantially equal magnitudes.
- FIG. 14 illustrate another advantageous aspect of the balancer circuit 430 .
- Operating the fifth and sixth transistors Q 5 , Q 6 at 50% duty cycles can provide energy transfer from the second capacitor C 2 to the first capacitor C 1 during the positive half-cycle 1410 of the input voltage V in , helping maintain the first DC voltage V 1 when the load draws current.
- a similar energy transfer from the first capacitor C 1 to the second capacitor C 2 can be provided during the negative half-cycle 1420 of the input voltage V in . This ability to transfer energy allows the capacitors C 1 , C 2 to have a relatively low capacitance for a given power rating in comparison to storage capacitors used in many conventional converter designs.
- each of the capacitors C 1 , C 2 allows each of the capacitors C 1 , C 2 to have a “per unit” capacitance of less than 1.
- Per unit capacitance may be described as follows. If nominal AC voltage is applied across one of the capacitors C 1 , C 2 , the capacitor would draw an AC current proportional to its capacitance. If this current is equal to a full load AC current for the power converter 400 at the nominal AC voltage, the capacitor may be described as having per unit capacitance of 1. If the current drawn at the nominal AC voltage is less than the full load AC current for the power converter 800 , however, the capacitor would have a per unit capacitance that is less than 1.
- the balancer circuit implementation of FIG. 13 utilizes a half-bridge configuration, and that similar functionality may be achieved using a full-bridge configuration.
- the diodes D 1 , D 2 of FIG. 13 may be replaced by seventh and eighth switching transistors Q 7 , Q 8 , the switching operations of which may be controlled, for example, by a switch control circuit along the lines of the switch control circuit 440 of FIG. 5.
- the embodiments of FIGS. 13 and 15 represent exemplary implementations, and that other circuit implementations fall within the scope of the present invention.
- the switching functions of the transistors Q 1 , Q 2 , Q 3 , Q 4 , Q 5 , Q 9 , Q 7 , Q 8 may be provided by a variety of switching devices, such as those described above with reference to FIG. 5 controlled using any of a number of different control circuit implementations, such as those described above with reference to FIG. 5.
- FIG. 16 illustrates an exemplary implementation of an AC voltage generating circuit 450 that may also act as a combined battery converter/battery charger circuit.
- the AC voltage generating circuit 450 includes four switching transistors Q a , Q b , Q c , Q d connected in a bridge configuration between the transformer winding L b and the battery 460 .
- the transistors Q a , Q b , Q c , Q d are controlled by a switch control circuit 452 .
- a current limiting inductor L cl is connected in series with the battery 460 and the switch S B , and a storage capacitor C S is connected across the series combination.
- the switch control circuit 452 may control switching operations of the transistors Q a , Q b , Q c , Q d responsive to a variety of state inputs, such as a sensed voltage of the battery 460 , and sensed AC input and output voltages for a battery converter in which the AC voltage generating circuit 450 is included.
- the switch control circuit 452 may be operated to selectively switch the transistors Q a , Q b , Q c , Q d such that a DC voltage produced by the battery 460 across the storage capacitor C S is inverted, producing an AC voltage across the winding L b . This may be achieved, for example, by switching a first pair of transistors Q a , Q b , in a complementary pattern to a second pair of transistors Q c , Q d at the desired AC line frequency.
- an AC voltage may be applied to the first winding L a of the transformer T 1 by action of the fifth and sixth transistors 431 , 432 of the balancer circuit 430 , inducing an AC voltage on the second winding L b .
- the switch control circuit 452 may selectively switch the transistors Q a , Q b , Q c , Q d such that the AC voltage induced on the winding L b is rectified to produce a DC voltage across the storage capacitor C S that can be used to charge the battery 460 .
- the magnitude of the voltage across the battery 460 can be controlled by the manner in which the transistors Q a , Q b , Q c , Q d are operated, such that when the battery approaches a full charge the current flowing into the battery 460 can be reduced to avoid overcharging.
- the switch S B can be opened.
- FIG. 16 represents an exemplary implementation, and that other circuit implementations fall within the scope of the present invention.
- the switching functions of the transistors Q a , Q b , Q c , Q d may be provided by any of a number of different switching devices such as those described in relation to FIG. 5, controlled using any of a number of different implementations of the switch control circuit 452 , such as ones similar to those described for the switch control circuit 440 of FIG. 5.
- functions of the switch control circuits 440 , 452 of FIGS. 5 and 16 may also be combined in one or more devices. It will further be understood that although the battery converter/battery charger circuit illustrated in 450 of FIG.
- circuit 16 is capable of a combined rectifier/inverter operation in order provide battery charging in addition to generation of an AC voltage across the winding L b
- circuit implementations of the AC voltage generating circuit 450 which provide only inversion of the DC voltage produced by the battery may also be used with the present invention.
- Such a circuit may provide battery charging via an ancillary battery charging circuit (not shown).
Abstract
Description
- The present invention relates to electrical power devices and methods of operation thereof, and more particularly, to power conversion devices and methods of operation thereof.
- Uninterruptible power supplies (UPSs) are power conversion devices that are commonly used to provide conditioned, reliable power for computer networks, telecommunications networks, medical equipment and the like. UPSs are widely used with computers and similar computing devices, including but not limited to personal computers, workstations, mini computers, network servers, disk arrays and mainframe computers, to insure that valuable data is not lost and that the device can continue to operate notwithstanding temporary loss of an AC utility source. UPSs typically provide power to such electronic equipment from a secondary source, such as a battery, in the event that a primary alternating current (AC) utility source drops out (blackout) or fails to provide a proper voltage (brownout).
- Conventional UPSs may be classified into categories. Referring to FIG. 1, a typical off-line UPS disconnects a load from a
primary AC source 10 when the primary AC source fails or is operating in a degraded manner, allowing the load to be served from a secondary source such as a battery. TheAC power source 10 is connected in series with a switch S1, producing an AC voltage across aload 20 when the switch S1 is closed, Energy storage is typically provided in the form of a storage capacitor CS. The secondary power source, here a battery B, is connected to theload 20 via alow voltage converter 30 and a transformer T. When theAC power source 10 fails, the switch S1 is opened, causing the load to draw power from the battery B. Thelow voltage converter 30 typically is an inverter that produces a quasi-square wave or sine wave voltage on a first winding L1 of the transformer T from a DC voltage produced by the battery B. The first winding L1 is coupled to a second winding L2 of the transformer T connected across theload 20. When the AC power source is operational, i.e., when the switch S1 is closed, the battery B may be charged using the low-voltage converter 30 or a separate battery charger circuit (not shown). - A line interactive (LIA) UPS topology is illustrated in FIG. 2. Here, the transformer T has a third winding L3 that may be connected in series with the
load 20 using switches S2, S3 to “buck” or “boost” the voltage applied to theload 20. As with the offline UPS topology of FIG. 1, when theAC power source 10 fails, the switch S1 can be opened to allow theload 20 to run off the battery B. - As illustrated in FIG. 3, a typical on-line UPS includes a
rectifier 40 that receives an AC voltage from anAC power source 10, producing a DC voltage across a storage capacitor CS at anintermediate node 45. Aninverter 50 is connected between theintermediate node 45, and is operative to produce an AC voltage across aload 20 from the DC voltage. As shown, a battery B is connected to theintermediate node 45 via a DC/DC converter 60, supplying auxiliary power. Alternatively, the DC/DC converter can be eliminated and a high-voltage battery (not shown) connected directly to theintermediate node 45. - Each of these topologies may have disadvantages. For example, typical conventional on-line and LIA UPSs for 60 Hz applications use 60 Hz magnetic components (e.g., transformers and inductors) that are sized for such frequencies, and thus may be large, heavy and expensive. LIA UPSs often exhibit step voltage changes that can affect the performance of the load. Conventional off-line, LIA and on-line UPSs often use large storage capacitors, which tend to be bulky and expensive, in order to maintain an acceptable output voltage under heavy loading conditions. Moreover, because conventional UPSs are typically designed to operate in only one of the above-described off-line, LIA or on-line modes, sellers of UPSs may be required to maintain large inventories including several different types of UPSs in order to meet a variety of different customer applications.
- In light of the foregoing, it is an object of the present invention to provide improved power converters and methods of operating power converters for use in devices such as uninterruptible power supplies (UPSs).
- It is another object of the present invention to provide power converters that can be operated in a number of different modes.
- It is yet another object of the present invention to provide power converters that can utilize smaller magnetic components and storage capacitors.
- These and other objects, features and advantages may be provided according to the present invention by power converters and methods of operation thereof in which a rectifier circuit produces first and second voltages (e.g., ±DC voltages) on first and second voltage busses from an AC input voltage produced by an AC power source, an inverter circuit produces an AC output voltage from the first and second voltages, and a balancer circuit controls the relative magnitudes of the first and second voltages responsive to respective first and second rates at which the balancer circuit couples the first and second voltage busses to a neutral bus of the AC power source. Preferably, the rectifier circuit includes first and second switches that selectively couple the first and second voltage busses to a phase bus of the AC power source through a first inductance, the inverter circuit includes third and fourth switches that selectively couple the first and second voltage busses to a load through a second inductance, and the balancer circuit includes fifth and sixth switches that selectively couple the first and second voltage bussed to the neutral bus through a third inductance, such as an inductor or transformer winding.
- The use of circuit topologies as described herein can provide several advantages. The balancer circuit enables energy transfer between first and second storage capacitors connected between the neutral bus and the first and second voltage busses, respectively, thus allowing the storage capacitors to be smaller than the storage capacitors typically used in conventional power converters with comparable power ratings. The switches in the rectifier, inductor and balancer can be controlled such that the power converter can be operated in a number of different power transfer modes. A secondary power source, such as a battery, may also be coupled to the power converter via a winding of a transformer that also serves as an inductance for the balancer circuit. In one embodiment, this coupling may be achieved through a combination battery converter/battery charger circuit that can also charge the battery when the converter is running off an AC power source. According to another aspect of the present invention, switches in the balancer circuit can be operated at varying duty cycles in positive and negative half-cycles of the AC input voltage, which can allow the power converter to be operated in a more efficient manner.
- In particular, according to one embodiment of the present invention, a power converter includes first and second voltage busses and a neutral bus. A first switching circuit, e.g., a rectifier circuit, is operative to selectively couple an input node thereof to the first and second voltage busses. A balancer circuit is operative to selectively couple the neutral bus to the first and second voltage busses such that relative magnitudes of respective first and second voltages on the first and second voltage busses are controlled responsive to respective first and second rates at which the balancer circuit couples the first and second voltage busses to the neutral bus. A second switching circuit, e.g., an inverter circuit, is operative to selectively couple the first and second voltage busses to a load at an output node thereof.
- The balancer circuit preferably includes first and second switches operative to selectively couple respective ones of the first and second voltage busses to the neutral bus through an inductance, such that the relative magnitudes of the first and second voltages are controlled responsive to respective first and second duty cycles of the first and second switches. The balancer circuit preferably is responsive to an AC input voltage applied to the input node to vary the respective duty cycles at which the first and second switches operate.
- In one embodiment of the present invention, a battery converter circuit may be switchably coupled to at least one of the first and second voltage busses. In one exemplary circuit implementation, the battery converter circuit may include an inductor configured to be connected in series with a battery. A first switch is operative to selectively couple one end of a series combination of a battery and the inductor to one of the first or second voltage busses. A second switch is operative to selectively couple the one end of the series combination of a battery and an inductor to another end of the series combination of a battery and the inductor.
- In another embodiment of the present invention, the balancer circuit includes a transformer including a first winding having a first tap coupled to the first and second switches and a second tap coupled to the neutral bus. The first and second switches selectively couple the first tap of the transformer to the first and second voltage busses. The power converter may further include first and second diodes, the first diode having a cathode coupled to the first voltage bus, the second diode having a cathode coupled to an anode of the first diode and an anode coupled to the second voltage bus. The first winding of the transformer may have a first end tap coupled to the first and second switches, a second end tap coupled to the anode of the first diode and the cathode of the second diode, and a center tap coupled to the neutral bus. Alternatively, the converter may further include third and fourth switches. The first winding of the transformer may have a first end tap coupled to the first and second switches, a second end tap coupled to the third and fourth switches, and a center tap coupled to the neutral bus, wherein the third switch is operative to couple and decouple the second end tap of the first winding and the first voltage bus and the fourth switch is operative to couple and decouple the second end tap of the first winding and the second voltage bus.
- In another embodiment according to the present invention, a second winding of the transformer is inductively coupled to the first winding. An AC voltage generating circuit is coupled to the second winding of the transformer and operative to apply an AC voltage thereto. The AC voltage generating circuit may include a battery converter circuit operative to generate an AC voltage on the second winding of the transformer from a DC voltage produced by a battery coupled to the battery converter circuit. The AC voltage generating circuit may include a combined battery converter/battery charger circuit that is operative to generate an AC voltage on the second winding of the transformer from a DC voltage produced by the battery, and to produce a DC voltage across the battery from an AC voltage induced on the second winding of the transformer.
- According to another aspect of the present invention, a power converter includes a rectifier circuit configured to connect to an AC power source and operative to produce first and second DC voltages at first and second voltage busses, respectively, by selectively coupling the first and second voltage busses to the AC power source through a first inductance. First and second capacitors couple the first and second voltage busses, respectively, to a neutral bus. An inverter circuit is configured to connect to a load and operative to selectively couple the first and second voltage busses to the load through a second inductance. A balancer circuit is operative to selectively couple the first and second voltage busses to the neutral bus through a third inductance such that relative magnitudes of the first and second DC voltages are controlled responsive to respective first and second rates at which the balancer circuit couples the first and second voltage busses to the neutral bus.
- In one embodiment of the present invention, the power converter includes a switch control circuit. The rectifier circuit includes a first inductor having a first terminal configured to receive an AC input voltage, a first switch responsive to the switch control circuit to couple and decouple a second terminal of the first inductor and the first voltage bus, and a second switch responsive to the switch control circuit to couple and decouple the second terminal of the first inductor and the second voltage bus. The inverter circuit includes a second inductor having a first terminal configured to connect to a load, a third switch responsive to the switch control circuit to couple and decouple a second terminal of the second inductor and the first voltage bus, and a fourth switch responsive to the switch control circuit to couple and decouple the second terminal of the second inductor and the second voltage bus. The balancer circuit may include a third inductor having a first terminal coupled to the neutral bus, a fifth switch responsive to the switch control circuit to couple and decouple a second terminal of the third inductor and the first voltage bus, and a sixth switch responsive to the switch control circuit to couple and decouple the second terminal of the third inductor and the second voltage bus.
- According to other aspects of the present invention, power transfer between a load and an AC power source that produces an AC input voltage between a phase bus and a neutral bus is controlled. First and second voltage busses are selectively coupled to the phase bus through a first inductance to produce first and second DC voltages at first and second voltage busses, respectively. The first and second voltage busses are selectively coupled to the load through a second inductance. The first and second voltage busses are selectively coupled to the neutral bus through a third inductance such that relative magnitudes of the first and second DC voltages are controlled responsive to respective first and second rates at which the first and second voltage busses are coupled to the neutral bus.
- According to yet another embodiment of the present invention, the first and second voltage busses are selectively coupled to phase bus by switching a first switch to couple and decouple the phase bus and the first voltage bus through a first inductor and by switching a second switch to couple and decouple the phase bus and the second voltage bus through the first inductor. The first and second voltage busses are selectively coupled to the load by switching a third switch to couple and decouple the first voltage bus and the load through a second inductor and by switching a fourth switch to couple and decouple the second voltage bus and the load bus through the second inductor. The switching is preferably performed responsive to the AC input voltage. In particular, the respective duty cycles at which the first, second, third and fourth switches are operated are varied responsive to the AC input voltage.
- In another embodiment of the present invention, a substantially continuous low impedance connection between the phase bus and the load is provided through selected combinations of the first, second, third and fourth switches when the AC input voltage is approximately at a nominal level. When the AC input voltage is less than the nominal level, the first and second switches are switched to boost the magnitudes of the first and second DC voltages while providing respective substantially continuous low impedance connections between the load and respective ones of the first and second voltages busses through respective ones of the third and fourth switches during respective positive and negative half-cycles of the AC input voltage. When the AC input voltage is greater than the nominal level, respective substantially continuous low-impedance connections between the AC power source and respective ones the first and second voltage busses are provided through respective ones of the first and second switches during respective positive and negative half-cycles of the AC input voltage, while bucking a voltage generated at the load from the first and second DC voltages.
- According to another embodiment of the present invention, fifth and sixth switches that couple and decouple respective ones of the first and second voltage busses and the neutral bus through a third inductor are switched such that the magnitude of the first DC voltage is substantially greater than the magnitude of the second DC voltage during a positive half-cycle of the AC input voltage and such that the magnitude of the second DC voltage is substantially greater than the magnitude of the first DC voltage during a negative half-cycle of the AC input voltage. A battery may be selectively coupled to at least one of the first inductance, the first voltage bus, or the second voltage bus to enable power transfer between the battery and the first and second voltage busses.
- FIGS.1-3 are schematic diagrams of power conversion circuit topologies used in typical conventional uninterruptible power supplies (UPSs).
- FIGS.4-5 are schematic diagrams illustrating power converters according to embodiments of the present invention.
- FIGS.6-8 are waveform diagrams illustrating exemplary operations of a power converter according to the embodiment of FIG. 4.
- FIGS.9-10 are schematic diagrams illustrating power converters according to other embodiments of the present invention.
- FIGS.11-12 are waveform diagrams illustrating exemplary operations for a power converter of FIG. 10.
- FIG. 13 is a schematic diagram illustrating a power converter according to another embodiment of the present invention.
- FIG. 14 is a waveform diagram illustrating exemplary operations for a power converter of FIG. 12.
- FIG. 15 is a schematic diagram illustrating a power converter according to another embodiment of the present invention.
- FIG. 16 is a schematic diagram illustrating an AC voltage generating circuit according to an embodiment of the present invention.
- The present invention now will be described more fully hereinafter with reference to the accompanying drawings, in which preferred embodiments of the invention are shown. This invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein; rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fullly convey the scope of the invention to those skilled in the art. In the drawings, like numbers refer to like elements.
- FIG. 4 is a schematic diagram illustrating a
power converter 400 according to an embodiment of the present invention. Thepower converter 400 includes arectifier circuit 410 that is coupled to anAC power source 10, producing first and second DC voltages V1, V2 on respective first and second voltage busses 402 a, 402 b that are coupled to a neutral bus N by first and second capacitors C1, C2. Therectifier circuit 410 includes first andsecond switches phase bus 401 of theAC power source 10 to the first and second power busses 402 a, 402 b through a first inductor L1, responsive to aswitch control circuit 440. Thepower converter 400 includes aninverter circuit 420 that produces an AC output voltage Vout across a load 20 (here shown as including capacitance CL and generalized impedance ZL) at anoutput 403 from the first and second DC voltages at the first and second voltage busses 402 a, 402 b. The inverter circuit includes third andfourth switches load 20 through a second inductor L2, responsive to theswitch control circuit 440. - The
power converter 400 also includes abalancer circuit 430 that is operative to control the relative magnitudes of the first and second DC voltages V1, V2 on the first and second voltage busses 402 a, 402 b by controlling respective first and second rates (e.g., duty cycles) at which the neutral bus N is connected to respective ones of the first and second voltage busses 402 a, 402 b through a third inductor L3. As used herein, a “balancer” circuit is a circuit that is capable of effecting a desired “balance” between voltages on different busses, such as the first and second voltage busses 402 a, 402 b of FIG. 4. As is discussed in greater detail below, this capability can allow power converters according to embodiments of the present invention, among other things, to operate more efficiently and/or to utilize smaller components, e.g., storage capacitors, than those used in many conventional devices. For the illustrated embodiment, this control is achieved by switching fifth andsixth switches switch control circuit 440. Theswitches switches - The
inverter circuit 420 can allow current flow from the first and second voltage busses 402 a, 402 b to theload 20, or vice versa, thus making the inverter circuit 420 a four-quadrant converter. Therectifier circuit 410 preferably has similar characteristics, except that different pulse-width modulation (PWM) patterns preferably are applied to theswitches rectifier circuit 410 than those applied to theswitches inverter circuit 420. The PWM patterns employed for theinverter circuit 420 preferably produce a voltage controlled, current limited output voltage, while the PWM patterns employed for therectifier circuit 410 preferably provide a controlled current to and from theAC power source 10. Therectifier circuit 410 can be operated such that a current is produced that causes power flow into theconverter 400 from theAC power source 10, or such that a current is produced that causes power flow into theAC power source 10 from thepower converter 400. - FIG. 5 illustrates a
power converter 500 according to another embodiment of the present invention. Portions of theconverter 500 of FIG. 5 that are the same as those illustrated in FIG. 4 are denoted with like reference numerals, and further detailed discussion of their operations will not be provided in light of the preceding discussion of FIG. 4. In FIG. 5, diode protected transistor switches Q1, Q2, Q3, Q4, Q5, Q6 are used for theswitches switch control circuit 440 that includes a microcontroller 442 and adriver circuit 444. Theswitch control circuit 440 may control the transistors Q1, Q2, Q3, Q4, Q5, Q6 responsive to sensed AC input voltage Vin,sensed and/or sensed AC output voltage Vout,sensed. - The
switch control circuit 440 may operate using different types of control techniques. For example, theswitch control circuit 440 may utilize an “open loop” control technique wherein the operations of the transistors Q1, Q2, Q3, Q4, Q5, Q6 are controlled based on a sensed AC input voltage Vin,sensed, without reference to the output voltage Vout produced by theconverter 500. Theswitch control circuit 500 may also operate on a “closed loop” basis, using a sensed AC output voltage Vout,sensed to guide control of the operations of the transistors Q1, Q2, Q3, Q4, Q5, Q6. The sensed AC input voltage Vin,sensed and sensed AC output voltage Vout,sensed may be provided to theswitch control circuit 440 in a number of different forms including, but not limited to, analog or digital representations of the input and output voltages Vin, Vout, or quantities related to and/or derived from the input and output voltages Vin, Vout. If battery-powered or battery-boosted operation is provided, as described below, theswitch control circuit 440 may also operate based on a sensed battery voltage (not shown), which may include an analog or digital representation of an actual battery voltage, or a quantity related to and/or derived from such a battery voltage. - It will be appreciated that the embodiment of FIG. 5 represents an exemplary implementation, and that other circuit implementations fall within the scope of the present invention. For example, the switching functions of the transistors Q1, Q2, Q3, Q4, Q5, Q6 may be provided by a variety of switching devices including, but not limited to, bipolar transistors, field-effect transistors (FETs), metal oxide semiconductor FETs (MOSFETs), gate turn-on devices (GTOs), and the like. The
driver circuit 444 may include variety of different components as well, and preferably includes components suitable for controlling the particular type(s) of switching devices used. - Other functions of the switching
control circuit 440 may be implemented in a number of different ways within the scope of the present invention. For example, functions of the microcontroller 442 may be implemented using discrete logic circuits or programmable logic circuits such as programmable logic devices (PLDs) instead or in conjunction with a microcontroller, microprocessor or similar device. Functions of the microcontroller 442 and thedriver circuit 444 may also be combined in one or more devices, such as an application-specific integrated circuit (ASIC) or a hybrid microcircuit. - FIGS.6-8, 11-12 and 14 are waveform diagrams illustrating exemplary operations of power converters according to embodiments of the present invention. For purposes of the discussion of FIGS. 6-8, 11-12, and 14, operation of the power converters of FIGS. 4, 10 and 13 will be described in terms of the control of the functions of
switches balancer circuits power converter 400 of FIG. 4 (and corresponding devices in the embodiments of FIGS. 5, 9-10, 13 and 15), and more particularly, in terms of the control of “duty cycles” at which the switches are switched (modulated). As used herein, “duty cycle” generally refers to a percentage of time during a switching cycle period (corresponding to the switching frequency described above) that a switch is in a “closed,” i.e., conductive, state. Thus, for example, a switch that is maintained at a 50% duty cycle is intermittently “on” for half of a switching cycle, while a switch approaching a 100% duty cycle, e.g., a 99% duty cycle, is on for nearly all of the switching cycle. It will be understood that, as described herein, the switches discussed may also operate at a 100% duty cycle, i.e., be maintained in an “on” state throughout one or more switching cycles, or at a 0% duty cycle, i.e., be maintained in an “off” state throughout one or more switching cycles. - It will be appreciated that this duty cycle control may be achieved, for example, by application of appropriate control signals to appropriate switching components. For example, in the embodiment of FIG. 5, duty cycle control of the switching transistors Q1, Q2, Q3, Q4, Q5, Q6 may be achieved by the microcontroller 442 and
driver circuit 444 applying appropriate base drive signals to the switching transistors Q1, Q2, Q3, Q4, Q5, Q6. It will be understood, however, that such duty cycle control may be achieved using any of a number of other switching devices and control signal generating circuits. - Preferably, each of the first, second, third, fourth, fifth and
sixth switches balancer circuits converter 400, more preferably, at one or more switching frequencies that each are at least 10 times higher than the frequency of the AC input voltage Vin. Using relatively high switching frequencies allows the magnetic components (e.g., the inductors L1, L2, L3) to be relatively small in size. It will be appreciated that although different switching frequencies can be used among theswitches load 20, as illustrated in FIG. 4, or may be incorporated in theconverter 400. - Preferably, the three switch pairs comprising the first and
second switches fourth switches sixth switches second switch 412 of FIG. 4 is preferably generally constrained to be “off” when thefirst switch 411 is “on,” and vice versa. However, it will be appreciated that, generally, practical circuit implementations within the scope of the present invention may be used in which the “complementary” switches are operated in an approximately or substantially complementary fashion. For example, a switch pair may be operated in a “break before make” fashion, such that one of the switches in the pair is turned off slightly before the other switch in the pair is turned on. In other circuit implementations, a slight amount of overlap of “on” periods of switches of a complementary pair may be allowed, e.g., a “make before break” mode of operation. Techniques for providing such “make before break” and “break before make” operations are known to those of skill in the art, and will not be discussed in greater detail herein. - FIG. 6 illustrates exemplary operations of the
converter 400 of FIG. 4 when the input voltage Vin is at or near a desired level for the output voltage Vout. During a positive half-cycle 610 of the input voltage Vin, thefirst switch 411 is operated at a duty cycle of approximately 100% (i.e., approaching a steady state “closed” state), while thesecond switch 412 is operated a complementary duty cycle of approximately 0% (i.e., approaching a steady state “open” state). Thethird switch 421 and thefourth switch 422 also operate at complementary duty cycles of approximately 100% and approximately 0%, respectively. Thus, a substantially continuous low impedance connection is provided between theinput node 401 and theload 20 at theoutput node 403 via thefirst voltage bus 402 a. As a result, the first DC voltage V1 and the output voltage Vout essentially track the input voltage Vin. Although thebalancer circuit 430 can be left in an inactive state, i.e., both the fifth and sixth switches can be left in an “off” state, thefifth switch 431 preferably is operated at a duty cycle of approximately 0% until the first DC voltage V1 falls below a first threshold voltage VT1 while thesixth switch 432 is operated at a complementary duty cycle of approximately 100%. - During the negative half-
cycle 620 of the input voltage Vin, the duty cycles of the first, second, third andfourth switches second switches fourth switches input node 401 and theoutput node 403 via thesecond voltage bus 402 b, such that the second DC voltage V2 and the output voltage Vout essentially track the input voltage Vin. As with the positive half-cycle, although thebalancer circuit 430 can be left inactive, thefifth switch 431 preferably is operated at a duty cycle of approximately 100% while the second DC voltage V2 is less than second threshold voltage VT2, with thesixth switch 432 being operating at a complementary duty cycle of approximately 0%. - The
switches balancer circuit 430 can be used to smooth transition of the output voltage Vout near zero volts, by varying the duty cycles of these switches between the threshold voltages VT1, VT2. As the first DC voltage V1 falls below the first threshold voltage VT1 during the positive half-cycle 620, thebalancer circuit 430 begins to increase the duty cycle of thefifth switch 431 while decreasing the duty cycle of thesixth switch 432 in a complementary fashion, thus driving the second DC voltage V2 negative before the actual zero crossing of the input voltage Vin. The duty cycles of the fifth andsixth switches inverter circuit 420 to be switched such that a relatively smooth transition of the output voltage Vout through zero volts can be achieved. - Similar zero-crossing control can be achieved as the AC input voltage Vin approaches zero volts during the
negative half cycle 620 by varying the duty cycles of the fifth andsixth switches fifth switch 431 is decreased while the duty cycle of thesixth switch 432 is increased, driving the first DC voltage V1 in a positive direction before the zero crossing of the input voltage Vin. The duty cycles of the fifth andsixth switches sixth switches - FIG. 7 illustrates exemplary operations of the
converter 400 of FIG. 4 when the input voltage Vin falls below a desired level for the output voltage Vout. During a positive half-cycle 710 of the input voltage Vin, the first andsecond switches switches third switch 421 and thefourth switch 422 operate at complementary duty cycles of approximately 100% and approximately 0%, respectively, providing a substantially continuous low-impedance connection between thefirst voltage bus 402 a and theoutput node 403. While the first DC voltage V1 is above a first threshold voltage VT1, the fifth andsixth switches balancer circuit 430 are operated at complementary duty cycles of approximately 0% and approximately 100%, respectively. However, when the first DC voltage V1 falls below the first threshold voltage VT1, the duty cycles of the fifth andsixth switches balancer circuit 430 can be increased and decreased, respectively, while the third andfourth switches inverter 420 are modulated to produce a smoothed zero crossing for the output voltage Vout. - During the negative half-
cycle 720 of the input voltage Vin, the duty cycles of the first, second, third andfourth switches second switches second switches fourth switches second voltage bus 402 b and theoutput node 403. While the second DC voltage V2 is less than the second threshold voltage VT2, the fifth andsixth switches balancer circuit 430 are operated at complementary duty cycles of approximately 100% and approximately 0%, respectively, allowing the output voltage Vout to essentially track the second DC voltage V2. However, when the second DC voltage V2 exceeds the second threshold voltage VT2, the duty cycles of the fifth andsixth switches balancer circuit 430 may be decreased and increased, respectively, while the third andfourth switches inverter 420 are modulated to produce a smoothed zero crossing for the output voltage Vout. - FIG. 8 illustrates exemplary operations of the
converter 400 of FIG. 4 when the input voltage Vin exceeds a desired level for the output voltage Vout. During a positive half-cycle 810 of the input voltage Vin, thefirst switch 411 is operated at a duty cycle of approximately 100%, while thesecond switch 412 is operated at a complementary duty cycle of approximately 0%, thus providing a substantially continuous low impedance connection between theinput node 401 to thefirst voltage bus 402 a. The third andfourth switches fourth switches load 20. While the first DC voltage V1 is above a first threshold voltage VT1, the fifth andsixth switches balancer circuit 430 are preferably operated at complementary duty cycles of approximately 0% and approximately 100%, respectively, which can provide an advantageous energy transfer between the capacitors C1, C2, as described above. However, when the first DC voltage V1 falls below the first threshold voltage VT1, the duty cycles of the fifth and sixth switches of thebalancer circuit 430 can be increased and decreased, respectively, while the third andfourth switches inverter 420 are modulated to produce a smoothed zero crossing for the output voltage Vout. - During the negative half-
cycle 820 of the input voltage Vin, the duty cycles of the first, second, third, andfourth switches second switches input node 401 and thesecond voltage bus 402 b. The third andfourth switches switches load 20. While the second DC voltage V2 is below the second threshold voltage VT1, the fifth andsixth switches balancer circuit 430 are operated at complementary duty cycles of approximately 100% and approximately 0%, respectively. However, when the second DC voltage V2 exceeds the second threshold voltage VT2, the duty cycles of the fifth and sixth switches of thebalancer circuit 430 can be decreased and increased, respectively, while he third andfourth switches inverter 420 are modulated to produce a smoothed zero crossing for the output voltage Vout. - FIG. 9 illustrates a
power converter 600 according to another embodiment of the present invention, in which a secondary power source, e.g., abattery 460, may be provided to generate the output voltage Vout when theAC power source 10 fails. As shown, either theAC power source 10 or thebattery 460 is connected to the input inductor L1 by a transfer switch ST. Techniques for controlling operation of such a transfer switch are known to those skilled in the art, and will not be discussed in greater detail herein. - When the transfer switch ST is in a state such that the
battery 460 is connected to the inductor L1, i.e., when theconverter 600 is operating off of DC power provided by thebattery 460, therectifier circuit 410 is operated as a battery boost circuit. To generate a positive half-cycle of a desired sinusoidal output voltage Vout, the first transistor Q1 is first switched at a duty cycle of approximately 100% and the second transistor Q2 is switched at a complementary duty cycle of approximately 0%, when the battery voltage VB is greater than the desired output voltage Vout. Thebalancer circuit 430 can be left inactive, but preferably is operated such that the fifth transistor Q5 is switched at a duty cycle of approximately 0% while the sixth transistor Q6 is switched at a complementary duty cycle of approximately 100%. The duty cycles of the transistors Q3, Q4 of theinverter 420 are varied such that the voltage V1 at thefirst voltage bus 402 a is bucked to produce the desired output voltage Vout. - Once the desired output voltage Vout is greater than the battery voltage VB, however, the duty cycles of the first and second transistors Q1, Q2 of the
rectifier circuit 410 are varied to provide an appropriate boost to the voltage V1 at thefirst voltage bus 402 a needed to track the desired output voltage Vout. The duty cycles of the transistors Q5, Q6 of thebalancer circuit 430 are changed to operate at duty cycles of approximately 0% and approximately 100%, respectively. The transistors Q3, Q4 of theinverter circuit 420 are operated at complementary duty cycles of approximately 100% and approximately 0%, respectively. - When the desired output voltage Vout is again less than the battery voltage VB (towards the end of the positive half-cycle), the rectifier transistors Q1, Q2 are operated again at complementary duty cycles of approximately 100% and approximately 0%, respectively. The
balancer circuit 430 can be left inactive, but preferably is operated such that the fifth transistor Q5 is switched at a duty cycle of approximately 0% and the sixth transistor Q6 is switched at a complementary duty cycle of approximately 100%. The transistors Q3, Q4 of theinverter 420 are switched such that the voltage Y1 at thefirst voltage bus 402 a is bucked to provide the desired output voltage Vout. - To generate the negative half-cycle of the desired output voltage Vout, the transistors Q1, Q2 of the
rectifier 410 are operated at complementary duty cycles of approximately 100% and approximately 0%, respectively, making the voltage V1 at thefirst voltage bus 402 a approximately the same as the battery voltage VB. The duty cycles of the transistors Q5, Q6 of thebalancer circuit 430 are varied such that the voltage V2 at thesecond voltage bus 402 b approximately tracks the desired output voltage Vout. The transistors Q3, Q4 of theinverter 420 are operated at duty cycles of approximately 0% and approximately 100%, respectively, as the voltage V2 at thesecond voltage bus 402 b is approximately the desired output voltage Vout. - It will be appreciated that the above-described operations can be used to generate a wider variety of output voltage waveforms. For example, by appropriately modulating the switching elements of the rectifier and
balancer circuits power converter 600 of FIG. 9 can produce sinusoidal, quasi-sinusoidal, square wave, quasi-square wave and a variety of other output waveforms. - FIG. 10 illustrates a
power converter 700 according to another embodiment of the present invention, which resembles the embodiment of FIG. 5 with the addition of abattery coupling circuit 470 including seventh and eight switching transistors Q7, Q8 that selectively couple abattery 460 to thefirst voltage bus 402 a and the neutral bus N via a switch SB and an inductor L4. The switch SB, which may be a mechanical or other switching device, is not required, but can reduce energy losses if opened when thebattery 460 does not require charging. - When the switch SB is closed, the first DC voltage V1 preferably is controlled such that it never falls below the battery voltage VB. The switching actions of the first and second switching transistors Q1, Q2 of the
rectifier circuit 410 and/or the fifth and sixth switching transistors Q5, Q6 of thebalancer circuit 430 can control the first DC voltage V1. When the first DC voltage V1 exceeds the battery voltage VB, the seventh switching transistor Q7 can be modulated (switched) to produce a current that charges thebattery 460. - FIG. 11 is a waveform diagram illustrating exemplary operations for the embodiment of FIG. 10 when the
battery 460 is charging. For a portion of a positive half-cycle 1110 of the input voltage Vin when the first DC voltage V1 is greater than the battery voltage VB, the seventh and eighth switching transistors Q7, Q8 are switched to allow current to flow from thefirst voltage bus 402 a to thebattery 460. During a negative half-cycle 1120, the fifth and sixth switching transistors Q5, Q6 of thebalancer circuit 430 are each operated at 50% duty cycles for the time the magnitude of the input voltage Vin is greater than the battery voltage VB. This keeps the first DC voltage V1 significantly greater than the battery voltage VB during this period, allowing the seventh and eighth transistors Q7, Q8 to be switched to provide current flow into thebattery 460 from thefirst voltage bus 402 a. In this manner, the current drawn from theAC source 10 for the positive and negative half-cycles - Referring to FIG. 12, when the
battery 460 does not need charging, the seventh and eighth transistors Q7, Q8 can be operated such that current does not flow between thebattery 460 and thefirst voltage bus 402 a during a positive half-cycle 1210 of the input voltage Vin. If the switch SB remains open, the operations of thebalancer circuit 430 during the negative half-cycle 1220 of the input voltage Vin can be changed from the operations described above with reference to FIG. 10 to provide more efficient operation while maintaining the first DC voltage V1 at or above the battery voltage VB and prevent current flow therebetween. If the switch SB is opened, further improvement in efficiency can be gained, as the first DC voltage V1 need no longer be constrained to be greater than or equal to the battery voltage VB. - FIG. 13 illustrates a
power converter 800 according to yet another embodiment of the present invention, wherein the inductance through which the fifth and sixth switching transistors Q5, Q6 couple the neutral bus N to the first and second voltage busses 402 a, 402 b is provided by a first winding La of a transformer T1. Specifically, the fifth and sixth switching transistors Q5, Q6 selectively couple the first and second voltage busses 402 a, 402 b to a first end tap of the winding La, a center tap of the winding La being coupled to the neutral bus N. A second end tap of the winding La is coupled to a node between a serial-connected pair of diodes D1, D2 coupled between the first and second voltage busses 402 a, 402 b. - According to another aspect of the present invention also illustrated by the embodiment of FIG. 13, a secondary power source, here a
battery 460, may be coupled to a second winding Lb of the transformer T1 via a switch SB and an AC voltage generating circuit 450 (e.g. a battery converter circuit). In a battery-powered or battery-boosted mode, the switch SB is closed and the ACvoltage generating circuit 450 produces an AC voltage (e.g., a square wave, quasi-square wave, sine wave, quasi sine wave, or other periodic or quasi-periodic voltage) across the second winding Lb from a DC voltage produced by thebattery 460. This induces a corresponding AC voltage across the first winding La. Under appropriate control of a switch control circuit (such as theswitch control circuit 440 of FIG. 5), the fifth and sixth transistors Q5, Q6, in conjunction with the diodes D1, D2, produce DC voltages V1, V2 on the first and second voltage busses 402 a, 402 b from the AC voltage induced across the first winding La. These DC voltages V1, V2 may be inverted by theinverter 420 to produce an AC voltage at theoutput bus 403. Thebattery 460 may be decoupled by opening the switch SB, as might be done when theconverter 800 is operated as an offline UPS. - In embodiments according to the present invention, the AC
voltage generating circuit 450 may act as a combined battery converter/battery charger circuit. In a battery charging mode, the ACvoltage generating circuit 450 may act as a rectifier, producing a DC voltage across thebattery 460 from an AC voltage induced on the second winding Lb, thus allowing thebattery 460 to be charged. An exemplary implementation for such dual purpose ACvoltage generating circuit 450 is described with reference to FIG. 15, below. - FIG. 14 is waveform diagram illustrating exemplary operations for the
converter 800 of FIG. 13 for a case in which the input voltage Vin is boosted to produce a desired output voltage Vout. Here, the fifth and sixth switching transistors Q5, Q6 are preferably both operated at 50% duty cycles in both positive and negative half-cycles - The operations of FIG. 14 illustrate another advantageous aspect of the
balancer circuit 430. Operating the fifth and sixth transistors Q5, Q6 at 50% duty cycles can provide energy transfer from the second capacitor C2 to the first capacitor C1 during the positive half-cycle 1410 of the input voltage Vin, helping maintain the first DC voltage V1 when the load draws current. A similar energy transfer from the first capacitor C1 to the second capacitor C2 can be provided during the negative half-cycle 1420 of the input voltage Vin. This ability to transfer energy allows the capacitors C1, C2 to have a relatively low capacitance for a given power rating in comparison to storage capacitors used in many conventional converter designs. - In particular, the energy transfer described above allows each of the capacitors C1, C2 to have a “per unit” capacitance of less than 1. Per unit capacitance may be described as follows. If nominal AC voltage is applied across one of the capacitors C1, C2, the capacitor would draw an AC current proportional to its capacitance. If this current is equal to a full load AC current for the
power converter 400 at the nominal AC voltage, the capacitor may be described as having per unit capacitance of 1. If the current drawn at the nominal AC voltage is less than the full load AC current for thepower converter 800, however, the capacitor would have a per unit capacitance that is less than 1. - It will be appreciated that the balancer circuit implementation of FIG. 13 utilizes a half-bridge configuration, and that similar functionality may be achieved using a full-bridge configuration. Referring to FIG. 15, in a
power converter 900 according to another embodiment of the present invention, the diodes D1, D2 of FIG. 13 may be replaced by seventh and eighth switching transistors Q7, Q8, the switching operations of which may be controlled, for example, by a switch control circuit along the lines of theswitch control circuit 440 of FIG. 5. It will be appreciated that the embodiments of FIGS. 13 and 15 represent exemplary implementations, and that other circuit implementations fall within the scope of the present invention. For example, the switching functions of the transistors Q1, Q2, Q3, Q4, Q5, Q9, Q7, Q8 may be provided by a variety of switching devices, such as those described above with reference to FIG. 5 controlled using any of a number of different control circuit implementations, such as those described above with reference to FIG. 5. - FIG. 16 illustrates an exemplary implementation of an AC
voltage generating circuit 450 that may also act as a combined battery converter/battery charger circuit. The ACvoltage generating circuit 450 includes four switching transistors Qa, Qb, Qc, Qd connected in a bridge configuration between the transformer winding Lb and thebattery 460. The transistors Qa, Qb, Qc, Qd are controlled by aswitch control circuit 452. A current limiting inductor Lcl is connected in series with thebattery 460 and the switch SB, and a storage capacitor CS is connected across the series combination. - The
switch control circuit 452 may control switching operations of the transistors Qa, Qb, Qc, Qd responsive to a variety of state inputs, such as a sensed voltage of thebattery 460, and sensed AC input and output voltages for a battery converter in which the ACvoltage generating circuit 450 is included. In a battery powered mode, for example, theswitch control circuit 452 may be operated to selectively switch the transistors Qa, Qb, Qc, Qd such that a DC voltage produced by thebattery 460 across the storage capacitor CS is inverted, producing an AC voltage across the winding Lb. This may be achieved, for example, by switching a first pair of transistors Qa, Qb, in a complementary pattern to a second pair of transistors Qc, Qd at the desired AC line frequency. - Referring to FIGS. 13 and 16, in a charging mode, an AC voltage may be applied to the first winding La of the transformer T1 by action of the fifth and
sixth transistors balancer circuit 430, inducing an AC voltage on the second winding Lb. Theswitch control circuit 452 may selectively switch the transistors Qa, Qb, Qc, Qd such that the AC voltage induced on the winding Lb is rectified to produce a DC voltage across the storage capacitor CS that can be used to charge thebattery 460. The magnitude of the voltage across thebattery 460 can be controlled by the manner in which the transistors Qa, Qb, Qc, Qd are operated, such that when the battery approaches a full charge the current flowing into thebattery 460 can be reduced to avoid overcharging. Alternatively, the switch SB can be opened. - It will be appreciated that the embodiment of FIG. 16 represents an exemplary implementation, and that other circuit implementations fall within the scope of the present invention. For example, the switching functions of the transistors Qa, Qb, Qc, Qd may be provided by any of a number of different switching devices such as those described in relation to FIG. 5, controlled using any of a number of different implementations of the
switch control circuit 452, such as ones similar to those described for theswitch control circuit 440 of FIG. 5. It will be appreciated that functions of theswitch control circuits voltage generating circuit 450 which provide only inversion of the DC voltage produced by the battery may also be used with the present invention. Such a circuit may provide battery charging via an ancillary battery charging circuit (not shown). - In the drawings and specification, there have been disclosed typical preferred embodiments of the invention and, although specific terms are employed, they are used in a generic and descriptive sense only and not for purposes of limitation, the scope of the invention being set forth in the following claims.
Claims (42)
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US09/756,505 US6314007B2 (en) | 1999-08-13 | 2001-01-08 | Multi-mode power converters incorporating balancer circuits and methods of operation thereof |
Applications Claiming Priority (3)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US37418099A | 1999-08-13 | 1999-08-13 | |
US57504200A | 2000-05-19 | 2000-05-19 | |
US09/756,505 US6314007B2 (en) | 1999-08-13 | 2001-01-08 | Multi-mode power converters incorporating balancer circuits and methods of operation thereof |
Related Parent Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US57504200A Continuation | 1999-08-13 | 2000-05-19 |
Publications (2)
Publication Number | Publication Date |
---|---|
US20010001535A1 true US20010001535A1 (en) | 2001-05-24 |
US6314007B2 US6314007B2 (en) | 2001-11-06 |
Family
ID=23475661
Family Applications (2)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US09/401,423 Expired - Lifetime US6160722A (en) | 1999-08-13 | 1999-09-22 | Uninterruptible power supplies with dual-sourcing capability and methods of operation thereof |
US09/756,505 Expired - Lifetime US6314007B2 (en) | 1999-08-13 | 2001-01-08 | Multi-mode power converters incorporating balancer circuits and methods of operation thereof |
Family Applications Before (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US09/401,423 Expired - Lifetime US6160722A (en) | 1999-08-13 | 1999-09-22 | Uninterruptible power supplies with dual-sourcing capability and methods of operation thereof |
Country Status (9)
Country | Link |
---|---|
US (2) | US6160722A (en) |
EP (1) | EP1076403A3 (en) |
JP (1) | JP2001086765A (en) |
CN (1) | CN1284777A (en) |
AU (1) | AU764387B2 (en) |
BR (1) | BR0003570A (en) |
CA (1) | CA2314782C (en) |
DE (1) | DE1076403T1 (en) |
MX (1) | MXPA00007910A (en) |
Cited By (17)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
EP1398867A1 (en) * | 2002-09-10 | 2004-03-17 | ABB Schweiz AG | Device for maintaining the voltage of an electric AC grid and method of operating the device |
US20060250724A1 (en) * | 2005-05-09 | 2006-11-09 | Katsunori Hayashi | Disk array device |
US20080012426A1 (en) * | 2006-07-12 | 2008-01-17 | Delta Electronics, Inc. | Method of controlling an uninterruptible power supply apparatus |
US20080089042A1 (en) * | 2005-05-13 | 2008-04-17 | Abb Research Ltd | Electronic circuit arrangement for control purposes |
US20090256534A1 (en) * | 2008-04-14 | 2009-10-15 | Twisthink, L.L.C. | Power supply control method and apparatus |
AU2009232242B2 (en) * | 2008-04-02 | 2014-04-03 | Schneider Electric It Corporation | Non-isolated charger with bi-polar inputs |
US20150048080A1 (en) * | 2008-09-15 | 2015-02-19 | The Boeing Company | Methods for fabrication of thermoplastic components |
US20170047773A1 (en) * | 2014-05-02 | 2017-02-16 | Schneider Electric It Corporation | Dc link voltage control |
US20180287504A1 (en) * | 2017-03-31 | 2018-10-04 | Schneider Electric It Corporation | Bi-directional dc-dc converter with load and source synchronized power control |
US10516342B1 (en) * | 2018-12-10 | 2019-12-24 | National Chung-Shan Institute Of Science And Technology | Three arm rectifier and inverter circuit |
US10574086B2 (en) * | 2016-04-08 | 2020-02-25 | Rhombus Energy Solutions, Inc. | Nonlinear control algorithm and system for a single-phase AC-AC converter with bidirectional isolated DC-DC converter |
CN112436723A (en) * | 2019-08-09 | 2021-03-02 | 株洲中车时代电气股份有限公司 | Method for inhibiting intermediate voltage oscillation of traction main circuit and traction main circuit |
US11157430B2 (en) * | 2019-12-19 | 2021-10-26 | Schneider Electric It Corporation | DC-DC power converter with four way power conversion |
US11201561B1 (en) * | 2018-03-02 | 2021-12-14 | Apple Inc. | Symmetric hybrid converters |
US20220014013A1 (en) * | 2020-07-07 | 2022-01-13 | Qingchang ZHONG | Power Electronic Converter with a Ground Fault Detection Unit that Shares a Common Ground with both DC Ports and AC Ports |
EP3847741A4 (en) * | 2018-09-06 | 2022-05-25 | Cornell University | High power density power converter and uninterruptible power supply circuit and methods |
US11374501B1 (en) * | 2021-03-26 | 2022-06-28 | Product Development Associates, Inc. | Phase balancer including power conversion circuits |
Families Citing this family (165)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US6630750B2 (en) | 1999-12-16 | 2003-10-07 | Jomahip, Llc | Spare bus power plant |
EP1262008B1 (en) * | 2000-01-28 | 2009-09-09 | Cummins Generator Technologies Limited | Ac power generating system |
US20010045779A1 (en) * | 2000-05-26 | 2001-11-29 | Huey Lee | Intelligent power system |
CA2410729A1 (en) * | 2000-05-31 | 2001-12-06 | Sure Power Corporation | Power system utilizing a dc bus |
US6603672B1 (en) * | 2000-11-10 | 2003-08-05 | Ballard Power Systems Corporation | Power converter system |
DE10103144A1 (en) * | 2001-01-24 | 2002-08-01 | Infineon Technologies Ag | Half-bridge circuit |
US6605879B2 (en) * | 2001-04-19 | 2003-08-12 | Powerware Corporation | Battery charger control circuit and an uninterruptible power supply utilizing same |
WO2002099948A1 (en) * | 2001-06-05 | 2002-12-12 | Koninklijke Philips Electronics N.V. | Power supply architecture with controlled power-on and power-off sequence |
DE60224477T2 (en) * | 2001-06-26 | 2009-02-26 | Sanyo Denki Co., Ltd. | Uninterruptible power supply and method for turning off an AC switch for an uninterruptible power supply |
WO2003032466A1 (en) * | 2001-10-03 | 2003-04-17 | Mitsubishi Denki Kabushiki Kaisha | Uninterruptible power supply and its starting method |
US20030076696A1 (en) * | 2001-10-18 | 2003-04-24 | Delta Electronics, Inc. | Device of uninterruptible power supply |
CN100541994C (en) * | 2002-03-25 | 2009-09-16 | 电力设备公司 | Utilize the power conversion device and the method for equalizer circuit |
KR20030083374A (en) * | 2002-04-22 | 2003-10-30 | 유지고하라 | Economy device of electric energy |
AU2003247484A1 (en) * | 2002-06-04 | 2003-12-19 | Sure Power Corporation | Load break dc power disconnect |
CN1333506C (en) * | 2002-08-14 | 2007-08-22 | 艾默生网络能源有限公司 | Non interrupted power supply system having bus uniform voltage function |
US7545120B2 (en) * | 2003-07-29 | 2009-06-09 | Dell Products L.P. | AC-DC adapter and battery charger integration for portable information handling systems |
US7592716B2 (en) * | 2003-07-29 | 2009-09-22 | Dell Products L.P. | Information handling system including a battery that reduces a voltage fluctuation |
US7084525B2 (en) * | 2003-08-28 | 2006-08-01 | Delphi Technologies, Inc. | Power system to transfer power between a plurality of power sources |
US7049870B2 (en) * | 2004-01-09 | 2006-05-23 | Potentia Semiconductor Corporation | Digital controllers for DC converters |
CA2502798C (en) * | 2004-03-31 | 2011-06-14 | University Of New Brunswick | Single-stage buck-boost inverter |
JP4556516B2 (en) * | 2004-07-08 | 2010-10-06 | 富士電機システムズ株式会社 | Power converter |
ATE484877T1 (en) | 2004-07-12 | 2010-10-15 | Siemens Ag | METHOD FOR OPERATING AN INVERTER AND ARRANGEMENT FOR IMPLEMENTING THE METHOD |
US7391188B2 (en) * | 2004-08-02 | 2008-06-24 | Jacobs James K | Current prediction in a switching power supply |
US20060132111A1 (en) * | 2004-08-02 | 2006-06-22 | Jacobs James K | Power supply with multiple modes of operation |
US7391132B2 (en) * | 2004-12-03 | 2008-06-24 | Huei-Jung Chen | Methods and apparatus providing double conversion/series-parallel hybrid operation in uninterruptible power supplies |
US7208891B2 (en) * | 2005-05-06 | 2007-04-24 | York International Corp. | Variable speed drive for a chiller system |
US7382114B2 (en) * | 2005-06-07 | 2008-06-03 | Intersil Americas Inc. | PFM-PWM DC-DC converter providing DC offset correction to PWM error amplifier and equalizing regulated voltage conditions when transitioning between PFM and PWM modes |
US7081734B1 (en) | 2005-09-02 | 2006-07-25 | York International Corporation | Ride-through method and system for HVACandR chillers |
US7332885B2 (en) * | 2005-09-02 | 2008-02-19 | Johnson Controls Technology Company | Ride-through method and system for HVAC&R chillers |
US7692417B2 (en) | 2005-09-19 | 2010-04-06 | Skyworks Solutions, Inc. | Switched mode power converter |
US10693415B2 (en) | 2007-12-05 | 2020-06-23 | Solaredge Technologies Ltd. | Testing of a photovoltaic panel |
US11881814B2 (en) | 2005-12-05 | 2024-01-23 | Solaredge Technologies Ltd. | Testing of a photovoltaic panel |
US20070151272A1 (en) * | 2006-01-03 | 2007-07-05 | York International Corporation | Electronic control transformer using DC link voltage |
EP1806819A1 (en) * | 2006-01-05 | 2007-07-11 | Constructions Electroniques + Telecommunications, en abrégé "C.E.+T" | Backup power system |
US7508094B2 (en) * | 2006-03-17 | 2009-03-24 | Eaton Corporation | UPS systems having multiple operation modes and methods of operating same |
US7869771B2 (en) * | 2006-06-23 | 2011-01-11 | Broadcom Corporation | Multi-band transformer for wireless transmitter |
US8280325B2 (en) * | 2006-06-23 | 2012-10-02 | Broadcom Corporation | Configurable transmitter |
US7336513B1 (en) | 2006-09-12 | 2008-02-26 | National Chung Cheng University | Method of compensating output voltage distortion of half-bridge inverter and device based on the method |
EP2074692B1 (en) * | 2006-09-25 | 2014-05-28 | Robert Bosch GmbH | Power inverter circuit for adjusting symmetry of the ac-voltage without load-coupling |
US7710081B2 (en) | 2006-10-27 | 2010-05-04 | Direct Drive Systems, Inc. | Electromechanical energy conversion systems |
US8319483B2 (en) | 2007-08-06 | 2012-11-27 | Solaredge Technologies Ltd. | Digital average input current control in power converter |
US8963369B2 (en) | 2007-12-04 | 2015-02-24 | Solaredge Technologies Ltd. | Distributed power harvesting systems using DC power sources |
US11855231B2 (en) | 2006-12-06 | 2023-12-26 | Solaredge Technologies Ltd. | Distributed power harvesting systems using DC power sources |
US11888387B2 (en) | 2006-12-06 | 2024-01-30 | Solaredge Technologies Ltd. | Safety mechanisms, wake up and shutdown methods in distributed power installations |
US9088178B2 (en) | 2006-12-06 | 2015-07-21 | Solaredge Technologies Ltd | Distributed power harvesting systems using DC power sources |
US8319471B2 (en) | 2006-12-06 | 2012-11-27 | Solaredge, Ltd. | Battery power delivery module |
US8816535B2 (en) | 2007-10-10 | 2014-08-26 | Solaredge Technologies, Ltd. | System and method for protection during inverter shutdown in distributed power installations |
US8384243B2 (en) | 2007-12-04 | 2013-02-26 | Solaredge Technologies Ltd. | Distributed power harvesting systems using DC power sources |
US11309832B2 (en) | 2006-12-06 | 2022-04-19 | Solaredge Technologies Ltd. | Distributed power harvesting systems using DC power sources |
US8473250B2 (en) | 2006-12-06 | 2013-06-25 | Solaredge, Ltd. | Monitoring of distributed power harvesting systems using DC power sources |
US11569659B2 (en) | 2006-12-06 | 2023-01-31 | Solaredge Technologies Ltd. | Distributed power harvesting systems using DC power sources |
US11687112B2 (en) | 2006-12-06 | 2023-06-27 | Solaredge Technologies Ltd. | Distributed power harvesting systems using DC power sources |
US11728768B2 (en) | 2006-12-06 | 2023-08-15 | Solaredge Technologies Ltd. | Pairing of components in a direct current distributed power generation system |
US8013472B2 (en) | 2006-12-06 | 2011-09-06 | Solaredge, Ltd. | Method for distributed power harvesting using DC power sources |
US8947194B2 (en) | 2009-05-26 | 2015-02-03 | Solaredge Technologies Ltd. | Theft detection and prevention in a power generation system |
US11735910B2 (en) | 2006-12-06 | 2023-08-22 | Solaredge Technologies Ltd. | Distributed power system using direct current power sources |
US8149579B2 (en) * | 2008-03-28 | 2012-04-03 | Johnson Controls Technology Company | Cooling member |
US7764041B2 (en) | 2007-01-22 | 2010-07-27 | Johnson Controls Technology Company | System and method to extend synchronous operation of an active converter in a variable speed drive |
US8495890B2 (en) * | 2007-01-22 | 2013-07-30 | Johnson Controls Technology Company | Cooling member |
US7800924B2 (en) * | 2007-03-27 | 2010-09-21 | Eaton Corporation | Power converter apparatus and methods using neutral coupling circuits with interleaved operation |
WO2008137276A1 (en) | 2007-05-08 | 2008-11-13 | Johnson Controls Technology Company | Variable speed drive |
US7573732B2 (en) * | 2007-05-25 | 2009-08-11 | General Electric Company | Protective circuit and method for multi-level converter |
EP3324505B1 (en) | 2007-10-15 | 2023-06-07 | Ampt, Llc | Systems for highly efficient solar power |
KR101405874B1 (en) * | 2007-10-22 | 2014-06-12 | 지멘스 악티엔게젤샤프트 | Electrical switchgear,particularly for connecting generators and thrusters in dynamically positioned vessels |
US7919953B2 (en) * | 2007-10-23 | 2011-04-05 | Ampt, Llc | Solar power capacitor alternative switch circuitry system for enhanced capacitor life |
US8174853B2 (en) * | 2007-10-30 | 2012-05-08 | Johnson Controls Technology Company | Variable speed drive |
US7957166B2 (en) * | 2007-10-30 | 2011-06-07 | Johnson Controls Technology Company | Variable speed drive |
CN101933209B (en) | 2007-12-05 | 2015-10-21 | 太阳能安吉有限公司 | Release mechanism in distributed electrical power apparatus, to wake up and method for closing |
US8049523B2 (en) | 2007-12-05 | 2011-11-01 | Solaredge Technologies Ltd. | Current sensing on a MOSFET |
US11264947B2 (en) | 2007-12-05 | 2022-03-01 | Solaredge Technologies Ltd. | Testing of a photovoltaic panel |
TW200934032A (en) * | 2008-01-30 | 2009-08-01 | Tsann Kuen Entpr Co Ltd | Induction type electrical power supply |
TW200935704A (en) * | 2008-02-01 | 2009-08-16 | Tsann Kuen Entpr Co Ltd | Induction type electrical power structure and system |
WO2009136358A1 (en) | 2008-05-05 | 2009-11-12 | Solaredge Technologies Ltd. | Direct current power combiner |
JP5190683B2 (en) * | 2008-06-11 | 2013-04-24 | サンケン電気株式会社 | AC power supply |
US8183734B2 (en) | 2008-07-28 | 2012-05-22 | Direct Drive Systems, Inc. | Hybrid winding configuration of an electric machine |
WO2010013322A1 (en) | 2008-07-30 | 2010-02-04 | 東芝三菱電機産業システム株式会社 | Power conversion device |
US8336323B2 (en) * | 2008-10-03 | 2012-12-25 | Johnson Controls Technology Company | Variable speed drive with pulse-width modulated speed control |
US20110210611A1 (en) * | 2008-10-10 | 2011-09-01 | Ampt, Llc | Novel Solar Power Circuits |
US8212402B2 (en) | 2009-01-27 | 2012-07-03 | American Power Conversion Corporation | System and method for limiting losses in an uninterruptible power supply |
WO2010111433A2 (en) | 2009-03-25 | 2010-09-30 | Powergetics, Inc. | Bidirectional energy converter |
JP5374210B2 (en) * | 2009-03-31 | 2013-12-25 | 本田技研工業株式会社 | DC / DC converter and power supply system using the same |
WO2010120315A1 (en) | 2009-04-17 | 2010-10-21 | Ampt, Llc | Methods and apparatus for adaptive operation of solar power systems |
US8604384B2 (en) | 2009-06-18 | 2013-12-10 | Illinois Tool Works Inc. | System and methods for efficient provision of arc welding power source |
WO2011008505A2 (en) | 2009-06-29 | 2011-01-20 | Powergetics, Inc | High speed feedback adjustment of power charge/discharge from energy storage system |
JP2012532583A (en) | 2009-06-29 | 2012-12-13 | パワージェティクス, インコーポレイテッド | High-speed feedback for power load reduction using variable generators |
WO2011049985A1 (en) | 2009-10-19 | 2011-04-28 | Ampt, Llc | Novel solar panel string converter topology |
US8148942B2 (en) * | 2009-11-05 | 2012-04-03 | O2Micro International Limited | Charging systems with cell balancing functions |
US8575778B2 (en) * | 2010-01-12 | 2013-11-05 | Ford Global Technologies, Llc | Variable voltage converter (VVC) with integrated battery charger |
KR101131664B1 (en) * | 2010-01-28 | 2012-03-28 | 장석호 | Charging equipment using switching arrangement and charging/discharging |
DE102010003797A1 (en) * | 2010-04-09 | 2011-10-13 | Tridonic Ag | Modular LED lighting system with emergency light function |
US20120074786A1 (en) | 2010-05-13 | 2012-03-29 | Eaton Corporation | Uninterruptible power supply systems and methods using isolated interface for variably available power source |
CN102005938B (en) * | 2010-08-25 | 2013-01-30 | 力博特公司 | Control method of bridge arm in UPS (uninterruptible power supply) in case of zero-crossing operation |
US8760078B2 (en) * | 2010-10-04 | 2014-06-24 | Earl W. McCune, Jr. | Power conversion and control systems and methods for solid-state lighting |
US10673229B2 (en) | 2010-11-09 | 2020-06-02 | Solaredge Technologies Ltd. | Arc detection and prevention in a power generation system |
US10673222B2 (en) | 2010-11-09 | 2020-06-02 | Solaredge Technologies Ltd. | Arc detection and prevention in a power generation system |
US10230310B2 (en) | 2016-04-05 | 2019-03-12 | Solaredge Technologies Ltd | Safety switch for photovoltaic systems |
GB2485527B (en) | 2010-11-09 | 2012-12-19 | Solaredge Technologies Ltd | Arc detection and prevention in a power generation system |
CA2822864A1 (en) | 2010-12-22 | 2012-06-28 | Converteam Technology Ltd. | Capacitor balancing circuit and control method for an electronic device such as a multilevel power inverter |
BR112013015890A2 (en) | 2010-12-22 | 2017-09-19 | Ge Energy Power Conversion Technology Ltd | mechanical arrangement of a multilevel power converter circuit. |
GB2483317B (en) | 2011-01-12 | 2012-08-22 | Solaredge Technologies Ltd | Serially connected inverters |
US8854004B2 (en) * | 2011-01-12 | 2014-10-07 | Samsung Sdi Co., Ltd. | Energy storage system and controlling method thereof |
US8816533B2 (en) * | 2011-02-16 | 2014-08-26 | Eaton Corporation | Uninterruptible power supply systems and methods using an isolated neutral reference |
US8730691B2 (en) | 2011-05-11 | 2014-05-20 | Eaton Corporation | Power conversion apparatus and methods employing variable-level inverters |
JP5800130B2 (en) * | 2011-06-20 | 2015-10-28 | 富士電機株式会社 | DC power supply system |
US8570005B2 (en) | 2011-09-12 | 2013-10-29 | Solaredge Technologies Ltd. | Direct current link circuit |
US9106103B2 (en) | 2011-09-23 | 2015-08-11 | Eaton Corporation | Unintteruptible power supply systems and methods employing on-demand energy storage |
US8803570B2 (en) | 2011-12-29 | 2014-08-12 | Stem, Inc | Multiphase electrical power assignment at minimal loss |
US8774977B2 (en) | 2011-12-29 | 2014-07-08 | Stem, Inc. | Multiphase electrical power construction and assignment at minimal loss |
US8922192B2 (en) | 2011-12-30 | 2014-12-30 | Stem, Inc. | Multiphase electrical power phase identification |
GB2498365A (en) | 2012-01-11 | 2013-07-17 | Solaredge Technologies Ltd | Photovoltaic module |
GB2498790A (en) | 2012-01-30 | 2013-07-31 | Solaredge Technologies Ltd | Maximising power in a photovoltaic distributed power system |
US9853565B2 (en) | 2012-01-30 | 2017-12-26 | Solaredge Technologies Ltd. | Maximized power in a photovoltaic distributed power system |
GB2498791A (en) | 2012-01-30 | 2013-07-31 | Solaredge Technologies Ltd | Photovoltaic panel circuitry |
FR2986917B1 (en) * | 2012-02-13 | 2014-02-21 | Converteam Technology Ltd | ELECTRIC POWER SUPPLY SYSTEM AND ELECTRIC POWER GENERATION PLANT COMPRISING SUCH A SYSTEM |
GB2499991A (en) | 2012-03-05 | 2013-09-11 | Solaredge Technologies Ltd | DC link circuit for photovoltaic array |
JP5403090B2 (en) * | 2012-03-09 | 2014-01-29 | 富士電機株式会社 | Power converter |
JP5370519B2 (en) * | 2012-03-15 | 2013-12-18 | 富士電機株式会社 | Power converter |
EP2647523A1 (en) * | 2012-04-04 | 2013-10-09 | Volvo Car Corporation | Circuit for charging a battery and for driving a three-phase electrical machine |
US9444320B1 (en) * | 2012-04-16 | 2016-09-13 | Performance Controls, Inc. | Power controller having active voltage balancing of a power supply |
CN102710006B (en) * | 2012-05-18 | 2014-12-24 | 深圳市健网科技有限公司 | Double-power supply system with balance bridge arm |
WO2014011706A1 (en) | 2012-07-09 | 2014-01-16 | Inertech Ip Llc | Transformerless multi-level medium-voltage uninterruptible power supply (ups) systems and methods |
US9490663B1 (en) * | 2012-07-16 | 2016-11-08 | Google Inc. | Apparatus and methodology for battery backup circuit and control in an uninterruptible power supply |
US9406094B2 (en) | 2012-08-14 | 2016-08-02 | Stem Inc. | Method and apparatus for delivering power using external data |
US10782721B2 (en) | 2012-08-27 | 2020-09-22 | Stem, Inc. | Method and apparatus for balancing power on a per phase basis in multi-phase electrical load facilities using an energy storage system |
US11454999B2 (en) | 2012-08-29 | 2022-09-27 | Stem, Inc. | Method and apparatus for automatically reconfiguring multi-phased networked energy storage devices at a site |
RU2513547C1 (en) * | 2012-09-07 | 2014-04-20 | Общество с ограниченной ответственностью "Гамем" (ООО Гамем") | Static reversible converter for power supply of alternating and direct-current consumers |
US10756543B2 (en) | 2012-09-13 | 2020-08-25 | Stem, Inc. | Method and apparatus for stabalizing power on an electrical grid using networked distributed energy storage systems |
US10389126B2 (en) | 2012-09-13 | 2019-08-20 | Stem, Inc. | Method and apparatus for damping power oscillations on an electrical grid using networked distributed energy storage systems |
US9634508B2 (en) | 2012-09-13 | 2017-04-25 | Stem, Inc. | Method for balancing frequency instability on an electric grid using networked distributed energy storage systems |
US10693294B2 (en) | 2012-09-26 | 2020-06-23 | Stem, Inc. | System for optimizing the charging of electric vehicles using networked distributed energy storage systems |
CN102882256B (en) * | 2012-10-12 | 2015-11-11 | 广东易事特电源股份有限公司 | A kind of ups power with double-bus charging circuit |
US9246411B2 (en) * | 2012-10-16 | 2016-01-26 | Rockwell Automation Technologies, Inc. | Regenerative voltage doubler rectifier, voltage sag/swell correction apparatus and operating methods |
RU2505917C1 (en) * | 2012-11-01 | 2014-01-27 | Федеральное государственное бюджетное образовательное учреждение высшего профессионального образования "Национальный минерально-сырьевой университет "Горный" | Self-contained electric power supply system |
WO2014082221A1 (en) * | 2012-11-28 | 2014-06-05 | Eaton Corporation | Multi-level converter apparatus with efficiency improving current bypass |
US9548619B2 (en) | 2013-03-14 | 2017-01-17 | Solaredge Technologies Ltd. | Method and apparatus for storing and depleting energy |
US9941813B2 (en) | 2013-03-14 | 2018-04-10 | Solaredge Technologies Ltd. | High frequency multi-level inverter |
US9397497B2 (en) | 2013-03-15 | 2016-07-19 | Ampt, Llc | High efficiency interleaved solar power supply system |
US9537332B2 (en) | 2013-05-30 | 2017-01-03 | Canara, Inc. | Apparatus, system and method for charge balancing of individual batteries in a string of batteries using battery voltage and temperature, and detecting and preventing thermal runaway |
CN103368231B (en) | 2013-07-05 | 2015-03-11 | 华为技术有限公司 | Uninterruptible power supply circuit |
CN103746425B (en) * | 2014-01-09 | 2016-02-17 | 成都芯源系统有限公司 | Mobile power supply circuit and method thereof |
CN104882913A (en) | 2014-02-27 | 2015-09-02 | 伊顿制造(格拉斯哥)有限合伙莫尔日分支机构 | UPS circuit |
US9318974B2 (en) | 2014-03-26 | 2016-04-19 | Solaredge Technologies Ltd. | Multi-level inverter with flying capacitor topology |
CN104092277A (en) | 2014-04-23 | 2014-10-08 | 矽力杰半导体技术(杭州)有限公司 | Power supply circuit including bidirectional DC converter and control method thereof |
WO2016065087A1 (en) | 2014-10-21 | 2016-04-28 | Inertech Ip Llc | Systems and methods for controlling multi-level diode-clamped inverters using space vector pulse width modulation (svpwm) |
US9479004B2 (en) * | 2015-03-13 | 2016-10-25 | Active-Semi, Inc. | Buck/boost circuit that charges and discharges multi-cell batteries of a power bank device |
US10120034B2 (en) | 2015-10-07 | 2018-11-06 | Canara, Inc. | Battery string monitoring system |
US10931190B2 (en) | 2015-10-22 | 2021-02-23 | Inertech Ip Llc | Systems and methods for mitigating harmonics in electrical systems by using active and passive filtering techniques |
US10734918B2 (en) | 2015-12-28 | 2020-08-04 | Illinois Tool Works Inc. | Systems and methods for efficient provision of arc welding power source |
US11177663B2 (en) | 2016-04-05 | 2021-11-16 | Solaredge Technologies Ltd. | Chain of power devices |
US11018623B2 (en) | 2016-04-05 | 2021-05-25 | Solaredge Technologies Ltd. | Safety switch for photovoltaic systems |
US10454381B2 (en) * | 2016-09-15 | 2019-10-22 | Virginia Tech Intellectual Properties, Inc. | Variable DC link converter and transformer for wide output voltage range applications |
RU180664U1 (en) * | 2017-05-30 | 2018-06-20 | Общество С Ограниченной Ответственностью Научно-Производственное Предприятие "Томская Электронная Компания" | Uninterruptible Power Supply |
TWI658678B (en) | 2017-12-25 | 2019-05-01 | 台達電子工業股份有限公司 | Uninterruptible power supply apparatus |
RU191898U1 (en) * | 2018-02-20 | 2019-08-27 | Общество с ограниченной ответственностью "АЕДОН" | MODULAR SECONDARY POWER SUPPLY |
RU186995U1 (en) * | 2018-06-07 | 2019-02-12 | Общество с ограниченной ответственностью "АСХ" (ООО "АСХ") | EXECUTIVE DEVICE FOR THE AUTOMATED SYSTEM OF DATA COLLECTION FROM ACCOUNTING AND RESOURCE MANAGEMENT IN HOUSING AND COMMUNAL SERVICES |
CN111181390A (en) * | 2018-11-13 | 2020-05-19 | 深圳市贝贝特科技实业有限公司 | Circuit equalizer and unmanned aerial vehicle |
CN111342677A (en) * | 2018-12-18 | 2020-06-26 | 协欣电子工业股份有限公司 | Power converter |
RU187091U1 (en) * | 2018-12-20 | 2019-02-19 | Федеральное государственное бюджетное образовательное учреждение высшего образования "Национальный исследовательский Мордовский государственный университет им. Н.П. Огарёва" | Three-phase rectifier-inverter module control device |
EP3672054B1 (en) | 2018-12-21 | 2021-06-16 | Eltek AS | Power converter and method of controlling a power converter |
EP3683943B1 (en) * | 2019-01-17 | 2021-03-24 | Cotek Electronic Ind. Co., Ltd. | Power converter |
RU2732280C1 (en) * | 2019-12-26 | 2020-09-15 | Акционерное общество "Научно-производственная корпорация "Космические системы мониторинга, информационно-управляющие и электромеханические комплексы" имени А.Г. Иосифьяна" АО "Корпорация "ВНИИЭМ" | Uninterrupted power supply - static reversible converter for supply of alternating and direct current consumers and charging (recharging) of storage battery |
KR102379157B1 (en) * | 2020-11-04 | 2022-03-25 | 한국항공우주연구원 | Integrated dc/dc and ac/dc converter system |
RU203769U1 (en) * | 2020-12-17 | 2021-04-21 | Общество с ограниченной ответственностью Научно-производственное предприятие «Томская электронная компания» | Uninterruptible Power Supply |
TWI794926B (en) * | 2021-08-12 | 2023-03-01 | 台達電子工業股份有限公司 | General-purpose current detection circuit and method of current detection the same |
CN115776241B (en) * | 2022-06-23 | 2023-07-21 | 中国科学院电工研究所 | AC-AC converter based on switch unit and control method |
Family Cites Families (27)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3775663A (en) * | 1972-08-24 | 1973-11-27 | Gen Electric | Inverter with electronically controlled neutral terminal |
US4502106A (en) * | 1983-10-17 | 1985-02-26 | Sundstrand Corporation | Current source sine wave inverter |
US4507724A (en) * | 1983-10-17 | 1985-03-26 | Sundstrand Corporation | Pulse width modulated inverter for unbalanced and variable power factor loads |
FI81465C (en) | 1986-12-22 | 1990-10-10 | Kone Oy | Device for coupling an accumulator battery to an elevator inverter DC direct current circuit |
US5237208A (en) * | 1988-10-25 | 1993-08-17 | Nishimu Electronics Industries Co., Ltd. | Apparatus for parallel operation of triport uninterruptable power source devices |
US5017800A (en) * | 1989-09-29 | 1991-05-21 | Wisconsin Alumni Research Foundation | AC to DC to AC power conversion apparatus with few active switches and input and output control |
US5126585A (en) * | 1990-06-19 | 1992-06-30 | Auckland Uniservices Limited | Uninterruptible power supplies |
US5111374A (en) * | 1990-06-22 | 1992-05-05 | The University Of Tennessee Research Corp. | High frequency quasi-resonant DC voltage notching scheme of a PWM voltage fed inverter for AC motor drives |
US5111376A (en) * | 1990-11-01 | 1992-05-05 | Sundstrand Corporation | Voltage balancing circuit |
US5229650A (en) * | 1990-11-07 | 1993-07-20 | Yuasa Battery Company Limited | Uniterruptible power system |
US5343079A (en) | 1991-02-25 | 1994-08-30 | Regents Of The University Of Minnesota | Standby power supply with load-current harmonics neutralizer |
US5119283A (en) * | 1991-06-10 | 1992-06-02 | General Electric Company | High power factor, voltage-doubler rectifier |
US5253157A (en) * | 1992-02-06 | 1993-10-12 | Premier Power, Inc. | Half-bridge inverter with capacitive voltage equalizer |
GB9400499D0 (en) * | 1994-01-12 | 1994-03-09 | Magnum Power Solutions Ltd | Improved uninterruptible power supply |
US5444356A (en) * | 1994-03-03 | 1995-08-22 | Miller Electric Mfg. Co. | Buck converter having a variable output and method for buck converting power with a variable output |
JP3203464B2 (en) * | 1994-06-11 | 2001-08-27 | サンケン電気株式会社 | AC power converter |
US5502630A (en) * | 1994-07-19 | 1996-03-26 | Transistor Devices, Inc. | Power factor corrected rectification |
JP3185846B2 (en) * | 1994-10-26 | 2001-07-11 | サンケン電気株式会社 | Power converter |
US5610805A (en) * | 1995-01-10 | 1997-03-11 | Cambridge Continuous Power | Uninterruptible power supply with a back-up battery coupled across the a.c. input |
US5644483A (en) * | 1995-05-22 | 1997-07-01 | Lockheed Martin Energy Systems, Inc. | Voltage balanced multilevel voltage source converter system |
AUPO009496A0 (en) * | 1996-05-24 | 1996-06-20 | Unisearch Limited | Photovoltaic to grid interconnection |
JP3414143B2 (en) * | 1996-08-15 | 2003-06-09 | 松下電工株式会社 | Power supply |
JP3497673B2 (en) * | 1996-08-30 | 2004-02-16 | 株式会社三社電機製作所 | Uninterruptible power supply cross current prevention circuit |
WO1998034314A1 (en) * | 1997-01-31 | 1998-08-06 | Silverline Power Conversion, Llc | Uninterruptible power supply |
JPH1169814A (en) * | 1997-08-14 | 1999-03-09 | Toshiba Corp | Power supply units and control circuit for parallel operations thereof |
US6005362A (en) * | 1998-02-13 | 1999-12-21 | The Texas A&M University Systems | Method and system for ride-through of an adjustable speed drive for voltage sags and short-term power interruption |
US6115276A (en) | 1998-11-24 | 2000-09-05 | Lucent Technologies Inc. | AC bus system with battery charger/inverter backup |
-
1999
- 1999-09-22 US US09/401,423 patent/US6160722A/en not_active Expired - Lifetime
-
2000
- 2000-07-27 DE DE1076403T patent/DE1076403T1/en active Pending
- 2000-07-27 EP EP00115097A patent/EP1076403A3/en not_active Withdrawn
- 2000-08-01 CA CA002314782A patent/CA2314782C/en not_active Expired - Lifetime
- 2000-08-09 JP JP2000240897A patent/JP2001086765A/en active Pending
- 2000-08-11 AU AU51954/00A patent/AU764387B2/en not_active Ceased
- 2000-08-11 MX MXPA00007910A patent/MXPA00007910A/en active IP Right Grant
- 2000-08-14 CN CN00124248A patent/CN1284777A/en active Pending
- 2000-08-14 BR BR0003570-0A patent/BR0003570A/en not_active Application Discontinuation
-
2001
- 2001-01-08 US US09/756,505 patent/US6314007B2/en not_active Expired - Lifetime
Cited By (26)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
EP1398867A1 (en) * | 2002-09-10 | 2004-03-17 | ABB Schweiz AG | Device for maintaining the voltage of an electric AC grid and method of operating the device |
US7667918B2 (en) * | 2005-05-09 | 2010-02-23 | Hitachi, Ltd. | Disk array device |
US20060250724A1 (en) * | 2005-05-09 | 2006-11-09 | Katsunori Hayashi | Disk array device |
US20080089042A1 (en) * | 2005-05-13 | 2008-04-17 | Abb Research Ltd | Electronic circuit arrangement for control purposes |
US8050053B2 (en) * | 2005-05-13 | 2011-11-01 | Abb Research Ltd | Electronic circuit arrangement for control purposes |
US20100308658A1 (en) * | 2006-07-12 | 2010-12-09 | Delta Electronics, Inc. | Uninterruptible power supply apparatus |
US8143743B2 (en) | 2006-07-12 | 2012-03-27 | Delta Electronics, Inc. | Uninterruptible power supply apparatus |
US20080012426A1 (en) * | 2006-07-12 | 2008-01-17 | Delta Electronics, Inc. | Method of controlling an uninterruptible power supply apparatus |
AU2009232242B2 (en) * | 2008-04-02 | 2014-04-03 | Schneider Electric It Corporation | Non-isolated charger with bi-polar inputs |
US20090256534A1 (en) * | 2008-04-14 | 2009-10-15 | Twisthink, L.L.C. | Power supply control method and apparatus |
US20150048080A1 (en) * | 2008-09-15 | 2015-02-19 | The Boeing Company | Methods for fabrication of thermoplastic components |
US20170047773A1 (en) * | 2014-05-02 | 2017-02-16 | Schneider Electric It Corporation | Dc link voltage control |
US10476299B2 (en) * | 2014-05-02 | 2019-11-12 | Schneider Electric It Corporation | DC link voltage control |
US10574086B2 (en) * | 2016-04-08 | 2020-02-25 | Rhombus Energy Solutions, Inc. | Nonlinear control algorithm and system for a single-phase AC-AC converter with bidirectional isolated DC-DC converter |
US20180287504A1 (en) * | 2017-03-31 | 2018-10-04 | Schneider Electric It Corporation | Bi-directional dc-dc converter with load and source synchronized power control |
US10811987B2 (en) * | 2017-03-31 | 2020-10-20 | Schneider Electric It Corporation | Bi-directional DC-DC converter with load and source synchronized power control |
US11201561B1 (en) * | 2018-03-02 | 2021-12-14 | Apple Inc. | Symmetric hybrid converters |
US11552578B1 (en) | 2018-03-02 | 2023-01-10 | Apple Inc. | Symmetric hybrid converters |
US11888410B1 (en) | 2018-03-02 | 2024-01-30 | Apple Inc. | Symmetric hybrid converters |
EP3847741A4 (en) * | 2018-09-06 | 2022-05-25 | Cornell University | High power density power converter and uninterruptible power supply circuit and methods |
US11381159B2 (en) | 2018-09-06 | 2022-07-05 | Cornell University | High power density power converter and uninterruptible power supply circuit and methods |
US10516342B1 (en) * | 2018-12-10 | 2019-12-24 | National Chung-Shan Institute Of Science And Technology | Three arm rectifier and inverter circuit |
CN112436723A (en) * | 2019-08-09 | 2021-03-02 | 株洲中车时代电气股份有限公司 | Method for inhibiting intermediate voltage oscillation of traction main circuit and traction main circuit |
US11157430B2 (en) * | 2019-12-19 | 2021-10-26 | Schneider Electric It Corporation | DC-DC power converter with four way power conversion |
US20220014013A1 (en) * | 2020-07-07 | 2022-01-13 | Qingchang ZHONG | Power Electronic Converter with a Ground Fault Detection Unit that Shares a Common Ground with both DC Ports and AC Ports |
US11374501B1 (en) * | 2021-03-26 | 2022-06-28 | Product Development Associates, Inc. | Phase balancer including power conversion circuits |
Also Published As
Publication number | Publication date |
---|---|
DE1076403T1 (en) | 2002-10-17 |
US6314007B2 (en) | 2001-11-06 |
US6160722A (en) | 2000-12-12 |
BR0003570A (en) | 2001-04-03 |
CN1284777A (en) | 2001-02-21 |
JP2001086765A (en) | 2001-03-30 |
AU764387B2 (en) | 2003-08-14 |
EP1076403A2 (en) | 2001-02-14 |
AU5195400A (en) | 2001-02-15 |
EP1076403A3 (en) | 2002-01-02 |
CA2314782A1 (en) | 2001-02-13 |
CA2314782C (en) | 2007-06-05 |
MXPA00007910A (en) | 2002-04-24 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US6314007B2 (en) | Multi-mode power converters incorporating balancer circuits and methods of operation thereof | |
US6483730B2 (en) | Power converters with AC and DC operating modes and methods of operation thereof | |
US8004240B2 (en) | Non-isolated charger with bi-polar inputs | |
US5440179A (en) | UPS with bi-directional power flow | |
US5057990A (en) | Bidirectional switching power apparatus with AC or DC output | |
Morrison et al. | A new power-factor-corrected single-transformer UPS design | |
EP1034614B1 (en) | Frequency converter and ups employing the same | |
US20230249564A1 (en) | Charging device and vehicle | |
EP1210758B1 (en) | Uninterruptible power supplies with dual-sourcing capability and methods of operation thereof | |
KR20190115364A (en) | Single and three phase combined charger | |
US6297971B1 (en) | Phase converter | |
EP1264385B1 (en) | Power converters with ac and dc operating modes and methods of operation thereof | |
US20230071003A1 (en) | Power factor correction circuits controlled using adjustable deadtime | |
Lin et al. | Implementation of the AC/AC converter based on neutral-point switch-clamped topology |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
STCF | Information on status: patent grant |
Free format text: PATENTED CASE |
|
FPAY | Fee payment |
Year of fee payment: 4 |
|
FPAY | Fee payment |
Year of fee payment: 8 |
|
FPAY | Fee payment |
Year of fee payment: 12 |
|
AS | Assignment |
Owner name: EATON CORPORATION, OHIO Free format text: MERGER;ASSIGNOR:EATON ELECTRICAL INC.;REEL/FRAME:047442/0286 Effective date: 20081128 Owner name: EATON ELECTRICAL INC., OHIO Free format text: MERGER;ASSIGNOR:EATON POWER QUALITY CORPORATION;REEL/FRAME:047442/0281 Effective date: 20060420 Owner name: EATON INTELLIGENT POWER LIMITED, IRELAND Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:EATON CORPORATION;REEL/FRAME:047442/0300 Effective date: 20171231 Owner name: EATON POWER QUALITY CORPORATION, OHIO Free format text: CHANGE OF NAME;ASSIGNOR:POWERWARE CORPORATION;REEL/FRAME:048100/0764 Effective date: 20041025 |