1229486 玖、發明說明： 相關申請案 本案係基於美國臨時專利申請案第60/391，987號，申請 日2002年，6月25日，名稱「不具電流感應裝置之主動式EMI 5濾波器」以及美國臨時專利申請案第60/408,534號，申請曰 2002年9月5日，名稱「主動式EMI渡波器」且申請該等專 利申請案之權益，並對前述二案請求優先權。 【發明所屬之技術領域3 發明領域 10 本發明係關於電源轉換系統之電磁干擾（EMI)之縮 減’特別係關於電動機驅動系統之EMI的降低。 【 發明背景 高速切換裝置例如兩極性電晶體、MOSFETs及IGBT，s 15 可提供電壓源PWM反相器之載波頻率，結果導致遠更佳的 操作特性。但高速切換造成下列嚴重問題，該等問題係來 自於電壓及/或電流的高速改變： a)地電流經由電動機内部之雜散電容器以及經由長電 纜而逃逸至大地； 20 b)傳導及輻射EMI ; c) 電動機承載電流及軸向電壓；以及 d) 電動機及變壓器之絕緣壽命縮短。 因寄生雜散電容無可避免地存在於負載内部，例如交 流電動機内部以及切換轉換器内部，故當切換裝置改變態 1229486 時，由高速切換引起的電壓及/或電流改變，產生高頻振盡 共模及#权電流。如此每次出現反相器切換事件，對應反 相器輸出端子電位快速相對於地電位移動，共模電流脈波 透過熱阱電動機纜線及電動機繞組接地而於直流鏈路流至 5反相器。此種用於B類（住宅）電動機驅動器之電流脈波振幅 典型為數百毫安培至數安培；脈衝寬度典型為25〇至5〇〇奈 秒。用於A類驅動器（工業用），且依據電動機大小以及電動 機纜線長度而定，脈波電流振幅用於脈波頻寬25〇奈秒至 500奈秒典型為數安培，直至用於脈波寬度丨至2微秒之2〇安 10 培或20安培以上。 共模振盪電流具有頻譜由轉換器切換頻率至數十MHZ 之範圍’振盪電流形成磁場，且將產生輻射電磁干擾(EMi) 遍布各處，如此對無線接收器、醫療器材等電子裝置造成 不良影響。 15 政府有許多規定限制某些電動機應用用途之容許線電 流EMI程度以及容許地電流。如此，於B類住宅（家電）應用 用途，地電流於0至30 kHz頻率範圍（於對數曲線)須維持低 於1至20毫安培；以及導線電流EMI於150 kHz至300 MHz 頻率範圍須維持低於指定值（小於約60 (ΙΒμν)。對於指定為 20 Α類工業用途之電動機驅動用途，地電流的限制較為嚴格， 但線電流EMI於150 kHz至30 MHz範圍仍然有限制。 通常基於被動元件，共模抗流線圈及EMI濾波器無法 完全解決此等問題。由共模電感器及「Y」電容器於輸入交 流線組成之被動式濾波器曾經用來於此種電動機驅動電路 1229486 濾波共模電流。被動式共模濾波器對可使用之PWM頻率上 有限制、實體尺寸大(經常占電動機驅動器結構之一大部分) 且叩貝。此外功能不完美，被動式驅動器於對所欲濾波作 用執行a十數器時，出現非期望之諧振。此外於一般用途之 5工業驅動器，驅動器與電動機間經常係利用長達1〇〇米或以 上的纜線連結。纜線愈長，則電動機缆線傳導之共模eMI 愈大，則要求之習知被動式共模輸入濾波器之尺寸也愈大。 有額外繞組藉電阻器短路之一種共模變壓器為已知， 該共模變壓器可阻尼振盪地電流。不幸仍有小量非週期地 10 電流仍然留在此種電路。 用於脈波寬度經調變(PW]V[)經控制之電動機驅動器電 路’用來控制共模電流之主動式濾波器為眾所周知。此種 主動式濾波器典型係述於報告：用於抵消藉PWM反相器產 生之共模電壓之主動電路，作者g；at〇shi Ogasawara等人， 15 IEES電源電子裝置會刊第13卷第5期（1998年9月）；以及 Ogasawara等人之專利案5,831，842。 第10圖顯示用於交流電動機之典型先前技術主動式濾 波電路或EMI及雜訊消除氣。如此，包含輸入端子L以及中 性端子之交聯電源連結至全波橋式連結整流器40之交流輸 2〇 入端子。雖然顯示單相電源供應，但本圖以及隨後將說明 之各圖之原理可以三相輸入或多相輸入進行。整流器4〇之 正及負匯流排分別含有點A及點D，且於反相器端子B及F 連結至三相橋式連結之PWM經控制之反相器41。反相器之 輸出交流端子連結至交流電動機42。濾波電容器40a也跨端 1229486 子B及端子F連結。電動機42有一接地殼體以接地端子43& 而連結至地線43。 由一對電晶體(^及仏組成之主動濾波器跨整流器4〇之 直流輸出線連結，而其射極係連結於節點E。如此界定放大 5器，該放大器係由差異變壓器之輸出繞組44控制，該差異 變壓器之輸入繞組45及46分別係連結於整流器4〇之正及負 輸出匯流排。繞組極性係以習知點符號指定。繞組44係連 結於電晶體Qi及Q2之控制端子與共通射極節點E間。直流隔 離電容器47係連結至地線43於節點c。 10 主動式濾波器包括電容器47界定一路徑用以分流大部 分之共模電流，否則共模電流將於路徑L或N、A、B、Μ(電 動機42)、43、43a及返回L或Ν之路徑流動；（或當極性顛倒 時於相反路徑）或於路徑L或N、D、F、Μ、43、43a流動（或 當極性顛倒時於相反路徑流動）。如此大部分共模電流可被 15 分流’對於來自正端子A之電流’對「正」電流於路徑B、 Μ、C、E、Q2、F、B分流；以及對「負」電流以b、μ、C、 Ε、Qi、Β之樣式分流，此種分流可經由適當控制電晶體& 及Q2而達成。流入負端子D之共模電流路徑對「正」電流遵 循路徑F、M、C、E、Q2、F，對「負」電流遵循路徑F、M、 20 C、E、Qi、B。分流程度對「正」電流而言係依據繞組44 之電流增益及Q2電流增益決定；以及對「負」電流而言分 流程度係依據繞組44及電流増益及(^之電流增益決定。為 了獲得共模電流之足夠分流程度，繞組44及電晶體(^及仏 之總電流增益需為高增益。 1229486 第10圖之感應變壓器44、45、46為了提供夠高的電流 增盈需變大且變昂貴。極為希望縮小變壓器尺寸之成本， 而未危及電路的操作。又另一項問題為因需要高增益，故 此種閉環電路傾向於產生非期望的振盈。 5 此外，發現電晶體Ql及Q2於電路所界限的「淨空高度」 以内無法於線性區域夠大範圍操作，如此不利於主動濾波 作用，淨空高度或電晶體(^及^之集極與射極間之電壓經 由考慮第10圖之近似等效電路將可最為明瞭，如第丨丨圖所 示，其中於C之地電位係等於第10圖中性線路之地電位。電 10晶體Ql及Q2顯不為電組器Ri及汉2，有個別並聯連結之二極 體。直流橋40顯示為二直流電源5〇及51，各自產生輸出電 壓VDC/2，此處VDC為於端子a及端子D之正匯流排與負匯流 排間之全輸出電壓，交流電源52具有尖峰交流電壓Vdc/2。 由第2圖可知於電源週期之不同部分，淨空高度可能消 15失。如此考慮第-種情況，其中電晶體^及仏之茂漏阻抗 為相等。此種情況下，第2圖電阻器心及化之值約略相等。 現在，當端子C之地電位相對於第2圖節點兄之直流中點， 介於(+)VDC/2與㈠VDC/2間擺盪時，若假設電容器47之阻抗 係選小於1及112，則於電晶體(^及仏涉及電位也介於 20 (+)VDC/2與㈠VDC/2間擺盪。因此於節點E之電位係接近於或 等於直流匯流排(於節點B或節點F)之電位之期間，對相關 電晶體QAQ2存在的電壓淨空高度不足以呈線性放大器操 作，故主動濾波作用喪失。 EMI渡波器於多種電磁應用用途特別於輸電系統為眾 1229486 所周知。涉及電力傳輸之系統典型包括電源反相器，其除 了用於電動機驅動器之外，也可用於電源供應器用途了^ 源反相器典型被供給電力通過輸電線，輪電線係以多相模 式操作。例如三相電源供應器典型係用於涉及反相器操作 5以及電動機驅動器之應用用途。三相電源供應器包括三條 輸電線，具有介於三對輸電線間之電壓電位。換言之，若 三相輸入係經由線Ll、L2及L3供應，則介於線11與1^間、 線L2與L3間以及線“與!^間有電壓電位。此等相至相電壓 典型為正弦波形，相對於彼此非為合規相位來提供有效電 10 力傳輸。 於類似前文說明之三相系統，輸電線係作為傳輸電源 信號之差異電壓對，該電源信號具有各成對線路間之電壓 值。此型輸電架構用以傳輸電源信號極為有用，對於同時 影響全部電源線之雜訊斷續具有免疫力。換言之，若全部 15電源線受到共同干擾或雜訊信號影響，則全部線路被影響 至相等程度’差異電壓維持相等。如此常見三相傳輸絲 有共模電壓，該共模電壓並非必然影響傳輸至反相器之電 源信號(舉例）。 當反1 目器用於供電且控制電動機驅動系統時，反相器 典型使用高頻切換，來導引適當電源信號之電動機繞組來 產生期望之操作效能。例如反相器可操作而控制電動機用 於載明之扭矩操作或預定速度操作。由於反㈣之高頻切 換’經常於電動機驅動線上有突然電壓變遷此乃讓之 特有來源。此種EMI產生共模雜訊，共模雜訊於電動機控 20 1229486 制信號、回授信號I/O、感應器等造成干擾。此外，與反相 器輪出端及接地端之電容耦合、或電動機接地之本身可產 生高頻地電流，提供與控制信號以及其它通訊信號之進_ 步干擾。高頻地電流也導致輻射干擾，產生接地回路，其 5作為環形天線而增加輻射雜訊的製造。高頻地電流也導致 兩個地電位點間之瞬間電壓差，而干擾控制信號及通訊信 號的適當參照。 已知多種措施可減少及控制共模雜訊及輻射EMI。例 如、、二過屏蔽之電源線用於將反相器連結至電動機來防止雜 10汛流流出電動機驅動系統之接地。電動機的電源線也扭絞 來提供平衡電容耦合，俾降低耦合接地之雜散電容。共模 抗流線圈用於電動機之電源線，也用來衰減共模雜訊。EMI 慮波器常附接至反相器之輸入，用作為低通濾波器去除來 自地電位之共模雜訊，否則該共模雜訊可能對電動機驅動 15系統之一或多個組成元件形成地電壓差。 另一項降低EMI雜訊之技術，係測量高頻雜訊電流， 對任何偵測得之電流提供補償。電流變壓器曾經用來感應 雜訊電流，俾測定控制EMI之適當補償。但具有適當尺寸 及額定值之電流變壓器體積龐大且價格昂貴，實際上產生 2〇非線性操作。希望提供一種未使用電流變壓器而降低EMI 之電路及技術。 【發明内容】 發明概要 本發明提供一種主動EMI濾波器，其未使用電流變壓 1229486 器而提供絕佳EMI降低特性。根據本發明之電路及方法感 應電壓，該電壓代表地線電流，且比較反相器輸入線之共 模電壓獲得差異信號。差異信號經放大，用於閉環控制作 為誤差信號，來將共模電壓與感應得之地電壓間之差異降 5 至零。如此本發明經由測量地電流感應之電壓然後補償地 電流，而測定高頻地電流。電路可用於單相電源系統或多 相電源系統，提供優於習知用於降低EMI之電流變壓器系 統或線性電壓調節器之改良效率。 根據本發明，提供一種共模電壓輸入至一誤差放大 10 器。一地電壓輸入信號也供給誤差放大器，來基於二信號 而獲得經放大之誤差信號。誤差放大器之輸出信號供給地 線，來補償高頻地電流，避免高頻地電流於電動機驅動系 統傳播。 誤差放大器係於輸入電源信號之上限與下限間相關範 15 圍操作，來自動調整誤差放大器之操作至輸入電源線之共 模電壓。經由操作MOSFET作為經過控制的電阻來提供恆 定電壓輸出，而與抵消的雜訊量無關，獲得高的開環增益。 經過控制之電阻可降低高頻電流變遷，俾進一步降低電路 產生的EMI。 20 圖式簡單說明 將參照附圖說明本發明如後，附圖中： 第1圖為電路圖顯示本發明； 第2圖為第1圖電路之摘要等效電路；1229486 发明 Description of the invention: The related application is based on US Provisional Patent Application No. 60 / 391,987, filed on June 25, 2002, under the name "Active EMI 5 Filter without Current Sensing Device" and US Provisional Patent Application No. 60 / 408,534, filed on September 5, 2002, named "Active EMI Transponder" and applied for the rights of these patent applications, and claimed priority in the two cases mentioned above. [Technical field to which the invention belongs 3 Field of invention 10 The present invention relates to the reduction of electromagnetic interference (EMI) of a power conversion system ', and particularly relates to the reduction of the EMI of a motor drive system. [Background of the Invention] High-speed switching devices such as bipolar transistors, MOSFETs, and IGBTs, s 15 can provide the carrier frequency of the voltage source PWM inverter, resulting in far better operating characteristics. However, high-speed switching causes the following serious problems, which are caused by high-speed changes in voltage and / or current: a) the ground current escapes to the ground through stray capacitors inside the motor and through long cables; 20 b) conducted and radiated EMI c) the motor carries current and axial voltage; and d) the insulation life of the motor and transformer is shortened. Because parasitic stray capacitance inevitably exists inside the load, such as inside an AC motor and inside a switching converter, when the switching device changes state 1229486, the voltage and / or current caused by high-speed switching changes, resulting in high-frequency vibration common mode And #RIGHT Current. In this way, each time an inverter switching event occurs, the potential of the corresponding output terminal of the inverter moves quickly relative to the ground potential. The common mode current pulse flows through the heat sink motor cable and the motor winding to ground and flows to the 5 inverter on the DC link. . The current pulse amplitude for this type B (residential) motor driver is typically several hundred milliamperes to several amperes; the pulse width is typically 25 to 500 nanoseconds. For Class A drives (industrial use), and depending on the size of the motor and the length of the motor cable, the pulse current amplitude is used for the pulse width from 25 nanoseconds to 500 nanoseconds, typically several amps, up to the pulse width丨 to 20 amps at or above 20 amps in 2 microseconds. The common mode oscillation current has a frequency range from the converter switching frequency to tens of MHZ. The oscillation current forms a magnetic field and will generate radiated electromagnetic interference (EMi) everywhere, which will adversely affect electronic devices such as wireless receivers and medical equipment. . 15 The government has many regulations that limit the allowable line current EMI level and allowable ground current for certain motor applications. Thus, for Class B residential (appliances) applications, the ground current must be maintained below 1 to 20 milliamps in the frequency range of 0 to 30 kHz (on a logarithmic curve); and the conductor current EMI must be maintained in the frequency range of 150 kHz to 300 MHz. Below the specified value (less than about 60 (ΙΒμν). For motor drive applications designated as Class 20 A industrial applications, the ground current limit is stricter, but the line current EMI is still limited in the range of 150 kHz to 30 MHz. Usually based on passive Components, common mode choke coils and EMI filters cannot completely solve these problems. Passive filters composed of common mode inductors and "Y" capacitors on the input AC line were used to filter common mode in this motor drive circuit 1229486 Current. Passive common-mode filters have limited PWM frequencies that can be used, have a large physical size (often occupying a large part of the structure of a motor driver), and have a small shell. In addition, the function is not perfect. Unexpected resonance occurs when the number is ten. In addition, for general-purpose 5 industrial drives, the drive and motor are often used up to 1〇. M or more cable connection. The longer the cable, the larger the common mode eMI conducted by the motor cable, and the larger the size of the passive common mode input filter required. The additional windings are shorted by the resistor. A common-mode transformer is known. This common-mode transformer can damp the oscillating ground current. Unfortunately, a small amount of non-periodic 10 current is still left in this circuit. It is used for pulse width modulated (PW) V [) Controlled motor driver circuits' active filters used to control common-mode current are well known. Such active filters are typically described in the report: Active Circuits Used to Cancel Common-Mode Voltage Generated by PWM Inverters, Author g; at〇shi Ogasawara et al., 15 IEES Journal of Power Electronics, Vol. 13 No. 5 (September 1998); and Ogasawara et al. Patent No. 5,831,842. Figure 10 shows a typical AC motor The prior art active filter circuit or EMI and noise cancellation gas. In this way, the cross-linked power supply including the input terminal L and the neutral terminal is connected to the AC input 20 input terminal of the full-wave bridge connection rectifier 40. Although shown Single-phase power supply, but the principle of this diagram and the diagrams to be described later can be performed with three-phase input or multi-phase input. The positive and negative busbars of the rectifier 40 contain points A and D, respectively, and are at the inverter terminals. B and F are connected to a three-phase bridge-connected PWM controlled inverter 41. The output AC terminal of the inverter is connected to the AC motor 42. The filter capacitor 40a is also connected across terminal 1229486 Sub B and terminal F. The motor 42 has a The grounding case is connected to the ground line 43 by a ground terminal 43. An active filter composed of a pair of transistors (^ and 仏) is connected across the DC output line of the rectifier 40, and its emitter is connected to the node E. The amplifier 5 is defined as such, the amplifier is controlled by the output winding 44 of the differential transformer, and the input windings 45 and 46 of the differential transformer are connected to the positive and negative output buses of the rectifier 40, respectively. The winding polarity is specified with the conventional dot symbol. The winding 44 is connected between the control terminals of the transistors Qi and Q2 and the common emitter node E. The DC isolation capacitor 47 is connected to the ground 43 at the node c. 10 The active filter includes a capacitor 47 to define a path to shunt most of the common mode current, otherwise the common mode current will be in the path L or N, A, B, M (motor 42), 43, 43a, and return L or N Flow on the path; (or on the opposite path when the polarity is reversed) or flow on the path L or N, D, F, M, 43, 43a (or on the opposite path when the polarity is reversed). In this way, most common-mode currents can be shunted by 15 'for the current from the positive terminal A' to "positive" currents in paths B, M, C, E, Q2, F, B; and "negative" currents by b, μ, C, Ε, Qi, B pattern shunting, this shunting can be achieved by appropriately controlling the transistor & and Q2. The common-mode current path flowing into the negative terminal D follows the paths F, M, C, E, Q2, and F for the “positive” current, and the paths F, M, 20 C, E, Qi, and B for the “negative” current. For the “positive” current, the degree of shunting is determined by the current gain of winding 44 and the Q2 current gain; and for the “negative” current, the degree of shunting is determined by the winding 44 and the current gain and the current gain of ^. Sufficient shunting of the modal current, the total current gain of the winding 44 and the transistor (^ and 仏 must be high gain. 1229486 Induction transformers 44, 45, 46 in Figure 10 need to be large and variable in order to provide sufficient current Expensive. It is extremely desirable to reduce the size of the transformer without compromising the operation of the circuit. Another problem is that such closed-loop circuits tend to produce undesired vibration due to the need for high gain. 5 In addition, transistors Ql and Q2 were found Within the "headroom height" bounded by the circuit, it is not possible to operate in a large range in the linear region. This is not conducive to active filtering. The headroom or the voltage between the collector and emitter of the transistor (^ and ^ is considered by considering Figure 10). The approximate equivalent circuit will be the most clear, as shown in Figure 丨 丨, where the ground potential at C is equal to the ground potential of the neutral line in Figure 10. Electric crystals Q1 and Q2 are not electrical Devices Ri and Han 2 have individual diodes connected in parallel. The DC bridge 40 is shown as two DC power sources 50 and 51, each generating an output voltage VDC / 2, where VDC is the positive bus at terminals a and D. For the full output voltage from the negative bus, the AC power supply 52 has a peak AC voltage Vdc / 2. As can be seen from Figure 2, the headroom may be reduced by 15 in different parts of the power cycle. So consider the first case where the transistor ^ And 仏 's leakage resistances are equal. In this case, the resistor values in Figure 2 are approximately equal. Now, when the ground potential of terminal C is relative to the DC midpoint of the node in Figure 2, When swinging between (+) VDC / 2 and ㈠VDC / 2, if it is assumed that the impedance of capacitor 47 is less than 1 and 112, then the potential of the transistor (^ and 仏 is also between 20 (+) VDC / 2 and ㈠VDC / 2 swings. Therefore, during the period when the potential of node E is close to or equal to the potential of the DC bus (at node B or node F), the voltage headroom for the related transistor QAQ2 is not high enough to operate as a linear amplifier. The filtering effect is lost. Unlike power transmission systems, which are well known by 1229486. Power transmission systems typically include power inverters, which can be used for power supplies in addition to motor drives. Source inverters are typically supplied with power through power transmission. Electric wires and wheels are operated in multi-phase mode. For example, three-phase power supplies are typically used in applications involving inverter operation5 and motor drives. Three-phase power supplies include three power lines with three The voltage potential between the wires. In other words, if the three-phase input is supplied via the lines L1, L2, and L3, there is a voltage potential between the lines 11 and 1 ^, the lines L2 and L3, and the lines "and! ^. The phase-to-phase voltages are typically sinusoidal waveforms that provide non-compliant phases relative to each other to provide effective electrical power transmission. In a three-phase system similar to that described above, the power transmission line is used as a differential voltage pair for transmitting a power signal, and the power signal has a voltage value between the paired lines. This type of transmission architecture is extremely useful for transmitting power signals and is immune to the intermittent noise that affects all power lines at the same time. In other words, if all 15 power lines are affected by common interference or noise signals, all lines are affected to the same degree 'and the difference voltages remain the same. Such a common three-phase transmission wire has a common-mode voltage, and the common-mode voltage does not necessarily affect the power signal transmitted to the inverter (for example). When an inverter is used to power and control the motor drive system, the inverter typically uses high frequency switching to steer the motor windings of the appropriate power signal to produce the desired operating efficiency. For example, the inverter can be operated to control the motor for specified torque operation or predetermined speed operation. This is a unique source because of the unexpected high-frequency switching, which often has sudden voltage changes on the motor drive line. This kind of EMI generates common mode noise, which causes interference in motor control signals, feedback signals I / O, and sensors. In addition, the capacitive coupling with the inverter wheel output and ground terminals, or the motor ground itself, can generate high-frequency ground current, providing further interference with control signals and other communication signals. High-frequency ground currents also cause radiated interference and generate ground loops, which act as loop antennas to increase the production of radiated noise. The high-frequency ground current also causes an instantaneous voltage difference between the two ground potential points, and interferes with the proper reference of control signals and communication signals. Various measures are known to reduce and control common mode noise and radiated EMI. For example, the two-shielded power lines are used to connect the inverter to the motor to prevent miscellaneous currents from flowing out of the motor drive system ground. The power cord of the motor is also twisted to provide balanced capacitive coupling and reduce stray capacitance from coupling to ground. The common mode anti-winding coil is used for the power line of the motor and also used to attenuate common mode noise. EMI filter is often attached to the input of the inverter, used as a low-pass filter to remove common mode noise from ground potential, otherwise the common mode noise may form one or more components of the motor drive 15 system Ground voltage difference. Another technology to reduce EMI noise is to measure high-frequency noise current and provide compensation for any detected current. Current transformers have been used to induce noise currents and determine the proper compensation for controlling EMI. However, current transformers with the appropriate size and rating are bulky and expensive, which actually produces 20 nonlinear operations. It is desirable to provide a circuit and technology that reduces EMI without using a current transformer. SUMMARY OF THE INVENTION The present invention provides an active EMI filter, which does not use a current transformer 1229486 and provides excellent EMI reduction characteristics. The circuit and method according to the present invention senses a voltage, which represents the ground current, and compares the common-mode voltage of the input lines of the inverter to obtain a difference signal. The difference signal is amplified and used for closed-loop control as an error signal to reduce the difference between the common-mode voltage and the induced ground voltage by 5 to zero. In this way, the present invention measures the high-frequency ground current by measuring the voltage induced by the ground current and then compensating the ground current. The circuit can be used in single-phase power systems or multi-phase power systems, providing improved efficiency over current transformer systems or linear voltage regulators conventionally used to reduce EMI. According to the present invention, a common-mode voltage input is provided to an error amplifier. A ground voltage input signal is also supplied to the error amplifier to obtain an amplified error signal based on the two signals. The output signal of the error amplifier is supplied to the ground wire to compensate the high-frequency ground current and prevent the high-frequency ground current from propagating in the motor drive system. The error amplifier operates within the range of the upper and lower limits of the input power signal to automatically adjust the operation of the error amplifier to the common-mode voltage of the input power line. A constant voltage output is provided by operating the MOSFET as a controlled resistor, regardless of the amount of noise canceled, and a high open-loop gain is obtained. The controlled resistance can reduce the high-frequency current transition and further reduce the EMI generated by the circuit. 20 Brief Description of the Drawings The present invention will be described later with reference to the drawings. In the drawings: FIG. 1 is a circuit diagram showing the present invention; FIG. 2 is a summary equivalent circuit of the circuit of FIG. 1;
第3圖為電路圖顯示根據本發明之複數個主動式Ε ΜI 12 1229486 濾波器之使用； 第4圖為本發明之摘要等效電路，說明雜訊之消除； 第5圖為全部摘要電路圖，顯示根據本發明之共模雜訊 之消除； 5 第6圖為電路圖，顯示根據本發明之三相共模主動式 EMI濾波器； 第7圖為電路圖，顯示根據本發明之三相輸入端之共模 及差異模主動式EMI濾波； 第8圖為線圖顯示根據本發明，主動式EMi滤波器之電 10 壓降調節器； 第9圖為線圖顯示根據本發明，電壓調節器之高頻雜訊 之降低； 第10圖為已知之主動式EMI濾波器之電路圖；以及 第11圖為第10圖所示等效電路之電路圖。 15 【實施方式】 較佳實施例之詳細說明 現在參照第1圖，顯示主動式EMI濾波電路100。來自 電動機機殼之共模電流係經由R1及C1之電壓而感應，共模 電流供給誤差放大器102之反相輸入端。誤差放大器102之 20 增益係由R1及C1與R2之組合決定。 電源線LI、L2及L3之共模電壓係透過電容器C5-C7感 應，供給差異變壓器102之非反相輸入端。若電容器C5-C7 感應之共模電壓相對於透過電阻器R1及電容器C1對參考 地電位測得之電壓略微正，則誤差放大器102輸出，且升高 13 1229486 電壓，俾驅使二電壓間之差異為零。此種情況之一項結果 為誤差放大器102之電壓執下降，輸入線之共模電壓透過電 容器C3-C7而降低。 此種電路之操作傾向於提供放大器1〇2之非反相輸入 5 端與反相輸入端間之理論短路。如此電路需避免地電位與 輸入線間之共模電壓。 轉向參照第2圖，共模等效電路概略顯示為電路2〇。等 效電路20包括輸入線阻抗21及誤差放大器22。如前文討 論，疾差放大器22插作而獲得反相輸入端與非反相輸入端 10間之理論短路。電壓VN及阻抗ZM表示輸入線驅動之電動機 系統之反相器。共模電流以電壓源Vc及阻抗Zc補償。由於 放大器22之二輸入端子間之虛擬短路，故與電壓共 通節點之電流和以如下方程式表示。 15 如方程式（1)可知，電壓Vc之調變允許方程式（1)平衡， 故電流抵消。如此共模消除電路可防止共模電流於地線傳 播。 現在參照第3圖，電路30係以主動式EMI濾波器31、32 濾波共模雜訊及差異模雜訊做說明。操作中，濾波器31、 20 32係於放大器33、34之非反相輸入點附接於電源線。放大 器33、34感應輸入線上之共模電壓，驅動器個別輸出端， 而產生相對於地電位之平衡電壓。此種組態及操作可消除 路由至輸入線之共模電流。此外，因濾波器31及32於各輪 1229486 入線上係獨立操作，故差異輸入線間 波差異模式EMI雜訊。 之差異抵消 ，藉此滤 差異圖’等效電路4〇用來說明第3圖之共模及 電壓42 d 5。提供共模雜賴模式，包括阻抗41及 電壓源42。雜輯有城43 =地電位與輪入線間之任何差異，且將一二入 輸入線來平衡共模雜訊。 現在參照第5圖，主動式刪遽波器50經模式化，顯示 主入具有線阻抗51之輸入線之共模雜訊之消除。放大找 1〇產生補償電壓VG結合補償隐G，來平《壓源VN結合阻 抗ZMt、、、，。的共換雜机電流。如此，主動式麵遽波電路別 經由提供-補償電流，其匹配雜訊源提供之電流，而防止 共模電流雜訊被注入傳輸線。 現在參照第6圖，舉例說明具有三相整流器61之三相共 模渡波器60。整流器61之輸出端供給操作電壓執至放大器 62 ’放大器62不具有回授路徑，只使用電容器63、64操作 來提供電壓信號平衡輸入線U_L3之共模電壓。如此，於電 谷器63感應地線電流，對來自輸入線u_L32共模電壓提供 的參考電流做比較。獲得跨電容器64之放大器62之輸出電 20壓，其傾向於提供平衡電壓，來批配線L1-L3感應之共模電 壓。 現在參照第7圖，顯示一種三相共模及差異模濾波器 70。渡波器70之組態為第3圖及第6圖所示濾波器的組合。 例如渡波器71提供輸入線L3之共模及差異模濾波，濾波器 15 1229486 72及73分別對輸入線L2及L1提供相同濾波。共模電壓由輸 入線L2供給放大器74之非反相輸入端，輸入線L2及L3對放 大器74供給電壓執電源。電容器C3及C4用來發展電壓，其 回授放大器74之反相輸入端，俾平衡輸入線L2感應之電 5 壓。經由補償於輸入線L2感應之共模電壓，濾波器71提供 相對於輸入線L3之共模濾波。因EMI濾波器71-73各自參照 地電位，具有相同主動式組態，故除了輸入線L1-L3各自之 共模雜訊之外，輸入線L1-L3間之差異模雜訊也被濾波。 現在參照第8圖，顯示根據本發明之共模EMI濾波器之 10電壓調節器電路80。電路80係類似第3圖所示，例外為供給 放大器81、82之執電壓係透過電壓調節器電路83導出，而 非直接由輸入線導出。第3圖顯示之EMI濾波器31、32係依 據咼阻抗電源供應器相對於輸入線之阻抗決定，獲得適當 最大開環增益。如此MOSFET Q20係呈恆定電阻源操作， 15來產生經過調節之直流電源供應給誤差放大器81、82。控 制電晶體Q21被驅動而對Q2〇維持特定電阻。結果為調節電 壓供給誤差放大器81、82。包括增納二極體D9及電晶體Q22 之電路提供恆定電壓回授，而與抵消之雜訊量無關。如此 某種量之電阻提供於跨電容器C13所得電源供應器與設置 20於一極體D7陽極之輸入線L間。維持此種電阻對於於主動 EMI濾波器80進行差異模雜訊消除相當重要。若未於輸入 線與C13之電源供應器間維持電阻，則誤差放大器喪失其回 路增益，所得差異模雜訊衰減不良。 電路80也提供一種特色，降低電壓調節器電路83產生 16 1229486 的差異模雜訊。換言之，供給電容器C13之充電電流具有高 di/dt波尖，該波尖係來自於電壓調節器與充電電容器C13 間之放電。電阻控制之回授包括電阻量，俾減小於二極體 D7觀察得之di/dt。 5 第9圖顯示於無電阻控制下，以及有電阻控制回授，於 二極體D7觀察得之電流充電波形。如此電晶體Q21及電阻 器R24-R27提供閉環電阻控制，進一步減小電流波尖，來避 免於電路80之主動式EMI濾波器導入額外差異模雜訊。 雖然已經就特定具體實施例說明本發明，但多種其它 10 變化及修改以及其它用途對熟諳技藝人士顯然易知。因此 較佳本發明並非受此處特定揭示所限，而僅由隨附之申請 專利範圍界定。 【圖式簡單說明】 第1圖為電路圖顯示本發明； 15 第2圖為第1圖電路之摘要等效電路； 第3圖為電路圖顯示根據本發明之複數個主動式EMI 濾波器之使用； 第4圖為本發明之摘要等效電路，說明雜訊之消除； 第5圖為全部摘要電路圖，顯示根據本發明之共模雜訊 20 之消除； 第6圖為電路圖，顯示根據本發明之三相共模主動式 EMI濾波器； 第7圖為電路圖，顯示根據本發明之三相輸入端之共模 及差異模主動式EMI濾波； 17 1229486 第8圖為線圖顯示根據本發明，主動式EMI濾波器之電 壓降調節器； 第9圖為線圖顯示根據本發明，電壓調節器之高頻雜訊 之降低； 5 第10圖為已知之主動式EMI濾波器之電路圖；以及 第11圖為第10圖所示等效電路之電路圖。 【圖式之主要元件代表符號表】 20，40…等效電路 44，45，46…感應變壓器 21…輸入線阻抗 45，46···輸入繞組 22，102…誤差放大器 47…直流隔離電容器 30…電路 50，51…直流電源 31，32，50…EMI濾波器 52…交流電源 33，34，44，52，74，81，82··· 53…節點 放大器 60···三相共模濾波器 40…全波橋式連結整流器 61…三相整流器 40a···濾波電容器 63，64…電容器 41，43，51…阻抗 70-73…濾波器 41…反相器 80，83…電壓調節電路 42…電壓源 81，82…放大器 42…交流電動機 100···主動式電磁干擾濾波電路 43…地線 C…電容器 43a···接地端子 L…線 44…輸出繞組 R…電阻器 18Fig. 3 is a circuit diagram showing the use of a plurality of active EMI 121229486 filters according to the present invention; Fig. 4 is an abstract equivalent circuit of the present invention, illustrating the elimination of noise; Fig. 5 is a full summary circuit diagram, showing Elimination of common mode noise according to the present invention; 5 FIG. 6 is a circuit diagram showing a three-phase common-mode active EMI filter according to the present invention; FIG. 7 is a circuit diagram showing a common three-phase input terminal according to the present invention. Mode and differential mode active EMI filtering; FIG. 8 is a line diagram showing the electric 10 voltage drop regulator of the active EMi filter according to the present invention; FIG. 9 is a line diagram showing the high frequency of the voltage regulator according to the present invention Noise reduction; Figure 10 is a circuit diagram of a known active EMI filter; and Figure 11 is a circuit diagram of an equivalent circuit shown in Figure 10. [Embodiment] Detailed description of the preferred embodiment Now referring to FIG. 1, an active EMI filter circuit 100 is shown. The common mode current from the motor case is induced through the voltages of R1 and C1, and the common mode current is supplied to the inverting input terminal of the error amplifier 102. The 20 gain of the error amplifier 102 is determined by R1 and the combination of C1 and R2. The common-mode voltages of the power lines LI, L2, and L3 are sensed through the capacitors C5-C7 and supplied to the non-inverting input terminal of the differential transformer 102. If the common-mode voltage induced by capacitors C5-C7 is slightly positive relative to the voltage measured through the resistor R1 and capacitor C1 to the reference ground potential, the error amplifier 102 will output and increase the voltage by 13 1229486, which will drive the difference between the two voltages. Is zero. One result of this situation is that the voltage of the error amplifier 102 decreases, and the common-mode voltage of the input lines decreases through the capacitors C3-C7. The operation of such a circuit tends to provide a theoretical short circuit between the non-inverting input 5 of the amplifier 102 and the inverting input. Such a circuit needs to avoid the common mode voltage between the ground potential and the input line. Turning to FIG. 2, the common-mode equivalent circuit is schematically shown as circuit 20. The equivalent circuit 20 includes an input line impedance 21 and an error amplifier 22. As discussed earlier, the difference amplifier 22 is inserted to obtain a theoretical short circuit between the inverting input terminal and the non-inverting input terminal 10. The voltage VN and impedance ZM represent the inverters of the motor system driven by the input lines. The common mode current is compensated by the voltage source Vc and the impedance Zc. Due to the virtual short circuit between the two input terminals of the amplifier 22, the sum of the currents at the nodes in common with the voltage is expressed by the following equation. 15 As can be seen from equation (1), the modulation of voltage Vc allows equation (1) to balance, so the current cancels. This common-mode cancellation circuit prevents common-mode currents from propagating on the ground. Referring now to FIG. 3, the circuit 30 is described using active EMI filters 31 and 32 to filter common mode noise and differential mode noise. In operation, the non-inverting input points of the filters 31, 20, 32 to the amplifiers 33, 34 are attached to the power line. The amplifiers 33 and 34 sense the common-mode voltage on the input line and the individual output terminals of the driver to generate a balanced voltage with respect to the ground potential. This configuration and operation eliminates common-mode currents routed to the input lines. In addition, since the filters 31 and 32 operate independently on the 1229486 input lines of each round, the differential input line wave difference mode EMI noise. The difference cancels out, thereby filtering the difference diagram 'equivalent circuit 40 to illustrate the common mode and voltage 42 d 5 of FIG. 3. Common mode hybrid mode is provided, including impedance 41 and voltage source 42. Noise 43 = Any difference between the ground potential and the turn-in line, and one or two input lines are used to balance common mode noise. Referring now to FIG. 5, the active erasing wave filter 50 is patterned to show the elimination of common mode noise that is mainly input to the input line with line impedance 51. Zoom in and find 10 to generate the compensation voltage VG combined with the compensation hidden G to level the voltage source VN combined with the impedances ZMt, ..., and. Common commuter current. In this way, the active surface wave circuit prevents the common-mode current noise from being injected into the transmission line by providing-compensating current, which matches the current provided by the noise source. Referring now to Fig. 6, a three-phase common-mode ferrule 60 having a three-phase rectifier 61 will be described by way of example. The output terminal of the rectifier 61 supplies the operating voltage to the amplifier 62. The amplifier 62 does not have a feedback path, and only operates with capacitors 63 and 64 to provide the common-mode voltage of the voltage signal balanced input line U_L3. In this way, the earth current is induced in the valley 63 and the reference current provided by the common-mode voltage of the input line u_L32 is compared. An output voltage of the amplifier 62 across the capacitor 64 is obtained, which tends to provide a balanced voltage to the common mode voltage induced by the wiring L1-L3. Referring now to FIG. 7, a three-phase common-mode and differential-mode filter 70 is shown. The configuration of the waver 70 is a combination of the filters shown in FIG. 3 and FIG. 6. For example, the wave filter 71 provides common mode and differential mode filtering of the input line L3, and the filters 15 1229486 72 and 73 provide the same filtering to the input lines L2 and L1, respectively. The common mode voltage is supplied to the non-inverting input terminal of the amplifier 74 through the input line L2, and the input lines L2 and L3 supply the voltage-supply power to the amplifier 74. Capacitors C3 and C4 are used to develop the voltage. The feedback input terminal of the amplifier 74 and the balanced input line L2 induce the voltage. By compensating the common mode voltage induced on the input line L2, the filter 71 provides common mode filtering with respect to the input line L3. Since the EMI filters 71-73 each refer to ground potential and have the same active configuration, in addition to the common mode noise of the input lines L1-L3, the differential mode noise between the input lines L1-L3 is also filtered. Referring now to Fig. 8, there is shown a voltage regulator circuit 80 of a common mode EMI filter according to the present invention. The circuit 80 is similar to that shown in FIG. 3, except that the voltages supplied to the amplifiers 81 and 82 are derived through the voltage regulator circuit 83 instead of being directly derived from the input lines. The EMI filters 31 and 32 shown in Figure 3 are determined according to the impedance of the 咼 impedance power supply with respect to the input line to obtain an appropriate maximum open loop gain. In this way, the MOSFET Q20 operates as a constant resistance source, and 15 to generate a regulated DC power supply to the error amplifiers 81 and 82. The control transistor Q21 is driven to maintain a specific resistance to Q20. As a result, the adjustment voltage is supplied to the error amplifiers 81 and 82. The circuit including the sonar diode D9 and transistor Q22 provides constant voltage feedback, regardless of the amount of noise canceled. Such a certain amount of resistance is provided between the power supply obtained from the capacitor C13 and the input line L provided at the anode of a pole D7. Maintaining such resistance is important for performing differential mode noise cancellation in the active EMI filter 80. If the resistance is not maintained between the input line and the power supply of C13, the error amplifier loses its circuit gain and the resulting differential mode noise attenuation is poor. The circuit 80 also provides a feature that reduces the voltage regulator circuit 83 to produce 16 1229486 differential mode noise. In other words, the charging current supplied to the capacitor C13 has a high di / dt peak, which is caused by the discharge between the voltage regulator and the charging capacitor C13. The feedback of resistance control includes the amount of resistance, which is reduced to the di / dt observed by diode D7. 5 Figure 9 shows the current charging waveform observed in diode D7 under the control of no resistance and feedback with resistance control. In this way, the transistor Q21 and the resistors R24-R27 provide closed-loop resistance control to further reduce the current wave tip to avoid the introduction of additional differential mode noise in the active EMI filter of circuit 80. Although the invention has been described in terms of specific embodiments, many other variations and modifications and other uses will be apparent to those skilled in the art. Therefore, the preferred invention is not limited by the specific disclosure herein, but only by the scope of the appended patents. [Schematic description] Figure 1 is a circuit diagram showing the invention; 15 Figure 2 is a summary equivalent circuit of the circuit of Figure 1; Figure 3 is a circuit diagram showing the use of a plurality of active EMI filters according to the invention; Fig. 4 is the abstract equivalent circuit of the present invention, explaining the elimination of noise; Fig. 5 is the entire abstract circuit diagram, showing the elimination of the common mode noise 20 according to the present invention; Fig. 6 is a circuit diagram, showing the elimination of noise according to the present invention Three-phase common-mode active EMI filter; Figure 7 is a circuit diagram showing the common-mode and differential-mode active EMI filtering of the three-phase input terminal according to the invention; 17 1229486 Figure 8 is a line diagram showing the active-mode filter according to the invention. Voltage drop regulator of the EMI filter; Figure 9 is a line diagram showing the reduction of high frequency noise of the voltage regulator according to the present invention; 5 Figure 10 is a circuit diagram of a known active EMI filter; and Figure 11 The figure is a circuit diagram of the equivalent circuit shown in FIG. [Representative symbol table of main components of the figure] 20, 40 ... Equivalent circuits 44, 45, 46 ... Induction transformer 21 ... Input line impedance 45, 46 ... Input winding 22, 102 ... Error amplifier 47 ... DC isolation capacitor 30 ... circuits 50,51 ... DC power supplies 31,32,50 ... EMI filters 52 ... AC power supplies 33, 34, 44, 52, 74, 81, 82 ... 53 node amplifiers 60 ... three-phase common mode filtering 40 ... full wave bridge connection rectifier 61 ... three-phase rectifier 40a ... filter capacitors 63,64 ... capacitors 41,43,51 ... impedance 70-73 ... filter 41 ... inverter 80,83 ... voltage regulation circuit 42 ... Voltage source 81, 82 ... Amplifier 42 ... AC motor 100 ... Active electromagnetic interference filter circuit 43 ... Ground C ... Capacitor 43a ... Ground terminal L ... Line 44 ... Output winding R ... Resistor 18