KR101031743B1 - Matrix converter and control method thereof - Google Patents

Abstract

A matrix converter and control method thereof are disclosed. According to an embodiment of the present invention, a matrix converter includes: a rectifier having a plurality of switches and rectifying a three-phase first AC voltage input from an input terminal into a DC voltage; A converter for converting the DC voltage into a second AC voltage of three phases and outputting the same to an output terminal; And a controller generating a first control signal for controlling on-off of the plurality of switches. In this case, the converter is located between a DC link terminal to which the DC voltage converted from the rectifier is applied and an inverter terminal having inverters corresponding to respective phases of the second AC voltage to amplify the DC voltage. And a network, the switches of the rectifier are switched when the inverter stage is in a zero voltage state and the DC link stage is in a zero current state.

Description

Matrix converter and control method thereof

The present invention relates to a matrix converter and a control method thereof, and more particularly, to a matrix converter and a method of controlling the same, which can reduce a layout area and power consumption by increasing the voltage transfer rate and reducing the number of power semiconductor switches required.

The matrix converter combines the rectifier stage and the inverter stage in one step by eliminating the electrolytic capacitor at the DC link stage of the converter. The input and output terminals are directly connected by a power semiconductor, allowing bidirectional power blur and independent control of the input power factor.

Matrix converters can be implemented in two ways. One is a direct matrix converter (DMC) and the other is an indirect matrix converter (IMC). Direct matrix converter (DMC) is an ac-ac direct conversion method that directly connects the input and output terminals using nine bidirectional switches. The indirect matrix converter (IMC) is an indirect conversion method that removes the electrolytic capacitor at the DC link stage from the existing ac-dc-ac converter. Electrolytic capacitors are polar, as is well known.

Matrix converters offer the advantages of bidirectional power blur, input power factor control, and minimal use of energy storage devices. However, despite these advantages of matrix converters, matrix converters are still not widely used commercially. Complex control algorithms and a large number of power switches are required and, above all, low voltage transfer rates.

In order to solve the problem of complicated control, various kinds of modulation techniques have been developed, such as indirect spatial vector modulation technique and direct spatial vector modulation technique. However, whatever modulation scheme is used, the matrix converter is directly connected to the input and output, so the output voltage must be included in the enveloping curve of the input voltage. Under these conditions, the matrix converter must follow the equation

Vo ≤ Vi (√3 / 2) ｜ cosφ ｜

Here, Vi and Vo represent the magnitudes of the maximum input voltage and the maximum output voltage, respectively, and φ represents the input displacement angle. Therefore, if the input power factor is fixed to 1 (cosφ = 1), the maximum output voltage that can be obtained without overmodulation is √3 / 2 (= 0.866) of the maximum input voltage.

In other words, due to the structure of the matrix converter, the limit of the intrinsic voltage transfer rate is 0.866. This low voltage transfer rate is the most serious limitation in not being able to use the matrix converter as a drive for a standard motor. In order to use the matrix converter according to the prior art, it is necessary to lower the output of a standard motor or to make a special motor for the matrix converter.

Many studies have been conducted to overcome the inherent limitations of voltage transfer rates, but most of these studies are limited to overmodulation techniques. However, the overmodulation technique has a problem of lowering the quality of the output voltage and the input current by generating lower harmonics in the output voltage and the input current waveform instead of obtaining a high voltage transfer rate.

An object of the present invention is to provide a matrix converter and a control method thereof that can improve the voltage transfer rate without degrading the quality of the output voltage and the input current.

According to an aspect of the present invention, there is provided a matrix converter including: a rectifier having a plurality of switches and rectifying a three-phase first AC voltage input from an input terminal into a DC voltage; A converter for converting the DC voltage into a second AC voltage of three phases and outputting the same to an output terminal; And a controller generating a first control signal for controlling on-off of the plurality of switches.

In this case, the converter is located between a DC link terminal to which the DC voltage converted from the rectifier is applied and an inverter terminal having inverters corresponding to respective phases of the second AC voltage to amplify the DC voltage. And a network, the switches of the rectifier are switched when the inverter stage is in a zero voltage state and the DC link stage is in a zero current state.

Advantageously, a control switch located between the DC link stage and one end of the impedance source network, the control switch being on when the inverter stage is nonshoot-through and off when the inverter stage is dark-short. May be further provided.

Preferably, the impedance source network comprises: a first inductor having one end connected to one end of the control switch and the other end connected to one end of the inverter end; A second inductor having one end connected to the other end of the DC link end and the other end connected to the other end of the inverter end; A first capacitor connected between one end of the first inductor and the other end of the second inductor; And a second capacitor connected between the other end of the first inductor and one end of the second inductor.

Inductances of the first inductor and the second inductor may be the same, and capacitances of the first capacitor and the second capacitor may be the same.

When the inductance and the capacitance are sufficiently large, the amount of change in current in the dark short state is a value obtained by dividing the inductance by the voltage applied across the first capacitor or the second capacitor by the dark short unit time, The change amount of the current in the non-dark short state is a value obtained by subtracting the inductance and multiplying the non-dark short unit time by a value obtained by subtracting the voltage across the first capacitor or the second capacitor from the instantaneous average voltage of the DC voltage, During the switching period in which the dark short unit time and the non-dark short unit time are added, the sum of the change amount of the current in the dark short state and the change amount of the current in the non-dark short state may be zero.

In this case, the voltage applied to the inverter stage may be a product of a boosting factor obtained by dividing the switching period by the non-dark short unit time minus the dark short unit time and the instantaneous average voltage of the DC voltage.

In this case, the second AC voltage is a space vector modulation technique for the voltage applied to the inverter stage, vi is the voltage applied to the inverter stage, V _dc is the product of the instantaneous average voltage of the DC voltage and m _i is When the modulation index of the inverter stage (modulation index) and B is the boost ratio, it can be represented by the following equation.

In this case, the voltage transfer rate of the matrix converter is divided by the maximum phase voltage Vm of the first AC voltage from the second AC voltage, and cosθ _in represents the maximum value of the absolute values of the input power factors of the first AC voltage. It can be represented by the following equation.

Preferably, the dark short state may be inserted when the inverter stage is in a non-dark short zero voltage state. In addition, the non-cancer short-circuit zero voltage state, the switches connected to one end of the impedance source network of the plurality of switches included in the inverter stage are all turned on, the switches connected to the other end of the impedance source network It can be formed when both are off.

Preferably, the plurality of switches, each of the power semiconductor switching element; And a plurality of diodes connected to the power semiconductor switching element in a forward and reverse direction.

Preferably, the second AC voltage may have an amplitude different from that of the first AC voltage.

Preferably, the matrix converter may be a sparse matrix converter.

The matrix converter and its control method according to the present invention have the advantage of overcoming the low voltage transfer rate while reducing the number of power semiconductor switches while maintaining the advantages of the existing matrix converter.

DETAILED DESCRIPTION In order to fully understand the present invention, the operational advantages of the present invention, and the objects achieved by the practice of the present invention, reference should be made to the accompanying drawings which illustrate preferred embodiments of the present invention and the contents described in the drawings.

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings. Like reference numerals in the drawings denote like elements.

1 is a diagram illustrating a matrix converter according to an embodiment of the present invention.

Referring to FIG. 1, the matrix converter 100 according to an embodiment of the present invention may be a sparse matrix converter having twelve semiconductor switching elements. The matrix converter 100 according to the embodiment of the present invention includes a rectifier 120, a controller 160, and a converter 140.

The rectifier 120 converts the first AC voltage of the three input phases as shown in FIG. 2B input to the input terminal 110 into a DC voltage as shown in FIG. 2A. To this end, the rectifier 120 includes a plurality of bidirectional switches SW.

Each of the bidirectional switches SW is implemented with a power diode D capable of bidirectional power flow and power semiconductor switching elements Sap, San, Sbp, Sbn, Scp, and Scn. As shown in FIG. 1, the power semiconductor switching elements Sap, San, Sbp, Sbn, Scp, and Scn may be connected to two forward power diodes and two reverse power diodes.

The controller 160 controls the on-off of the bidirectional switches SW, that is, the on-off of the power semiconductor switching elements Sap, San, Sbp, Sbn, Scp, Scn provided in the bidirectional switches SW. In order to control the off, the first control signal XCON1 applied to the gates of the power semiconductor switching elements Sap, San, Sbp, Sbn, Scp, and Scn is generated.

Specifically, referring to FIG. 1 and FIG. 2B, in order to maximize the absolute value of the voltage applied to the DC link terminal XY, the phases Va, Vb, and Vc of the first AC voltage are respectively, The three terminals a, b, and c of the input terminal 110 are input at intervals of? / 3. One of the phases Va, Vb, and Vc of the first alternating voltage is fixed to the anode X or the cathode Y of the DC link terminal XY according to the sign, and the semiconductor switching for power for the other two phases is performed. Through modulation of the devices, a sinusoidal input current is obtained.

For example, as shown in (c) of FIG. 2, in the section A, the first phase Va of the first alternating current having a positive sign is fixed to the anode X of the DC link terminal XY. The second phase Vb and the third phase Vc are alternately connected to the cathode Y of the DC link terminal XY. At this time, of the two power semiconductor switching elements Sap and San for the first phase Va, the power semiconductor switching element Sap connected to the anode X of the terminal a and the DC link terminal XY. Is on, the power semiconductor switching element Sbn connected to the cathode Y of the DC link terminal XY of the two power semiconductor switching elements Sbp and Sbn for the second phase Vb, and The power semiconductor switching elements Scn connected to the cathode Y of the DC link terminal XY of the two power semiconductor switching elements Scp and Scn for the third phase Vc are alternately turned on and off. Preferably, the first control signal XCON1 is generated. The other power semiconductor switching elements San, Sbp, Scp are turned off.

Next, in the section B, the third phase Vc of the first alternating current having a negative sign is fixed to the cathode Y of the DC link terminal XY, and the first phase Va and the second phase ( Vb) is alternately connected to the anode X of the DC link terminal XY. At this time, the power semiconductor switching element Scn connected to the terminal c and the cathode Y of the DC link terminal XY for the third phase Vc is turned on, and the anode X of the DC link terminal XY is turned on. The power semiconductor switching element Sap for the first phase Va and the power semiconductor switching element Sbp for the second phase Vb are alternately turned on and off so that the first control signal XCON1) is generated. The other power semiconductor switching elements San, Sbn, and Scp are turned off.

In the same manner, the on-off of the bidirectional switches SW, that is, the power semiconductor switching elements Sap, San, Sbp, Sbn, Scp, Scn, for the C to G sections is controlled.

The average output voltage (instantaneous average value) Vdc of the DC link stage XY in each section is alternately the voltage of the phase applied to the anode X of the DC link stage XY, and the cathode Y of the DC link stage XY. It has a value of the difference of the voltage of the phase applied to.

The controller 160 specifically controls the bidirectional switches SW to perform the switching operation only when the current I of the DC link terminal X-Y is "0". Specifically, the controller 160 when the inverter stage 144 of the converter 140, which will be described later, is switched to a free wheeling state, when no current flows in the DC link stage XY, the rectifier 120 The switching operation for the switches of the control may be performed.

1 and 2, by the above switching operation, the first AC voltage of the three input phases as shown in FIG. 2B is converted into a DC voltage as shown in FIG. 2A. As described above, Vdc of FIG. 2A represents an instantaneous average value of the DC link stage XY that can be obtained during the switching period T, and Udc represents the DC link stage that can be obtained during the switching period T XY) represents the total average value. The overall average value Udc of the DC link stage X-Y is about 1.5 times greater than the absolute value of the first AC voltage of the three phases. Hereinafter, unless otherwise stated, the voltage at the DC link terminal X-Y is represented by an instantaneous average value Vdc.

The average output voltage (instantaneous average value) Vdc of the DC link stage (X-Y) that can be obtained during one switching period in the rectifier 120 is expressed by Equation 1 below.

Vdc = 3Vm / (2 | cosθ _in |)

Vm of Equation 1 is the maximum value of the amplitude of each phase of the first AC voltage, cosθ _in represents the maximum value of the absolute value of the power factor of the first AC voltage of the three phases.

Referring back to FIG. 1, the converter 140 amplifies the DC voltage of the DC link terminal X-Y and converts it into a second AC voltage of three phases. The second alternating voltage may have an amplitude different from the first alternating voltage. The converted second phase AC voltage is output to the output terminal 150.

Converter 140 includes an inverter stage 144 that includes an impedance source network 142 and inverters IVT that are each connected to corresponding output terminals A, B, and C of output 150. .

The impedance source network 142 is connected between the rectifier 120 and the inverter stage 144, amplifies the voltage of the DC link stage X-Y and transfers the voltage to the inverter stage 144. By the boost function of the impedance source network 142, a wide voltage transfer rate can be obtained. Due to the high switching frequency, the impedance source network 142 can be operated as a voltage source inverter to which a constant direct current voltage is applied during one switching period.

The inverter stage 144 converts the voltage amplified by the impedance source network 142 into an alternating voltage and applies it to the output stage 150.

For example, the inverter IVT connected to the output terminal A of the output terminal 150 may change the ON / OFF of the two switches S _Ap and S _An so that the voltage of both ends of the impedance source network 142 may be changed. Apply one to output terminal A. The same applies to inverters connected to the other output terminals B and C. At this time, the third to control the on-off of the pair of switches (S _Ap , S _An ), (S _Bp , S _Bn ), (S _Cp , S _Cn ) included in each inverter IVT The control signal XCON3 or the inverted signal / XCON3 of the third control signal may be provided from the controller 160.

Impedance source network by controlling the on-off of a pair of switches (S _Ap , S _An ), (S _Bp , S _Bn ), (S _Cp , S _Cn ) included in each inverter IVT The voltage amplified by 142 is converted into an alternating voltage.

Thus, one of the pair of switches ((S _Ap , S _An ), (S _Bp , S _Bn ), (S _Cp , S _Cn )) (eg S _Ap ) is on and the other (eg , S _Ap is in a nonshoot-through state in which the switches S _Ap , S _Bp , and S _Cp connected to one end of the impedance source network 142 are all turned on and the impedance source network ( When all of the switches (S _An , S _Bn , S _Cn ) connected to the other end of the 142 are turned off, the inverter stage 144 may operate in a nonshoot-through zero-voltage state. have.

While the inverter stage 144 is operating in a non-dark short zero voltage state, the controller 160 applies the switches S _An , S _Bn , and S _Cn connected to the other ends of the impedance source network 142. At least one of the inversion signals / XCON3 of the control signal may be generated at the same logic level as the third control signal XCON3. Accordingly, when all of the switches S _Ap , S _Bp , and S _Cp connected to one end of the impedance source network 142 are turned on, the switches S _An , connected to the other end of the impedance source network 142 are turned on. At least one switch of S _Bn , S _Cn ) may be turned on.

In this case, both ends of the impedance source network 142 may be shorted so that the voltage across the impedance source network 142 may be "0". This state may be referred to as a shoot-through state or a shoot-through zero-voltage state.

The control unit 160 of the matrix converter 100 according to an embodiment of the present invention is when the control stage (S1) is turned on when the inverter stage 144 is in a non-dark short state, and the inverter stage 144 is in a dark short state. The second control signal XCON2 may be applied to the gate of the control switch S1 so that the control switch S1 is turned off. The matrix converter 100 according to an embodiment of the present invention charges energy to the impedance source network 142 when the inverter stage 144 is in a dark short state and applies a voltage to the inverter stage 144 when in the non-dark short state. Step up. When the inverter stage 144 is in the dark short state, the impedance source network 142 turns off the control switch SW1 and electrically disconnects the inverter stage 144 from the DC link terminal XY, thereby inverting the inverter stage 144. ) Can prevent overcurrent from flowing.

When the inverter stage 144 is in a shoot-through state, the converter 140 may be equivalently converted as shown in FIG. In addition, the converter 140 when the inverter stage 144 is in a nonshoot-through state may be equivalently converted as shown in FIG.

 Referring to FIG. 3, assuming that the inductors L1 and L2 have the same inductance L and the capacitors C1 and C2 have the same capacitance C, due to the symmetrical structure of the impedance source network 142, the following equation 2 is obtained. Can be.

VL1 = VL2 = VL

VC1 = VC2 = VC

Since the inductors L1 and L2 have the same inductance L, as shown in Equation 2, the voltage VL1 across the inductor L1 and the voltage VL2 across the inductor L2 may be the same as VL. Similarly, since it has the same capacitance C as capacitors C1 and C2, both voltage VC1 across capacitor C1 and voltage VC2 across capacitor C2 can be equal to VC.

Referring to FIG. 3A, as described above, the control switch S1 is turned off during the dark short state period (denoted by T0) during the switching period (denoted by T). The following equation (3) can be obtained from the equivalent circuit in the dark short state.

VL = VC

Vd = 2VC

vi = 0

In Equation 3, Vd denotes a voltage applied to the input terminal of the impedance source network 142, and vi denotes a voltage applied to the inverter terminal 144 from the impedance source network 142.

Referring to FIG. 3B, as described above, during the non-dark short state period T1, the inverter stage 144 is non-dark short. The following equation (4) can be obtained from the equivalent circuit in the non-dark short state.

VL = Vdc-VC

Vd = Vdc

vi = VC-VL = 2VC-Vdc

Equation 5 below shows the currents IL and IL 'of the inductor in the dark and non-short states.

IL0 and IL0 'in Equation 5 are initial current values of the inductor, respectively.

Assuming that the inductance L and the capacitance C are sufficiently large, the amount of current change in the dark short state and the non dark short state of Equation 5 can be obtained by using Equations 3 and 4 as shown in Equation 6 below.

ΔIL = (VL / L) T0 = (VC / L) T0

ΔIL '= (VL / L) T1 = ((Vdc-VC) / L) T1

During one switching period T = T0 + T1 in the steady state, the average voltage across the inductor should be " 0 ", so the amount of change in current flowing through the inductor during one switching period is also " 0 ". Therefore, the following equation (7) is established.

ΔIL + ΔIL '= (VC / L) T0 + ((Vdc-VC) / L) T1 = 0

From Equation 4 and Equation 7, the voltage vi applied to the inverter stage 144 may be expressed as Equation 8 below.

vi = 2VC-Vdc = 2 (T1 / (T1-T0)) Vdc-Vdc = (T / (T1-T0)) Vdc

In Equation 8, T / (T1-T0) is referred to as a boosting factor, and may have a value of 0 to infinity.

As shown in Equation 8, the DC voltage applied to the inverter stage 144 is boosted, and when using the space vector modulation method using the boosted voltage, the output voltage which is the three-phase AC voltage output to the output stage 150 is Equation 9 is as follows. Space vector modulation is a well known technique, and a detailed description thereof will be omitted.

Mi in Equation 9 represents a modulation index of the inverter stage 144, and B represents a boost ratio in Equation 8. Therefore, from the equations (1) and (9), the voltage transfer rate Gv of the matrix converter 100 according to the embodiment of the present invention can be represented by the following equation (10).

This is in contrast to the voltage transfer rate of the matrix converter according to the prior art does not exceed the maximum √3 / 2 (= 0.866), the matrix converter 100 according to an embodiment of the present invention may theoretically obtain an infinite gain.

4 is a diagram for describing an example of a control method of the matrix converter 100 of FIG. 1.

1 and 4, the first phase Va of the first AC voltage is fixed to the anode X or the cathode Y of the DC link terminal XY, and the first phase Va of the first AC voltage ( The second phase Vb or the third phase Vc of the first AC voltage is applied to the anode X or the cathode Y of the DC link terminal XY in which Va is not fixed. That is, while the power semiconductor switching element Sa (one of Sap and San) for the first phase Va of the first AC voltage is on, the power semiconductor switching for the second phase Vb of the first AC voltage is on. The power semiconductor switching elements Sc (one of Scp and Scn) for the elements Sb (one of Sbp and Sbn) and the third phase Vc of the first alternating voltage are controlled to be alternately turned on.

4, the switches of the rectifier 120 (Sa (one of Sap and San), Sb (one of Sbp and Sbn), Sc (one of Scp and Scn)) and the inverter stage 144 by on-off of the switches (S _a (S one of _Ap and S _An), one of the S _B (S _Bp and S _Bn), S _C (S _Cp and S either _Cn)), in interval t1 to t12 Are distinguished. At this time, all of the upper switches (S _Ap , S _Bp , S _Cp ) of the inverter stage 144 connected to one end of the impedance source network 142 are turned on or lower switches S _An of the inverter stage 144. In the periods t4 and t9 of the non-cancer short-circuit zero voltage state in which all of S _Bn and S _Cn )) are turned on, the control switch S1 is turned off and the upper and lower switches of the inverter stage 144 are simultaneously turned on. A dark short zero voltage state (hatched portions in sections t4 and t9) can be inserted.

In this way, the inverter stage 144 inserts the dark short state in the non-dark short zero voltage state, so that the inverter stage is in the zero voltage state and the current I of the DC link terminal XY becomes "0" for power. The semiconductor switching elements Sap, San, Sbp, Sbn, Scp, and Scn may be switched.

As such, the matrix converter and the control method thereof according to the embodiment of the present invention may include an impedance source network 142 to insert a dark short zero voltage state. Therefore, as described above, voltage boosting can be performed while preventing overcurrent from flowing between the impedance source network 142 and the DC link terminal X-Y.

That is, the DC-link zero current communication technique can be stably performed, and the number of switching elements of the matrix converter can be reduced. In addition, the matrix converter and the control method thereof according to an embodiment of the present invention can increase the voltage transfer rate by performing a boost function for the voltage of the DC link terminal by the impedance source network 142.

As described above, optimal embodiments have been disclosed in the drawings and the specification. Although specific terms are employed herein, they are used for purposes of describing the present invention only and are not used to limit the scope of the present invention. Therefore, those skilled in the art will understand that various modifications and equivalent other embodiments are possible therefrom. Therefore, the true technical protection scope of the present invention will be defined by the technical spirit of the appended claims.

BRIEF DESCRIPTION OF THE DRAWINGS In order to better understand the drawings cited in the detailed description of the invention, a brief description of each drawing is provided.

1 is a diagram illustrating a matrix converter according to an embodiment of the present invention.

FIG. 2 is a diagram for describing a switching operation of the matrix converter of FIG. 1.

FIG. 3 is an equivalent conversion circuit for the dark and non-short states of the converter of FIG. 1.

FIG. 4 is a diagram for describing an example of a control method of the matrix converter of FIG. 1.

Claims (13)

A rectifier having a plurality of switches and rectifying a three-phase first AC voltage input from an input terminal into a DC voltage; A converter for converting the DC voltage into a second AC voltage of three phases and outputting the same to an output terminal; And A controller for generating a first control signal for controlling the on-off of the switches of said rectifier, said converter comprising: An impedance source network positioned between the DC link terminal to which the DC voltage converted from the rectifier is applied and the inverter terminal including inverters corresponding to each phase of the second AC voltage to amplify the DC voltage, The switches of the rectifier are switched when the inverter stage is in a zero voltage state and the DC link stage is in a zero current state, The impedance source network, A first inductor having one end connected to one end of the control switch and the other end connected to one end of the inverter end; A second inductor having one end connected to the other end of the DC link end and the other end connected to the other end of the inverter end; A first capacitor connected between one end of the first inductor and the other end of the second inductor; And A second capacitor connected between the other end of the first inductor and one end of the second inductor, Inductances of the first inductor and the second inductor are the same, and the capacitances of the first capacitor and the second capacitor are the same, The change amount of the current of the inductor in the dark short state is a value obtained by dividing the inductance by the voltage applied across the first capacitor or the second capacitor by the dark short unit time, The amount of change in the current of the inductor in the non-dark short state is a value obtained by subtracting the inductance and multiplying the non-dark short unit time by a value obtained by subtracting the voltage across the first capacitor or the second capacitor from the instantaneous average voltage of the DC voltage. ego, During the switching period in which the dark short unit time and the non-dark short unit time are added, the sum of the amount of change in current in the dark short state and the amount of change in current in the non-short short state is zero, The voltage across the inverter stage is And the switching period is a product of a boosting factor divided by the non-dark short unit time minus the dark short unit time and the instantaneous average voltage of the DC voltage. According to claim 1, And a control switch located between the DC link stage and one end of the impedance source network and turned on when the inverter stage is in a nonshoot-through state and turned off when the inverter stage is in a dark short state. Matrix converter, characterized in that. The method of claim 2, wherein the dark short state, And the inverter stage is inserted when there is a non-dark short zero voltage state. The method of claim 3, wherein the non-dark short zero voltage state, Among the plurality of switches included in the inverter stage, the switches connected to one end of the impedance source network are all turned on, and the switches connected to the other end of the impedance source network are all formed when the off Matrix converter. delete delete delete delete The method of claim 1, wherein the second AC voltage, By performing a space vector modulation technique on the voltage across the inverter stage, vi is the voltage across the inverter stage, V _dc is the product of the instantaneous average voltage of the DC voltage, and m _i is the modulation index of the inverter stage. And B is expressed by the following equation when said boosting rate.

10. The method of claim 9, wherein the voltage transfer rate of the matrix converter, Dividing by the maximum phase voltage Vm of the first AC voltage from the second AC voltage, cosθ _in is represented by the following equation when representing the maximum value of the absolute value of the input power factor of the first AC voltage Matrix converter.

The method of claim 1, wherein each of the plurality of switches, Power semiconductor switching elements; And And a plurality of diodes connected to the power semiconductor switching element in a forward and a reverse direction. The method of claim 1, wherein the second AC voltage, And an amplitude different from said first alternating voltage. The method of claim 1, wherein the matrix converter, A matrix converter characterized by being a sparse matrix converter.

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