JPH11262269A - Control of pulse width modulated inverter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To prevent the decline in supply voltage for driving by keeping a second semiconductor switching element in an on-state in a 120 deg. region centered around a negative-side peak point of a sine wave power command for one phase, while alternately turning the switching element on and off so that the voltage command is equal to line voltages between the one phase and the other phases, in the other regions. SOLUTION: In a 120 deg. region A centered around a negative-side peak point of a U-phase fundamental voltage command, an actual U-phase voltage command is a command for keeping a second semiconductor switching element for U-phase connected to the negative pole side of a first DC power supply in an on-state. In the regions other than the one where the second semiconductor switching element for U-phase is kept at an on-state, the actual U-phase voltage command is a voltage command corresponding to the negative pole side potential of the first DC power supply as a reference with addition of UV line voltage value in this region. It is moreover similar with W-phase. With this method, there is no region where the second semiconductor switching element is continuous in an off-state so that the decline in supply voltage for driving can be prevented.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method for controlling a pulse width modulation type inverter device.

[0002]

2. Description of the Related Art In a pulse width modulation type inverter device, FIG. 6 shows a pulse width modulation output form in a first conventional example, and FIG. 7 shows a pulse width modulation output form in a second conventional example. It is a thing. FIG. 8 shows a pulse width modulation output form in the third conventional example described in Japanese Patent Application Laid-Open No. 1-267668. FIG. 9 shows a configuration example of an inverter device implementing the pulse width modulation output mode in the first to third conventional examples. In FIG. 6 showing the first conventional example, a basic triangular wave for generating a U-phase pulse width modulation output, a U-phase sine wave (fundamental wave) voltage command, and a third harmonic with respect to the U-phase fundamental wave voltage command are shown. Component command, an actual U-phase voltage command obtained by synthesizing the fundamental wave voltage command and the third harmonic component command, and an IGBT transistor Q1 from the microcomputer 108 generated by comparing the magnitude of the actual U-phase voltage command with the basic triangular wave. An on / off command to Q4 is shown. Although omitted in the figure,
Regarding the ON / OFF command from the microcomputer 108 to the IGBT transistors Q1 and Q4, Q1 and Q4
A dead time is provided during which both of the IGBT transistors are turned off so that both are not turned on at the same time. In FIG. 6, the microcomputer 10
In order to generate the output voltage from the U-phase terminal by pulse width modulation, a comparison is made between the actual U-phase voltage command and the basic triangular wave.
An ON command for the BT transistor Q4 is output (an OFF command for Q1), and an ON command for the IGBT transistor Q1 (an OFF command for Q4) is output in a section where the basic triangular wave is smaller. . As a result, the output voltage of the U-phase terminal during the ON period of the IGBT transistor Q1 becomes the positive-side potential of the first DC power supply 101 in FIG.
During the ON period of the T-transistor Q4, the potential is on the negative electrode side, but the average output voltage matches the actual U-phase voltage command value. In order to increase the maximum sine wave voltage that can be output, the actual U-phase voltage command generated by the microcomputer 108 is generated by combining the U-phase fundamental voltage command and the third harmonic component command as shown in FIG.
In FIG. 7 showing a second conventional example (this is referred to as a two-phase modulation output method), a basic triangular wave, a U-phase sine wave (basic wave) voltage command for producing a U-phase pulse width modulation output, and an actual U-phase A voltage command, and an on / off command to the IGBT transistors Q1 and Q4 from the microcomputer 108, which are created by comparing the actual U-phase voltage command with the basic triangular wave, are shown. Although not shown in the figure, the microcomputer 108 supplies the IGBT transistors Q1 and Q4
A dead time is provided for the ON / OFF command to turn off both IGBT transistors so that Q1 and Q4 do not turn on at the same time. Also in FIG. 7, as in the embodiment of FIG. 6, the microcomputer 108 compares the actual U-phase voltage command with the basic triangular wave to generate an output voltage from the U-phase terminal by pulse width modulation. To increase the maximum sine wave voltage that can be output, the microcomputer 10
8 produces the actual U-phase voltage command as shown in FIG.
The IGBT transistor Q1 is continuously turned on in the 60-degree section around the positive peak point of the phase fundamental wave voltage command, and the IGBT transistor Q4 is continuously turned on in the 60-degree section around the negative peak point. It has become. By doing so, the maximum sine wave voltage that can be output can be increased, and the number of switching operations of the IGBT transistor is reduced to reduce the switching loss. FIG. 8 (which is also a two-phase modulation output system) showing a third conventional example described in JP-A-1-274668 discloses a basic triangular wave and a U-phase sine wave for producing a U-phase pulse width modulation output. (Fundamental wave) voltage command, actual U-phase voltage command, and on / off command from microcomputer 108 to IGBT transistors Q1, Q4 generated by comparing actual U-phase voltage command with basic triangular wave. . Although omitted in the figure,
Regarding the ON / OFF command from the microcomputer 108 to the IGBT transistors Q1 and Q4, Q1 and Q4
A dead time is provided during which both of the IGBT transistors are turned off so that both are not turned on at the same time. Similarly, in FIG. 8, the microcomputer 108 compares the actual U-phase voltage command with the basic triangular wave to generate an output voltage from the U-phase terminal by pulse width modulation. In order to increase the maximum sine wave voltage that can be output, the actual U-phase voltage command generated by the microcomputer 108 is, as shown in FIG. 8, an IGBT in the 60-degree section starting from the positive peak point of the U-phase fundamental wave voltage command. The transistor Q1 is continuously turned on,
Conversely, the 60-degree section starting from the negative peak point is I
The GBT transistor Q4 is turned on continuously. By doing so, the maximum sine wave voltage that can be output can be increased, and the switching frequency of the IGBT transistor is reduced to reduce the switching loss.
When the output current (motor phase current) is small, the switching operation of the IGBT transistor can be concentrated when the peak point of the motor phase current, which has a phase delay of about 0 degree, coincides with the continuous ON operation period of the IGBT transistor.
Further, the switching loss is reduced.

[0003]

However, the first aspect
In the pulse width modulation output form shown in the conventional example, there is a problem that the switching loss of the IGBT transistor is large, and in the pulse width modulation output forms shown in the second and third conventional examples, as shown in FIG. In FIG. 3, the IGBT transistor Q4 (or Q5 or Q6) is turned on in FIG.
IGBT transistor in a bootstrap power supply circuit for charging current to a capacitor 15 (or 16 or 17) serving as a power supply for driving the IGBT transistor Q1 (Q2 or Q3) via a diode 24 (or 25 or 26) There is a section where Q4 is continuously turned off for a long time (see FIG. 7 and FIG. 8). Therefore, during this section, there is no current charging to the capacitor 15, so that the voltage between the terminals of the capacitor 15 decreases due to discharging. There is a problem that it is difficult to secure a drive power supply voltage for IGBT transistor Q1. Further, since the U-phase output current detection from the voltage value between the terminals of the shunt resistor 19 is possible only when the transistor Q4 is on (including the reflux mode in which the motor current flows through the diode D4),
Conversely, if there is a section where the IGBT transistor Q4 is continuously turned off for a long time (see FIGS. 7 and 8), there is a problem that the phase output current cannot be detected during that section.

[0004]

In order to solve the above-mentioned problems, the present invention provides a first semiconductor switching element connected to a positive electrode of a DC power supply, and a first semiconductor switching element connected in parallel to the first semiconductor switching element. A freewheeling diode, a second semiconductor switching element connected to the negative side of the DC power supply, a second freewheeling diode connected in parallel to the second semiconductor switching element, and both first and second semiconductor switching elements. The elements are connected in series and the connection point is used as the output terminal of the inverter device. The output terminal having such a configuration is provided for three phases, and the ON / OFF time of the first and second semiconductor switching elements for three phases is provided. And a drive circuit for driving each of the semiconductor switching elements on and off in response to an on / off signal from the processing apparatus to each of the semiconductor switching elements In the pulse width modulation type inverter device, the second semiconductor in the phase is a 120-degree section centered on the negative peak point of the phase output voltage for each cycle of the sine wave output voltage command frequency of the inverter device. The switching element is continuously turned on, and after the elapse of the 120-degree section in the phase, the continuous on operation is sequentially switched to the next phase, and the output voltage of the phase during the continuous on operation is set to the negative potential of the DC power supply. As there are
For the other two phases, the first in each phase
And the second semiconductor switching element are turned on and off alternately within one carrier cycle at a ratio equal to the voltage between the sinusoidal lines with the phase.

[0005]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS In the control method of a pulse width modulation type inverter device according to the present invention, a pulse width modulation output form is shown in FIGS. In FIG. 1, a U-phase fundamental (sine wave) voltage command, a V-phase fundamental (sine wave) voltage command,
A W-phase fundamental (sine wave) voltage command and an actual U-phase voltage command are shown. In the section A in FIG. 1, the actual U-phase voltage command is fixed to the negative potential of the first DC power supply. The actual V and W phase voltage commands are commands obtained by adding the VU and WU line voltages, respectively, based on the negative potential. In the section B, the actual V-phase voltage command is fixed to the negative potential of the first DC power supply. The actual U and W phase voltage commands are commands obtained by adding the UV and WV line voltages, respectively, based on the negative electrode potential. In the section C, the actual W-phase voltage command is fixed to the negative potential of the first DC power supply. The actual U and V phase voltage commands are commands obtained by adding the UW and VW line voltages, respectively, based on the negative potential. In FIG. 2, an actual U-phase voltage command, an actual V-phase voltage command, an actual W-phase voltage command,
A basic triangular wave for generating an on / off command to the semiconductor switching element by comparing the magnitude of the actual phase voltage command with the actual phase voltage command by the arithmetic unit, and is generated by the arithmetic unit by comparing the magnitude of the actual phase voltage command with the basic triangular wave. ON / OFF to the first and second semiconductor switching elements in each phase
3 shows an off drive signal.

First, in a 120-degree section (section A) centered on the negative peak point of the U-phase fundamental wave voltage command,
The actual U-phase voltage command is a command for continuously turning on the second semiconductor switching element in the U-phase connected to the negative electrode side of the first DC power supply. After the elapse of this 120 degree section, in the next 120 degree section (120 section centering on the negative peak point of the V-phase voltage command: section B), a command to continuously turn on the second semiconductor switching element in the V phase is issued. Then, the next 120-degree section (120 section around the negative peak point of the W-phase voltage command:
In the section C), the command is switched to the command for continuously turning on the second semiconductor switching element in the W phase, and after this section, the procedure returns to the U phase again and the same command is repeated. In the remaining section other than the section in which the actual U-phase voltage command is a continuous ON operation command to the second semiconductor switching element in the U phase, first, the section in which the second semiconductor switching element in the V phase is continuously turned on is Since the V-phase terminal voltage becomes the negative-side potential of the first DC power supply,
A voltage command is obtained by adding the UV line voltage value (see FIG. 1) in this section with reference to the negative electrode side potential of the DC power supply of the DC power supply. In the W phase, the section in which the second semiconductor switching element is continuously turned on is the W phase. Since the terminal voltage becomes the negative potential of the first DC power supply, the voltage command is obtained by adding the UW line voltage value (see FIG. 1) in this section with reference to the negative potential of the first DC power supply.

When this is expressed by a specific mathematical expression, in the configuration of the inverter device shown in FIG. 3, the first DC power supply voltage of the first DC power supply 1 made by rectifying the AC 200 V commercial AC power supply 14 with the rectifier diode 13 is: V ₁ = 200 × 2
^1/2 (V). Here, assuming that the effective value of the output voltage between the sine wave lines of the inverter is VO, the effective value of the output voltage of the fundamental wave (sine wave) of each phase of the inverter is V
O / 3 ^1/2 U-phase fundamental voltage command is EU1 = 2 ^1/2 x VO / 3 ^1/2 x
SINθ V-phase fundamental wave voltage command is EV1 = 2 ^1/2 × VO / 3 ^1/2 ×
SIN (θ-120) W phase fundamental wave voltage command is EW1 = 2 ^1/2 × VO / 3 ^1/2 ×
SIN (θ-240). When the actual U-phase voltage command is EU, the actual V-phase voltage command is EV, and the actual U-phase voltage command is EW,

(1) In the section U where the second semiconductor switching element in the 210 ≦ θ ≦ 330-U phase is continuously on,
The actual voltage commands for the V and W phases are respectively EU = 0 EV = (VU line voltage) + (actual U-phase voltage command) =
(EV1−EU1) + EU = 2 ^1/2 × VO × SIN (θ
-150) EW = (WU line voltage) + (actual U-phase voltage command) =
(EW1-EU1) + EU = 2 ^1/2 × VO × SIN (θ
+150). Also, for each phase at that time, the actual line voltage command is the actual UV line voltage command = EU-EV = EU-((EV
1-EU1) + EU) = EU1-EV1 Actual VW line voltage command = EV-EW = ((EV1-E
U1) + EU)-((EW1-EU1) + EU) = EV
1-EW1 Actual WU line voltage command = EW-EU = ((EW1-E
U1) + EU) -EU = EW1-EU1, which is equal to the line voltage command synthesized by the fundamental (sine wave) voltage command of each phase.

(2) A section U in which the second semiconductor switching element in the 90 ≦ θ ≦ 210-W phase is continuously ON.
The actual voltage commands for the V and W phases are respectively EU = (UW line voltage) + (actual W-phase voltage command) =
(EU1-EW1) + EW = 2 ^1/2 × VO × SIN (θ
-30) EV = (VW line voltage) + (actual W-phase voltage command) =
(EV1−EW1) + EW = 2 ^1/2 × VO × SIN (θ
−90) EW = 0. Also, for each phase at that time,
The actual line voltage command is the actual UV line voltage command = EU-EV = ((EU1-E
W1) + EW)-((EV1-EW1) + EW) = EU
1-EV1 Actual VW line voltage command = EV-EW = ((EV1-E
W1) + EW) -EW = EV1-EW1 Actual WU line voltage command = EW-EU = EW-((EU
1−EW1) + EW) = EW1−EU1, which is also equal to the line voltage command synthesized by the fundamental (sine wave) voltage command of each phase.

(3) In a section U in which the second semiconductor switching element in the -30 ≦ θ ≦ 90-V phase is continuously on,
The actual voltage commands for the V and W phases are respectively EU = (UV line voltage) + (actual V-phase voltage command) =
(EU1−EV1) + EV = 2 ^1/2 × VO × SIN (θ
+30) EV = 0 EW = (WV line voltage) + (actual V-phase voltage command) =
(EW1−EV1) + EV = 2 ^1/2 × VO × SIN (θ
+90). Also, for each phase at that time, the actual line voltage command is the actual UV line voltage command = EU-EV = ((EU1-E
V1) + EV) -EV = EU1-EV1 Actual VW line voltage command = EV-EW = EV-((EW
1-EV1) + EV) = EV1-EW1 Actual WU line voltage command = EW-EU = ((EW1-E
V1) + EV)-((EU1-EV1) + EV) = EW
1-EU1, which is also equal to the line voltage command synthesized by the fundamental (sine wave) voltage command of each phase.

As described above, in all the sections (1) to (3), the actual line voltage command of each phase and the line voltage command synthesized from the fundamental (sine wave) voltage command of each phase are as follows. It turns out that they are always equal. Therefore, (1) to (3)
, The fundamental (sine wave) phase voltage is output from each phase output terminal of the inverter device.
Furthermore, V in each of the above equations in the actual phase voltage command
As can be seen from O, up to 200 (V), which is the effective value of the AC voltage of the commercial AC power supply 14, can be actually set and output as the maximum value. In other words, VO, which is the sine wave line output voltage effective value of the inverter device, can be output up to 200 (V), which is the AC voltage effective value of the AC power supply 14 at the maximum. Therefore, when the first and second semiconductor switching elements are turned on and off as described in claim 1, the phase output voltage of the inverter device is set to a sine wave voltage output, and the effective line-to-line voltage output value of the inverter device is determined. Can output up to the effective value of the AC power supply voltage at the maximum. Further, since there is no section where the second semiconductor switching element is continuously turned off as shown in FIG. 2, the voltage between terminals of the capacitor 15 serving as a power supply for driving the first semiconductor switching element in the U phase in FIG. Nothing to do.

In the U-phase output current detection in the configuration of FIG. 3, if the inverter device does not output up to the maximum outputtable voltage, the second semiconductor switching element in the U-phase must be at the peak point of the basic triangular wave shown in FIG. As a result, the U-phase output current can always be detected, and two of the three phases having the smaller actual phase voltage command value (for one of the three phases, the second semiconductor switching element is operated during the continuous ON operation). 2) is selected by the arithmetic unit in each case, and the phase output current of the two phases is detected at the peak point of the basic triangular wave shown in FIG. 2, so that the AC power supply voltage effective value which is the maximum outputtable voltage of the inverter device is obtained. Since the two phase output currents can be detected even when the voltage is being output, the remaining one phase output current can be detected for three phases. The sum of the force current is can be performed is obvious a is thus always U-phase output current detection by utilizing the relationship of zero.

FIG. 3 shows an example of the configuration of an inverter device implementing the pulse width modulation output mode according to the present invention. FIG.
, The microcomputer 8 compares the magnitude of the built-in basic triangular wave with the actual phase voltage command generated by the operation to generate ON / OFF signals to the IGBT transistors Q1 to Q6, and divides the signal into the respective IGBTs. The signal is transmitted to the on / off drive circuit units 2 to 7 of the transistors. According to the transmitted on / off command signal, each of the on / off drive circuit units performs the on / off drive operation of each IGBT transistor. At this time, for example, in the U-phase, while the IGBT transistor Q4 is on, the U-phase output terminal voltage becomes substantially equal to the negative potential of the first DC power supply 1, and as shown in FIG. 18 to diode 24, capacitor 15, I
A current will flow to the GBT transistor Q4,
The charging operation of the capacitor 15 is performed by this current,
Immediately, the voltage between the terminals of the capacitor 15 is
8 is equal to the voltage value. On the other hand, IGBT transistor Q
4 is turned off, the voltage across the terminals of the capacitor 15 decreases due to discharge.
Since the off operation of the GBT transistor Q4 does not last for a long time, the capacitor 15 is immediately charged again, so that the voltage between the terminals of the capacitor 15 serving as the power supply for driving the IGBT transistor Q1 is greatly reduced. do not do. Further, the microcomputer 8 selects two phases having small actual phase voltage command values and detects the phase output currents of the two phases at the upper peak point of the built-in basic triangular wave (the phase currents of the three phases in the case of the three-phase motor drive). Is zero, the phase output current value of the remaining one phase becomes obvious if the phase output currents of two phases can be detected. Therefore, the phase output current detection is sufficient for two phases). Further, the microcomputer 8 detects the instantaneous phase output current at the same time, so that the sample and hold circuits 22 and 23 for the V-phase and W-phase current detection outputs from the operational amplifiers 31 and 32 respectively. After that, the detected value of the phase output current is taken in by the built-in A / D conversion port.

[0014]

As described above, according to the present invention, the 120-degree section centered on the negative peak point of the phase fundamental wave (sine wave) voltage command is connected to the negative side of the first DC power supply. The second semiconductor switching element is continuously turned on, and for the other sections in the other sections during the continuous on operation, the negative potential of the first DC power supply, which is the output potential of the phase, is set as a reference potential. By performing pulse width modulation output so that the voltage output becomes equal to the line voltage with the phase, IGB
The switching loss of the T transistor can be reduced, and the section in which the second semiconductor switching element connected to the negative electrode side of the first DC power supply is continuously turned off is eliminated, so that the first semiconductor switching element is provided by the bootstrap power supply circuit. In the case where the drive power supply of the present invention is configured, the drive power supply voltage can be prevented from lowering, and the phase output current can always be detected even when the current detection circuit of the inverter device is configured as shown in FIG. is there.

[Brief description of the drawings]

FIG. 1 is an explanatory diagram of a voltage command to each phase in an embodiment of the present invention.

FIG. 2 shows a pulse width modulation output form in an embodiment of the present invention.

FIG. 3 is a configuration example of an inverter device that performs pulse width modulation output according to the present invention.

FIG. 4 shows a charging current route to the capacitor 15 during the ON operation of the IGBT transistor Q4 (the present invention).

FIG. 5 is a diagram showing selection by a two-phase arithmetic unit with a small actual phase voltage command (the present invention)

FIG. 6 shows a pulse width modulation output form in a first conventional example of a pulse width modulation type inverter device.

FIG. 7 is a pulse width modulation output form in a second conventional example of a pulse width modulation type inverter device.

FIG. 8 shows a pulse width modulation output form in a third conventional example of a pulse width modulation type inverter device.

FIG. 9 is a configuration example of an inverter device that performs pulse width modulation output according to a conventional example.

[Explanation of symbols]

Q1 to Q6 IGBT transistors D1 to D6 Freewheeling diode 1 First DC power supply 2 to 7 On / off drive circuit section of each IGBT transistor Q1 to Q6 8 Microcomputer 9 Three-phase induction motor 10 to 12 Each phase output of inverter device Terminal 13 Rectifier diode section 14 Commercial AC power supply 15 to 17 Capacitor to be a drive power supply for each IGBT transistor Q1 to Q3 18 Second DC power supply 19 to 21 Output current detection shunt resistor 22, 23 V phase for each phase And W phase current detection section sample and hold circuit 24 to 26 Diode 27 to 29 Reference voltage of each phase current detection section 30 to 32 Operational amplifier of each phase current detection section 33 to 38 Resistance of each phase current detection section 101 First DC Power supply 102-107 ON / OFF drive circuit for each IGBT transistor Q1-Q6 Unit 108 Microcomputer 109 Three-phase induction motor 110-112 Each-phase output terminal of inverter device 113 Rectifier diode unit 114 Commercial AC power supply 115-117 Capacitor serving as power supply for driving IGBT transistors Q1-Q3 118 Second DC power supply 119, 120 U-phase and V-phase output current detector 121 V-phase current detection section sample hold circuit

Claims (1)

[Claims] 1. A first semiconductor switching element connected to a positive electrode of a DC power supply, a first freewheeling diode connected in parallel to the first semiconductor switching element, and a negative electrode connected to the DC power supply. A second semiconductor switching element, a second freewheeling diode connected in parallel to the second semiconductor switching element, and a series connection of the first and second semiconductor switching elements and a connection point of the inverter device. An output terminal, an output device having such a configuration for three phases, an operation device for controlling the on / off time of the first and second semiconductor switching elements for the three phases, and an output device from the operation device. ON / OFF to each semiconductor switching element
Each of the semiconductor switching elements is turned on / off by an off signal.
In the control method of the pulse width modulation type inverter device including a driving circuit unit that is turned off, a 120-degree section centered on the negative peak point of the phase output voltage for each cycle of the sine wave output voltage command frequency The second semiconductor switching element in a phase is continuously turned on, and after a lapse of a 120-degree interval in the phase, the continuous on operation is sequentially switched to the next phase, and the output voltage of the phase during the continuous on operation is the Assuming that the potential is at the negative electrode side potential of the DC power supply, for the remaining two phases, the first and second semiconductor switching elements in each phase are set to 1 at a ratio that is equal to the sine-wave line voltage between the phases. A method for controlling a pulse width modulation type inverter device, wherein the operation is alternately turned on and off within a carrier cycle.