JP3788346B2 - Voltage type PWM inverter control device - Google Patents

Description

[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a control apparatus for a voltage-type PWM inverter that outputs an AC voltage having a variable voltage and a variable frequency in order to drive an AC motor or the like.
[0002]
[Prior art]
In a voltage-type PWM inverter that drives an AC motor with an AC voltage of variable voltage and variable frequency, the incompleteness of the freewheeling diode connected in reverse parallel to the switching element during the dead time period provided to prevent the short circuit of the upper and lower arms of the inverter Due to the turn-on, the output voltage becomes discontinuous at and near the zero crossing point of the phase current (hereinafter simply referred to as the zero crossing point), and the phase current also becomes discontinuous (current discontinuity phenomenon or current stagnation phenomenon). The waveform may be distorted. As a result, inconveniences such as the pulsation of the torque of the AC motor as the load and the fluctuation of the rotation speed occur in the vicinity of the zero cross point of the phase current.
[0003]
Therefore, conventionally, (1) each phase output voltage of the inverter is detected and the inverter output voltage command value is compensated by the voltage regulator. (2) each phase output current of the inverter is detected and the current regulator is used. The method of compensating the output current command value is adopted, and the torque pulsation and speed fluctuation in the vicinity of the current zero cross point are suppressed by these compensation operations (referred to as dead time compensation).
[0004]
[Problems to be solved by the invention]
Among the conventional methods described above, (1) Compensation by a voltage regulator requires a highly accurate and highly responsive voltage detector for detecting the output voltage, and this type of voltage detector is generally expensive. Therefore, there is a problem that the cost of the entire control device increases.
In addition, (2) compensation by the current regulator requires the electric constant of the motor in order to create a current command value, so that there is a problem that calculation for compensation becomes complicated.
These problems are described in, for example, “3. Configuration of each speed estimation method” in “Current Status of Research and Development at Manufacturers (Current Situation and Challenges in Application of Induction Motor Speed Sensorless Vector Control)” Is also pointed out.
[0005]
Accordingly, the present invention provides a control device for a voltage-type PWM inverter that performs dead time compensation without requiring complicated calculation and suppresses torque pulsation and speed fluctuation in the vicinity of the current zero cross point at a relatively low cost. It is something to try.
[0006]
In order to solve the above problems, the present invention is basically based on generating a voltage compensation amount that cancels the distortion in the vicinity of the zero cross point where the phase current is distorted, and adding this compensation amount to the voltage command value.
That is, in order to prevent the phase current from stagnating near zero near the zero cross point, the phase current is decomposed into a vector based on the voltage command value generated by the voltage-type PWM inverter, and the phase advanced from the decomposed vector. A current command value is generated, and a voltage compensation amount is created based on the current command value and the phase current detection value.
[0007]
If the phase current amplitude value decreases during the dead time, the phase current stagnates near zero, and an interval in which the output voltage becomes unstable occurs. This is because the voltage at the dead time is determined by the polarity and magnitude of the phase current. This operation will be described below.
[0008]
FIG. 3 shows one phase (for example, U phase) of the main circuit of the voltage source PWM inverter.
In FIG. 3, E _dc is a DC power source, T ₁ is an upper arm comprising a switching element S ₁ such as an IGBT and a free-wheeling diode _DF antiparallel to this, and T ₄ is also the switching element S ₄ and the free-wheeling diode D _F. , P is a positive terminal, N is a negative terminal, and U is a U-phase output terminal.
[0009]
Further, the equivalent model of the freewheeling diode _{D F} in FIG. 4, (the voltage between the terminals _UN) arms voltage when an operation mode of the phase current _{i U} in FIG. 5, the phase current _{i U} in FIG. 6 large _{E UN} FIG. 7 shows the behavior of the arm voltage E _UN when the phase current i _U is small.
[0010]
Now, the phase current i _U flows in the direction shown in FIG. 3, the switching elements S ₁ and S ₄ are ideal switches, and the free wheel diode _DF is a parallel circuit of a resistor R _D and a capacitor C _j as shown in FIG. It shall be represented by Note that C _j1 and C _j4 in FIG. 5 correspond to the capacitor C _j in FIG.
[0011]
The operation mode of the phase current i _U when the switching element S ₁ changes from the on state to the off state and the switching element S ₄ changes from the off state to the on state is as shown in FIG. 5 (a) → (b) → (c). It becomes like this.
In this operation process, the arm voltage E _UN in the section in which both the switching elements S ₁ and S ₄ are off (corresponding to the section 2 in FIG. 5B and FIG. 6) is the phase flowing in the free wheel diode _DF. Determined by the magnitude of the current. When this phase current is large, the charge accumulated in the pn junction of the freewheeling diode _DF is rapidly discharged, so that the freewheeling diode _DF rapidly recovers and the change in arm voltage (dv / dt) becomes large.
At this time, as shown in FIG. 6, the arm voltage E _UN changes in substantially the same manner as the PWM signal, so that the output voltage error is small.
[0012]
However, when the phase current is small, the charge discharge rate of the pn junction described above becomes slow and the change in arm voltage (dv / dt) becomes small. That is, as shown in FIG. 7, the arm voltage E _UN takes a value like (1) or (2) in the section 2 and does not change in the same manner as the PWM signal, so that the output voltage error becomes large.
The arm voltage E _UN varies depending on the magnitude of the phase current. When the phase current is small, the arm voltage E _UN is as shown in (1), and changes from (1) → (2) → (3) as the phase current increases. .
[0013]
That is, the voltage error in the interval 2 depends on the phase current and the reverse recovery charge of the freewheeling diode _DF . Further, due to these relationships, the voltage error maximum value ε _v ^max can be expressed by the following Equation 1. Here, t _d : dead time, f _c : carrier frequency, E _dc : DC intermediate voltage.
[0014]
[Equation 1]
ε _v ^max = t _d · f _c · E _dc
[0015]
Further, the phase current _{ID in which} a voltage error occurs (changes) can be expressed by Equation 2. Here, Q _rr is the reverse recovery charge of the freewheeling diode _DF .
[0016]
[Equation 2]
I _D ≦ 2 · Q _rr / t _d
[0017]
In the present invention, in order to suppress the current distortion by compensating for the voltage error, it introduces a new current command value in a section 2 corresponding to the dead time t _d.
According to the first aspect of the present invention, a new phase current command value is generated so that a phase current equal to or greater than a current amplitude value causing a voltage error from the phase detection current value flows in a leading phase. Then, a deviation between the current command value and the current detection value is obtained, and when the amplitude value of the current command value is equal to or smaller than the current amplitude value at which the voltage error occurs, a gain capable of compensating the maximum error voltage for the deviation is obtained. The multiplied amount is added as the voltage compensation amount to the original voltage command value.
[0018]
That is, the invention according to claim 1 is directed to prevent the occurrence of a current discontinuity phenomenon before and after the phase current zero crossing point including the phase current zero cross point during the dead time provided for preventing the short circuit between the upper and lower arms of the voltage type PWM inverter. In the control device of the voltage type PWM inverter adapted to compensate the voltage command value,
Means for generating a current command value of a leading phase than the current detection value output from each phase of the inverter;
Means for obtaining a deviation between the current command value and the current detection value;
Means for calculating the gain from the dead time and the carrier frequency and the DC intermediate voltage of the inverter;
When the amplitude value of the current command value is equal to or less than the amplitude value of the phase current determined by the reverse recovery charge of the freewheeling diode connected in reverse parallel to the switching element of each arm of the inverter and the dead time, the gain is added to the deviation. Means for adding the voltage compensation amount obtained by multiplying to the original voltage command value of the inverter to compensate the voltage command value;
It is equipped with.
[0019]
The invention according to claim 2 is a means for generating a current command value of a phase that is more advanced than the current detection value output from each phase of the inverter capable of two-phase modulation;
Means for obtaining a deviation between the current command value and the current detection value;
Means for calculating the gain from the dead time and the carrier frequency and the DC intermediate voltage of the inverter;
When the amplitude value of the current command value in a certain phase is equal to or less than the amplitude value of the phase current determined by the reverse recovery charge of the freewheeling diode connected in reverse parallel to the switching element of each arm of the inverter and the dead time, Means for compensating the voltage command value by dividing the voltage compensation amount obtained by multiplying the gain by 2 and adding it to the original voltage command value for the other two modulated phases;
It is equipped with.
[0020]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 is a block diagram showing a first embodiment of the present invention, and corresponds to the embodiment of the invention described in claim 1.
In FIG. 1, a frequency command value ω ₁ ^* is input to a voltage amplitude value calculator 11 and a phase angle calculator 12 and converted into an amplitude command value | V | ^* and a phase angle command value θ ^* , respectively. The amplitude command value | V | ^* and the phase angle command value θ ^* are input to the coordinate converter 13 and input to the adder 21 as a three-phase voltage command value to be output by the PWM inverter 22.
[0021]
Each phase current of the AC motor 30 driven by the PWM inverter 22 is detected by a current detector 23, and these currents are in phase with a voltage command value vector by a coordinate converter 15 to which a phase angle command value θ ^* is input. The component is decomposed into a component and a component orthogonal thereto.
[0022]
On the other hand, the advance angle command value Δθ ^* is added by the adder 16 to the phase angle command value θ ^* output from the phase angle calculator 12, and the addition result is input to the coordinate converter 14.
This coordinate converter 14 is for further advancing the phase angle of the detected current vector output from the preceding coordinate converter 15 by Δθ ^* to obtain a current command value vector for each phase. The current command value vector output from the detector 14 is input to the current amplitude detector 17.
[0023]
When the current amplitude detector 17 detects the current amplitude value of each phase from the current command value vector and determines that the detected current amplitude value is a value that causes a voltage error (the degree to which the above-described Equation 2 is satisfied). When the phase current amplitude value is small), the output is set to “1”, and when it is determined that the voltage error does not occur, the output is set to “0”.
[0024]
The current command value of each phase output from the coordinate converter 14 and the detected current of each phase by the current detector 23 are input to the adder 18 by the sign shown in the figure, and the deviation between the two currents is calculated to the multiplier 19. Have been entered. Further, the DC intermediate voltage E _dc inverter _22, the carrier frequency f _c and the dead time t _d is input to the gain calculator 24, obtained as gain voltage error maximum value epsilon _v ^max by calculation of equation 1 above ing.
[0025]
The gain ε _v ^max and the deviation output from the adder 18 are multiplied by a multiplier 19 to obtain a voltage compensation amount corresponding to the deviation. This voltage compensation amount is input to the adder 21 via the multiplier 20 when it is determined that a voltage error has occurred (when the output of the current amplitude detector 17 is “1”), and the original voltage command value Is added to
[0026]
For this reason, even if the phase current amplitude value becomes small during the dead time and a current stagnation phenomenon may occur, a compensation amount that eliminates the voltage error due to the phase current amplitude value and the reverse recovery charge of the freewheeling diode is set. By compensating for the original voltage command value generated by the adder 18, the gain calculator 24, and the multipliers 19 and 20, the torque pulsation of the AC motor 30 and the fluctuation of the rotational speed caused by the current stagnation phenomenon can be prevented. Can do.
[0027]
Next, a second embodiment of the present invention will be described with reference to FIG. In FIG. 2, the same components as those in FIG. 1 are denoted by the same reference numerals and the description thereof will be omitted.
In this embodiment, two-phase modulation can be selected as the modulation method of the PWM inverter 26 in addition to the three-phase modulation, and the DC intermediate voltage E _dc of the inverter 26 is input to the divider 25 and the output is E _dc. / 2 is input to the gain calculator 24 as described above. Other configurations are the same as those in FIG.
Here, a control device for a PWM inverter that can select three-phase / two-phase modulation is disclosed in, for example, FIG. 4 of Japanese Patent Application No. 2001-46280 by the present applicant.
[0028]
Although not shown in FIG. 2, the PWM inverter 26 includes a three-phase / two-phase modulation selector to which the output of the adder 21 is added, and a voltage command value output from the adder 21 during three-phase modulation. Is used to create PWM signals for all three-phase switching elements of the inverter, and during two-phase modulation, the output voltage of one phase is fixed and only the other two phases are modulated.
[0029]
Here, the two-phase modulation (two-arm modulation) is for the purpose of improving the voltage utilization rate, and the switching element of the upper and lower arms of one phase having a specific section in one cycle is set to all on or all off, and the voltage of the phase And the other two phases are modulated, the voltage waveform of the two phases to be modulated is distorted so that the line voltage becomes a desired waveform (for example, a sine wave), and the line The peak value of the inter-voltage is increased to the DC power supply voltage (see “6. Voltage utilization rate of three-phase PWM inverter” in Chapter 6 “Self-excited inverter” of “Semiconductor power conversion circuit” published by the Institute of Electrical Engineers of Japan. (See “Improved” and “Two-arm modulation” in “Table 6.3.1”).
Further, during normal three-phase modulation, a PWM pulse is generated by comparing each phase voltage command with a carrier to drive the switching element.
[0030]
In the configuration of FIG. 2, the gain calculator 24 outputs a gain (voltage error maximum value ε _v ^max ) during the dead time t _d including the zero cross point of the current, and this gain is input to the multiplier 19 and the same as described above. A voltage compensation amount is obtained. This voltage compensation amount is input to the adder 21 via the multiplier 20 when it is determined that a voltage error has occurred (when the output of the current amplitude detector 17 is “1”), and the original voltage command value Is added to
[0031]
In the PWM inverter 26, when performing two-phase modulation, for example, when the U-phase voltage is fixed and other V-phase and W-phase two phases are modulated, the voltage compensation amount to be added to the U-phase voltage command value Is divided into the V phase and the W phase and superimposed on the voltage command values of the respective phases. For this reason, the voltage compensation amount for these V phase and W phase may be 1/2 of Equation 1. Accordingly, the DC intermediate voltage E _dc input to the gain calculator 24 is halved by the divider 25 to obtain the voltage compensation amount from this gain.
According to this embodiment, it is possible to prevent torque pulsation and fluctuations in the rotational speed of the AC motor 30 due to the current stagnation phenomenon even for the voltage-type PWM inverter of the two-phase modulation system.
[0032]
【The invention's effect】
As described above, according to the present invention, a current command value having a phase that is more advanced than the phase current detection value of the inverter is generated, and the amplitude value is equal to or less than the amplitude value of the phase current determined by the reverse recovery charge and dead time of the freewheeling diode. In some cases, the gain obtained from the dead time, carrier frequency, and DC intermediate voltage is multiplied by the deviation between the current command value and the current detection value to obtain a voltage compensation amount, and this compensation amount is added to the original voltage command value. Is.
[0033]
Therefore, it is possible to provide a low-cost control device that does not use a relatively expensive voltage detector and that can simply perform calculation for compensation, and suppresses torque pulsation and speed fluctuation in the vicinity of the current zero-crossing point. A stable torque can be generated.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a first embodiment of the present invention.
FIG. 2 is a block diagram showing a second embodiment of the present invention.
FIG. 3 is a circuit diagram showing one phase of a voltage-type PWM inverter main circuit.
4 is a diagram showing an equivalent model of the freewheeling diode in FIG. 3. FIG.
5 is an explanatory diagram of an operation mode of a phase current in FIG. 3. FIG.
FIG. 6 is a diagram illustrating the behavior of an arm voltage before and after a dead time.
FIG. 7 is a diagram illustrating the behavior of an arm voltage before and after a dead time.
[Explanation of symbols]
11 Voltage amplitude value calculator 12 Phase angle calculators 13 to 15 Coordinate converters 16, 18, 21 Adder 17 Current amplitude detectors 19, 20 Multipliers 22, 26 Voltage source PWM inverter 23 Current detector 24 Gain calculator 25 Divider 30 AC motor

Claims (2)

Voltage that compensates for the voltage command value of the inverter to prevent the occurrence of current discontinuity before and after the phase current zero crossing point during the dead time provided to prevent the short circuit between the upper and lower arms of the voltage type PWM inverter In the control device of the type PWM inverter,
Means for generating a current command value of a leading phase than the current detection value output from each phase of the inverter;
Means for obtaining a deviation between the current command value and the current detection value;
Means for calculating the gain from the dead time and the carrier frequency and the DC intermediate voltage of the inverter;
When the amplitude value of the current command value is equal to or less than the amplitude value of the phase current determined by the reverse recovery charge of the freewheeling diode connected in reverse parallel to the switching element of each arm of the inverter and the dead time, the gain is added to the deviation. Means for adding the voltage compensation amount obtained by multiplying to the original voltage command value of the inverter to compensate the voltage command value;
A control apparatus for a voltage-type PWM inverter. Voltage that compensates for the voltage command value of the inverter to prevent the occurrence of current discontinuity before and after the phase current zero crossing point during the dead time provided to prevent the short circuit between the upper and lower arms of the voltage type PWM inverter In the control device of the type PWM inverter,
Means for generating a current command value in a phase that is more advanced than the current detection value output from each phase of the inverter capable of two-phase modulation;
Means for obtaining a deviation between the current command value and the current detection value;
Means for calculating the gain from the dead time and the carrier frequency and the DC intermediate voltage of the inverter;
When the amplitude value of the current command value in a certain phase is equal to or less than the amplitude value of the phase current determined by the reverse recovery charge of the freewheeling diode connected in reverse parallel to the switching element of each arm of the inverter and the dead time, Means for compensating the voltage command value by dividing the voltage compensation amount obtained by multiplying the gain by 2 and adding it to the original voltage command value for the other two modulated phases;
A control apparatus for a voltage-type PWM inverter.

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