JP6213318B2 - Current resonance type DC voltage converter, control integrated circuit, and current resonance type DC voltage conversion method - Google Patents

Current resonance type DC voltage converter, control integrated circuit, and current resonance type DC voltage conversion method Download PDF

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JP6213318B2
JP6213318B2 JP2014050634A JP2014050634A JP6213318B2 JP 6213318 B2 JP6213318 B2 JP 6213318B2 JP 2014050634 A JP2014050634 A JP 2014050634A JP 2014050634 A JP2014050634 A JP 2014050634A JP 6213318 B2 JP6213318 B2 JP 6213318B2
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JP2015177595A (en
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西川　武男
武男 西川
貴志 兵頭
貴志 兵頭
都司如 横井
都司如 横井
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Omron Corp
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Priority to PCT/JP2015/054523 priority patent/WO2015137069A1/en
Priority to KR1020167022258A priority patent/KR101865488B1/en
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H02M3/015—Resonant DC/DC converters with means for adaptation of resonance frequency, e.g. by modification of capacitance or inductance of resonance circuit
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0016—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters
    • H02M1/0022—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters the disturbance parameters being input voltage fluctuations
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • H02M7/4818—Resonant converters with means for adaptation of resonance frequency, e.g. by modification of capacitance or inductance of resonance circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33571—Half-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573—Full-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Description

本発明は、直流電圧を別の直流電圧に変換する直流電圧変換器（ＤＣ／ＤＣコンバータ）に関し、特に、絶縁型ＤＣ／ＤＣコンバータの代表的な回路の一つである電流共振型直流電圧変換器、制御用集積回路および電流共振型直流電圧変換方法に関する。   The present invention relates to a DC voltage converter (DC / DC converter) that converts a DC voltage into another DC voltage, and in particular, a current resonance type DC voltage conversion that is one of typical circuits of an insulated DC / DC converter. The present invention relates to a voltage regulator, a control integrated circuit, and a current resonance type DC voltage conversion method.

従来、直流電圧を別の直流電圧に変換するＤＣ／ＤＣコンバータのうちで、特に、絶縁型ＤＣ／ＤＣコンバータの代表的な回路の一つである電流共振型コンバータ（ＬＬＣ方式等）ではソフトスイッチング動作が可能であり、基本的には高効率を実現しやすい回路方式である（例えば特許文献１参照）。   Conventionally, among DC / DC converters that convert a DC voltage into another DC voltage, soft switching is particularly performed in a current resonance type converter (LLC method, etc.) that is one of typical circuits of an isolated DC / DC converter. It is a circuit system that can operate and is basically easy to realize high efficiency (for example, see Patent Document 1).

この特許文献１に記載の電流共振型コンバータは、スイッチング周波数を有する制御信号によってオン／オフ動作するスイッチング素子を有し、前記スイッチング素子によって直流の入力電圧を交流に変換して第１の交流電圧を出力するスイッチング回路と、共振用のインダクタ及びコンデンサを有し、前記第１の交流電圧を入力して所定の共振周波数にて共振して共振信号を出力する共振回路と、前記共振信号を入力する１次巻線、及び前記１次巻線に対して絶縁された２次巻線を有する変圧器と、前記２次巻線から出力される第２の交流電圧を直流に変換して出力電圧を生成し、前記出力電圧を出力側から出力する整流回路と、前記出力電圧を検出し、前記制御信号を生成して前記スイッチング素子をオン／オフ動作させる制御回路とを備え、前記出力電圧に対する前記スイッチング周波数における所定の出力電圧対スイッチング周波数特性を有する電流共振型コンバータにおいて、前記制御回路は、起動時の前記出力電圧を検出してこの検出結果を求める出力電圧検出手段と、前記出力電圧対スイッチング周波数特性及び前記検出結果に基づき、前記出力側に対して電力の供給可能なソフトスタート開始の前記スイッチング周波数を決定する周波数決定手段と、決定された前記スイッチング周波数にて起動を開始させた後に前記スイッチング周波数を漸減する前記制御信号により、前記スイッチング素子をオン／オフ動作させて前記電流共振型コンバータの起動を制御する周波数制御手段と、を有することを特徴とするものである。   The current resonance type converter described in Patent Document 1 includes a switching element that is turned on / off by a control signal having a switching frequency, and converts the DC input voltage to AC by the switching element to generate a first AC voltage. A switching circuit that outputs a resonance signal, an inductor and a capacitor for resonance, a resonance circuit that inputs the first AC voltage, resonates at a predetermined resonance frequency, and outputs a resonance signal; and inputs the resonance signal A transformer having a primary winding and a secondary winding insulated from the primary winding, and an output voltage obtained by converting a second AC voltage output from the secondary winding into a DC A rectifier circuit that outputs the output voltage from an output side, and a control circuit that detects the output voltage and generates the control signal to turn on / off the switching element. In the current resonance type converter having a predetermined output voltage vs. switching frequency characteristic at the switching frequency with respect to the output voltage, the control circuit detects the output voltage at start-up and obtains the detection result. And, based on the output voltage vs. switching frequency characteristics and the detection result, frequency determining means for determining the switching frequency at the start of soft start that can supply power to the output side, and at the determined switching frequency Frequency control means for controlling the activation of the current resonance converter by turning on / off the switching element by the control signal for gradually decreasing the switching frequency after starting the activation. It is.

しかし、この特許文献１に記載されているような電流共振型コンバータでは、太陽光パネルや蓄電池といった入出力電圧が大きく変動する用途に用いる場合、回路の変圧器一次側に並列で存在する共振インダクタンス値を小さくして共振電流を大きくすることで、数倍以上の幅広いゲイン（出力電圧／入力電圧）調整を実現する。しかし、この場合、共振電流による損失が増大し、高効率化が困難であるという課題があった。   However, in the current resonance type converter as described in Patent Document 1, when used for an application in which the input / output voltage fluctuates greatly, such as a solar panel or a storage battery, the resonance inductance existing in parallel on the primary side of the transformer of the circuit. By reducing the value and increasing the resonance current, a wide gain (output voltage / input voltage) adjustment of several times or more is realized. However, in this case, there is a problem that loss due to resonance current increases and it is difficult to achieve high efficiency.

また別の方法として、電圧の変動を直列補償する追加回路を設ける方式が提案されている（例えば、非特許文献１参照）
しかし、この非特許文献１に記載されている双方向絶縁型ＤＣ／ＤＣコンバータでは、スイッチング素子の増加や補償回路での損失増加といった課題があった。 As another method, a method of providing an additional circuit for series compensation of voltage fluctuation has been proposed (for example, see Non-Patent Document 1).
However, the bidirectional insulated DC / DC converter described in Non-Patent Document 1 has problems such as an increase in switching elements and an increase in loss in the compensation circuit.

特開２０１２−０２９４３６号公報JP 2012-029436 A

宮脇慧、他２名、「直列補償方式を用いた双方向絶縁型ＤＣ／ＤＣコンバータの動作検証」、半導体電力変換研究会の講演資料、２０１２年１月２７日、ＳＰＣ−１２-０２５Satoshi Miyawaki and two others, “Verification of Operation of Bidirectional Insulation DC / DC Converter Using Series Compensation Method”, Presentation Material of Semiconductor Power Conversion Study Group, January 27, 2012, SPC-12-025

従来技術のこのような課題に鑑み、本発明の目的は、入出力電圧が大きく変動する用途であっても、スイッチング素子の増加や補償回路などでの損失増加を最小限にしながら高効率を実現可能な電流共振型直流電圧変換器、制御用集積回路および電流共振型直流電圧変換方法を提供することである。   In view of such problems of the prior art, the object of the present invention is to achieve high efficiency while minimizing an increase in switching elements and an increase in loss in a compensation circuit even in applications where the input / output voltage fluctuates greatly. A current resonance type DC voltage converter, a control integrated circuit, and a current resonance type DC voltage conversion method are provided.

上記目的を達成するため、本発明の電流共振型直流電圧変換器は、直流の入力電圧から交流電圧を生成するスイッチング部と、共振用キャパシタンスおよび第１共振用インダクタンスに前記交流電圧が印加されて共振する共振部と、この共振部に１次側が直列接続された変圧器と、この変圧器と並列に存在する第２共振用インダクタンスと、前記変圧器の２次側に現れる電流を整流して直流の出力電圧を生成する整流部と、前記第２共振用インダクタンスを変化させる並列共振インダクタンス調整部とを備えることを特徴とする。   In order to achieve the above object, a current resonance type DC voltage converter according to the present invention includes a switching unit that generates an AC voltage from a DC input voltage, and the AC voltage applied to a resonance capacitance and a first resonance inductance. Rectifying the resonant part that resonates, the transformer whose primary side is connected in series to the resonant part, the second resonance inductance that exists in parallel with the transformer, and the current that appears on the secondary side of the transformer A rectifying unit that generates a DC output voltage and a parallel resonance inductance adjusting unit that changes the second resonance inductance are provided.

ここで、本発明の電流共振型直流電圧変換器は、前記入力電圧もしくは前記出力電圧を検出する電圧検出手段を備え、前記並列共振インダクタンス調整部は、前記変圧器に並列に存在する共振インダクタンス（変圧器の励磁インダクタンスを用いてもよいし、外付けしてもよい）と、前記変圧器の前記１次側に第２スイッチを介して並列接続された調整用インダクタンスを有しており、前記電圧検出手段で検出された前記入力電圧もしくは前記出力電圧に基づいて前記第２スイッチの開閉を制御する第２スイッチ制御部をさらに備えてもよい。また、前記第２スイッチ制御部では、前記第２スイッチの開閉状態遷移の条件となる遷移条件電圧範囲が定められており、前記第２スイッチ制御部は、前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の内であれば前記第２スイッチを開くとともに、前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の内でなければ前記第２スイッチを閉じるように制御してもよい。   Here, the current resonance type DC voltage converter of the present invention includes voltage detection means for detecting the input voltage or the output voltage, and the parallel resonance inductance adjusting unit is configured to have a resonance inductance (in parallel with the transformer) A magnetizing inductance of a transformer may be used or may be externally attached), and an adjustment inductance connected in parallel via a second switch to the primary side of the transformer, You may further provide the 2nd switch control part which controls opening and closing of a said 2nd switch based on the said input voltage or the said output voltage detected by the voltage detection means. In the second switch control unit, a transition condition voltage range that is a condition for the open / close state transition of the second switch is determined, and the second switch control unit determines whether the input voltage or the output voltage is the transition The second switch may be opened if it is within the condition voltage range, and the second switch may be closed if the input voltage or the output voltage is not within the transition condition voltage range.

このような構成の電流共振型直流電圧変換器によれば、最小限の追加回路によって特に定格付近の各種損失が大幅に低減され、これにより効率を向上させることが可能となる。   According to the current resonance type DC voltage converter having such a configuration, various losses particularly in the vicinity of the rating can be greatly reduced by a minimum additional circuit, thereby improving the efficiency.

また、本発明の電流共振型直流電圧変換器において、前記第２スイッチ制御部は、前記第２スイッチが開いているときの前記遷移条件電圧範囲に対して、前記第２スイッチが閉じているときの前記遷移条件電圧範囲を変化させて、前記第２スイッチの開閉制御にヒステリシスを持たせるようにしてもよい。例えば、前記第２スイッチ制御部は、前記第２スイッチが開いているときの前記遷移条件電圧範囲に対して、前記第２スイッチが閉じているときの前記遷移条件電圧範囲をその両端の少なくとも一方で相対的に狭めてもよい。   In the current resonance type DC voltage converter according to the present invention, the second switch control unit may be configured such that the second switch is closed with respect to the transition condition voltage range when the second switch is open. The transition condition voltage range may be changed to provide hysteresis in the opening / closing control of the second switch. For example, the second switch control unit sets the transition condition voltage range when the second switch is closed to at least one of both ends of the transition condition voltage range when the second switch is closed. You may narrow relatively.

このような構成の電流共振型直流電圧変換器によれば、たとえ前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の下限または上限の近傍で細かく変動しても、前記第２スイッチがその度に開閉を繰り返すような現象を極力抑止できる。   According to the current resonance type DC voltage converter having such a configuration, even if the input voltage or the output voltage fluctuates in the vicinity of the lower limit or the upper limit of the transition condition voltage range, the second switch is changed each time. The phenomenon that repeatedly opens and closes can be suppressed as much as possible.

また、本発明の電流共振型直流電圧変換器において、前記第２スイッチは、第１整流手段と、この第１整流手段に直列接続された第１方向スイッチと、第２整流手段と、この第２整流手段に直列接続された第２方向スイッチとを備え、前記第１整流手段の整流方向を第１方向とし、前記第２整流手段の整流方向を第２方向とすると、前記第１方向と前記第２方向とが互いに逆向きとなるように、前記第１整流手段および前記第１方向スイッチが前記第２整流手段および前記第２方向スイッチに並列接続されていてもよい。   Further, in the current resonance type DC voltage converter of the present invention, the second switch includes a first rectifier, a first direction switch connected in series to the first rectifier, a second rectifier, and a second rectifier. A second directional switch connected in series to the two rectifying means, wherein the rectifying direction of the first rectifying means is the first direction, and the rectifying direction of the second rectifying means is the second direction, The first rectifier and the first direction switch may be connected in parallel to the second rectifier and the second direction switch so that the second direction is opposite to each other.

ここで、本発明の電流共振型直流電圧変換器は、前記変圧器の並列共振インダクタンスに流れる並列共振電流値および並列共振電流方向を検出する並列共振電流検出手段をさらに備え、前記第２スイッチ制御部は、前記第１方向スイッチおよび前記第２方向スイッチがいずれも閉じている場合に前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の外から内へと変化すると、前記並列共振電流方向が前記第１方向であれば、まず前記第２方向スイッチを開き、次に前記並列共振電流値が０になった後で前記第１方向スイッチを開くとともに、前記並列共振電流方向が前記第２方向であれば、まず前記第１方向スイッチを開き、次に前記並列共振電流値が０になった後で前記第２方向スイッチを開くように、前記第１方向スイッチおよび前記第２方向スイッチを制御してもよい。さらに、前記第２スイッチ制御部は、前記第１方向スイッチおよび前記第２方向スイッチがいずれも開いている場合に前記入力電圧が前記遷移条件電圧範囲の内から外へと変化すると、前記第１方向スイッチおよび前記第２方向スイッチを同時に閉じるように制御してもよい。   Here, the current resonance type DC voltage converter of the present invention further includes parallel resonance current detection means for detecting a parallel resonance current value and a parallel resonance current direction flowing in the parallel resonance inductance of the transformer, and the second switch control. When the input voltage or the output voltage changes from outside to inside of the transition condition voltage range when both the first direction switch and the second direction switch are closed, the parallel resonance current direction is If it is the first direction, the second direction switch is first opened, and then the first direction switch is opened after the parallel resonant current value becomes 0, and the parallel resonant current direction is the second direction. If the first direction switch and the first direction switch are opened, then the second direction switch is opened after the parallel resonant current value becomes zero. It may control the second direction switch. Further, the second switch control unit may change the first switch when the input voltage changes from the transition condition voltage range to the outside when both the first direction switch and the second direction switch are open. You may control to close a direction switch and the said 2nd direction switch simultaneously.

このような構成の電流共振型直流電圧変換器によれば、前記調整用インダクタに電流が流れていない状態で前記第１方向スイッチおよび前記第２方向スイッチを含む前記第２スイッチを開くため、サージ電圧の発生を抑止できるとともに、さらに前記第１整流手段および前記第２整流手段の逆回復損失も小さくすることができる。   According to the current resonance type DC voltage converter having such a configuration, since the second switch including the first direction switch and the second direction switch is opened in a state where no current flows through the adjustment inductor, a surge is generated. The generation of voltage can be suppressed, and the reverse recovery loss of the first rectifier and the second rectifier can be further reduced.

また、上記の電流共振型直流電圧変換器の前記第２スイッチ制御部を内蔵する制御用集積回路も本発明の範疇である。   A control integrated circuit incorporating the second switch control unit of the current resonance type DC voltage converter is also within the scope of the present invention.

このような構成の制御用集積回路によれば、本発明の電流共振型直流電圧変換器を容易に実現できる。   According to the control integrated circuit having such a configuration, the current resonance type DC voltage converter of the present invention can be easily realized.

あるいは、本発明の電流共振型直流電圧変換方法は、直流の入力電圧から交流電圧を生成するスイッチング工程と、共振用キャパシタンスおよび第１共振用インダクタンスを有する共振部に前記交流電圧が印加されて共振する共振工程と、前記共振部に１次側が直列接続された変圧器で変圧を行う変圧工程と、前記変圧器の２次側に現れる電流を整流して直流の出力電圧を生成する整流工程と、前記変圧器と並列に存在する第２共振用インダクタンスを変化させる共振インダクタンス調整工程とを備えることを特徴とする。   Alternatively, according to the current resonance type DC voltage conversion method of the present invention, the AC voltage is applied to a switching step for generating an AC voltage from a DC input voltage, and the resonance part having the resonance capacitance and the first resonance inductance, and the resonance is performed. A resonating step, a transforming step of transforming with a transformer having a primary side connected in series to the resonating unit, a rectifying step of rectifying a current appearing on the secondary side of the transformer to generate a DC output voltage, And a resonance inductance adjusting step of changing a second resonance inductance existing in parallel with the transformer.

このような構成の電流共振型直流電圧変換方法によれば、最小限の追加回路によって特に定格付近の各種損失が大幅に低減され、これにより効率を向上させることが可能となる。   According to the current resonance type DC voltage conversion method having such a configuration, various losses particularly near the rating can be greatly reduced by a minimum additional circuit, thereby improving the efficiency.

本発明の電流共振型直流電圧変換器および電流共振型直流電圧変換方法によれば、最小限の追加回路によって特に定格付近の各種損失が大幅に低減され、これにより効率を向上させることが可能となる。   According to the current resonance type DC voltage converter and the current resonance type DC voltage conversion method of the present invention, it is possible to greatly reduce various losses particularly near the rating by a minimum additional circuit, thereby improving the efficiency. Become.

本発明の制御用集積回路によれば、本発明の電流共振型直流電圧変換器を容易に実現できる。   According to the control integrated circuit of the present invention, the current resonance type DC voltage converter of the present invention can be easily realized.

本発明の第１実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータ１００の概観構成図である。1 is an overview configuration diagram of a current resonant converter 100 of a parallel resonant inductance adjustment method according to a first embodiment of the present invention. 電流共振型コンバータ１００における本発明のコンセプトを示しており、図２（ａ）は並列共振インダクタンス調整用のスイッチＳＷ５がＯＦＦの場合の説明図であり、図２（ｂ）はスイッチＳＷ５がＯＮの場合の説明図である。The concept of the present invention in the current resonance type converter 100 is shown, FIG. 2A is an explanatory diagram when the switch SW5 for adjusting the parallel resonance inductance is OFF, and FIG. 2B is a diagram when the switch SW5 is ON. It is explanatory drawing in the case. 入力電圧Ｖｉｎに応じた電流共振型コンバータ１００による損失を従来設計との相対比で例示するグラフである。It is a graph which illustrates the loss by the current resonance type converter 100 according to input voltage Vin by relative ratio with the conventional design. 入力電圧Ｖｉｎが３５０Ｖ、出力が４ｋＷのときの電流共振型コンバータ１００による損失詳細の分析結果を従来設計と対比して例示するグラフである。It is a graph which illustrates the analysis result of the loss details by the current resonance type converter 100 when the input voltage Vin is 350 V and the output is 4 kW in comparison with the conventional design. 入力電圧Ｖｉｎが３５０Ｖ、出力が４ｋＷのときの電流共振型コンバータ１００におけるコイルおよびトランスの電流波形を従来設計と対比して例示しており、図５（ａ）が従来設計における各電流波形のグラフであり、図５（ｂ）が電流共振型コンバータ１００における各電流波形のグラフである。The current waveforms of the coil and transformer in the current resonance type converter 100 when the input voltage Vin is 350 V and the output is 4 kW are illustrated in comparison with the conventional design, and FIG. 5A is a graph of each current waveform in the conventional design. FIG. 5B is a graph of each current waveform in the current resonance type converter 100. 本発明の第１実施形態の第１変形例に係る並列共振インダクタンス調整方式の電流共振型コンバータ１００Ａの概観構成図である。It is a general | schematic block diagram of 100 A of current resonance type converters of a parallel resonance inductance adjustment system which concerns on the 1st modification of 1st Embodiment of this invention. 本発明の第１実施形態の第２変形例に係る並列共振インダクタンス調整方式の電流共振型コンバータ１００Ｂの概観構成図である。It is a general | schematic block diagram of the current resonance type converter 100B of the parallel resonance inductance adjustment system which concerns on the 2nd modification of 1st Embodiment of this invention. 本発明の第２実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおけるスイッチＳＷ５のＯＮ／ＯＦＦ制御の概観説明図である。FIG. 7 is an overview explanatory diagram of ON / OFF control of a switch SW5 in a parallel resonance inductance adjustment type current resonance converter according to a second embodiment of the present invention. 本発明の第３実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおける並列共振インダクタンス調整用スイッチＳＷ５Ｃの概観構成図である。It is a general | schematic block diagram of parallel resonance inductance adjustment switch SW5C in the current resonance type | mold converter of the parallel resonance inductance adjustment system which concerns on 3rd Embodiment of this invention. 本発明の第３実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおける並列共振インダクタンス調整用スイッチＳＷ５Ｃの動作シーケンスの概観説明図である。It is a general explanatory view of an operation sequence of the parallel resonance inductance adjustment switch SW5C in the parallel resonance inductance adjustment type current resonance converter according to the third embodiment of the present invention. 本発明の第３実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおいて、並列共振インダクタンス調整用スイッチＳＷ５ＣのＯＮ／ＯＦＦ切り替えにヒステリシスも持たせるようにした場合の状態遷移を示す概観説明図である。Overview explanatory diagram showing state transition in the case of providing hysteresis for ON / OFF switching of the parallel resonance inductance adjustment switch SW5C in the parallel resonance inductance adjustment type current resonance converter according to the third embodiment of the present invention. It is. 本発明の第３実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおいて、並列共振インダクタンス調整用スイッチＳＷ５ＣのＯＮ／ＯＦＦ切り替えにヒステリシスも持たせるようにした場合の動作の概略フローチャートである。14 is a schematic flowchart of an operation when a parallel resonance inductance adjustment switch SW5C is provided with hysteresis in the parallel resonance inductance adjustment switch SW5C in the parallel resonance inductance adjustment type current resonance converter according to the third embodiment of the present invention.

以下、本発明に係る電流共振型直流電圧変換器および制御用集積回路のいくつかの実施形態を、図面を参照して説明する。   Hereinafter, several embodiments of a current resonance type DC voltage converter and a control integrated circuit according to the present invention will be described with reference to the drawings.

＜第１実施形態の概略構成＞
図１は本発明の第１実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータ１００の概観構成図である。 <Schematic configuration of the first embodiment>
FIG. 1 is a schematic configuration diagram of a parallel resonant inductance adjustment type current resonant converter 100 according to a first embodiment of the present invention.

この図１に示すように、電流共振型コンバータ１００は、スイッチング回路１０と、ＬＣ共振回路２０と、トランス３０と、整流回路４０と、並列共振インダクタンス調整回路５０とを備えている。   As shown in FIG. 1, the current resonance type converter 100 includes a switching circuit 10, an LC resonance circuit 20, a transformer 30, a rectifier circuit 40, and a parallel resonance inductance adjustment circuit 50.

スイッチング回路１０は、直流の入力電圧Ｖｉｎが入力される入力端子対１１ａ、１１ｂに、入力平滑用のコンデンサ１２と、直列接続されたスイッチＳＷ１およびスイッチＳＷ２と、直列接続されたスイッチＳＷ３およびスイッチＳＷ４とがそれぞれ並列に接続されている。これらのスイッチＳＷ１〜ＳＷ４によってフルブリッジ型のスイッチング回路が構成されている。   The switching circuit 10 includes an input smoothing capacitor 12, a switch SW1 and a switch SW2 connected in series, a switch SW3 and a switch SW4 connected in series to an input terminal pair 11a and 11b to which a DC input voltage Vin is input. Are connected in parallel. These switches SW1 to SW4 constitute a full bridge type switching circuit.

スイッチＳＷ１〜ＳＷ４としては、例えば電界効果トランジスタ（ＦＥＴ）やＩＧＢＴなどのスイッチング素子が挙げられるが、これらに限らない。   Examples of the switches SW1 to SW4 include, but are not limited to, switching elements such as field effect transistors (FETs) and IGBTs.

このスイッチング回路１０は、スイッチＳＷ１〜ＳＷ４の各状態をスイッチング周波数ｆで所定タイミングで時系列的に切り替えることによって、スイッチＳＷ１およびスイッチＳＷ２間の接続点１３ａと、スイッチＳＷ３およびスイッチＳＷ４間の接続点１３ｂとの間に交流電圧を生成する。なお、スイッチング周波数ｆの制御手段は図示していないが、出力電圧や出力電流を監視してスイッチング周波数ｆの調節を行うものが挙げられる。ただし、これに限らない。この制御手段は、後述する制御ユニット５４で兼用してもよい。   The switching circuit 10 switches each state of the switches SW1 to SW4 in time series at a predetermined timing with the switching frequency f, thereby connecting the connection point 13a between the switch SW1 and the switch SW2 and the connection point between the switch SW3 and the switch SW4. An alternating voltage is generated between 13b and 13b. In addition, although the control means of the switching frequency f is not shown in figure, what adjusts the switching frequency f by monitoring an output voltage or an output current is mentioned. However, it is not limited to this. This control means may be shared by the control unit 54 described later.

ＬＣ共振回路２０は、直列接続された共振用コンデンサ２１（キャパシタンスＣｒ）および共振用コイル２２（インダクタンスＬｒ）を有しており、これらによって第１の共振回路を構成している。スイッチング回路１０によって生成された交流電圧が接続点１３ａ（共振用コンデンサ２１側に接続）および接続点１３ｂ（後述するトランス３０の１次側巻線３１を介して共振用コイル２２側に接続）から印加されることで、ＬＣ共振回路２０は固有の共振周波数ｆｒで共振する。   The LC resonance circuit 20 has a resonance capacitor 21 (capacitance Cr) and a resonance coil 22 (inductance Lr) connected in series, and these constitute a first resonance circuit. The AC voltage generated by the switching circuit 10 is connected from the connection point 13a (connected to the resonance capacitor 21 side) and the connection point 13b (connected to the resonance coil 22 side via the primary winding 31 of the transformer 30 described later). By being applied, the LC resonance circuit 20 resonates at a specific resonance frequency fr.

トランス３０は、１次側巻線３１および２次側巻線３２を有しており、これらの１次側巻線３１および２次側巻線３２は互いに絶縁されている。１次側巻線３１の一端がＬＣ共振回路２０の共振用コイル２２側に直列に接続され、１次側巻線３１の他端がスイッチング回路１０の接続点１３ｂに接続されている。このトランス３０では、１次側巻線３１に印加された１次電圧と巻線比に応じた２次電圧が２次側巻線３２に現れる。   The transformer 30 has a primary side winding 31 and a secondary side winding 32, and the primary side winding 31 and the secondary side winding 32 are insulated from each other. One end of the primary winding 31 is connected in series to the resonance coil 22 side of the LC resonance circuit 20, and the other end of the primary winding 31 is connected to the connection point 13 b of the switching circuit 10. In the transformer 30, a secondary voltage corresponding to the primary voltage applied to the primary winding 31 and the winding ratio appears in the secondary winding 32.

なお、トランス３０の１次側巻線３１には、並列に共振インダクタンス成分を設け（トランスの励磁インダクタンスを利用してもよいし、外付けでもよい）、その並列共振インダクタンスＬｍ１に対応するコイル３３が等価的に存在することになる。ＬＬＣ回路では、前述の共振用コンデンサ２１、共振用コイルおよび並列共振インダクタンスによって第二の共振回路を構成しており、このＬＬＣ共振回路は固有の共振周波数ｆｒ２で共振する。   A primary side winding 31 of the transformer 30 is provided with a resonance inductance component in parallel (excitation inductance of the transformer may be used or may be externally attached), and a coil 33 corresponding to the parallel resonance inductance Lm1. Exist equivalently. In the LLC circuit, the above-described resonance capacitor 21, resonance coil, and parallel resonance inductance constitute a second resonance circuit, and this LLC resonance circuit resonates at a specific resonance frequency fr2.

整流回路４０は、直流の出力電圧Ｖｏｕｔが出力される出力端子対４４ａ、４４ｂに、出力平滑用のコンデンサ４３と、整流方向が一致するように（この図１では上向き）直列接続された整流素子４１ａおよび整流素子４１ｂと、これらの整流方向といずれも一致するように（この図１では上向き）直列接続された整流素子４１ｃおよび整流素子４１ｄとがそれぞれ並列に接続されている。整流素子４１ａ〜４１ｄとしては、例えばダイオードが挙げられるが、これに限らない。   The rectifier circuit 40 is connected in series to an output smoothing capacitor 43 (upward in FIG. 1) in series with an output smoothing capacitor 43 to an output terminal pair 44a, 44b from which a DC output voltage Vout is output. 41a and the rectifying element 41b, and the rectifying element 41c and the rectifying element 41d connected in series are connected in parallel so as to coincide with the rectifying directions (upward in FIG. 1). Examples of the rectifying elements 41a to 41d include diodes, but are not limited thereto.

この整流回路４０の整流素子４１ａおよび整流素子４１ｂ間の接続点４２ａにはトランス３０の２次側巻線３２の一端が接続され、整流素子４１ｃおよび整流素子４１ｄ間の接続点４２ｂにはトランス３０の２次側巻線３２の他端が接続されている。これにより、整流回路４０は出力端子対４４ａ、４４ｂに直流の出力電圧Ｖｏｕｔを生成する。   One end of the secondary winding 32 of the transformer 30 is connected to a connection point 42a between the rectifying element 41a and the rectifying element 41b of the rectifier circuit 40, and the transformer 30 is connected to a connection point 42b between the rectifying element 41c and the rectifying element 41d. The other end of the secondary side winding 32 is connected. As a result, the rectifier circuit 40 generates a DC output voltage Vout at the output terminal pair 44a, 44b.

並列共振インダクタンス調整回路５０は、並列共振インダクタンス調整用コイル５１と、このコイル５１に直列に接続されて電流の通過／遮断を切り替えることで並列共振インダクタンスＬｍ１を実質的に変化させるための並列共振インダクタンス調整用スイッチＳＷ５とを有しており、これらコイル５１およびスイッチＳＷ５が並列共振インダクタンスＬｍ１に対応するコイル３３に並列に接続されている。並列共振インダクタンス調整回路５０はさらに、コイル３３を通過する電流値およびその方向を検出する電流センサ５２と、スイッチング回路１０の入力端子対１１ａ、１１ｂにおける入力電圧Ｖｉｎの電圧値を検出する電圧センサ５３と、電流センサ５２および電圧センサ５３からの検出信号を受けるとともにそれらの検出結果に基づいてスイッチＳＷ５の切り替えを制御する制御ユニット５４とを有している。   The parallel resonance inductance adjustment circuit 50 is connected to the parallel resonance inductance adjustment coil 51 and the parallel resonance inductance Lm1 that is connected in series to the coil 51 and substantially changes the parallel resonance inductance Lm1 by switching between current passage and interruption. The adjustment switch SW5 is included, and the coil 51 and the switch SW5 are connected in parallel to the coil 33 corresponding to the parallel resonance inductance Lm1. The parallel resonant inductance adjustment circuit 50 further includes a current sensor 52 that detects the current value passing through the coil 33 and its direction, and a voltage sensor 53 that detects the voltage value of the input voltage Vin at the input terminal pair 11a and 11b of the switching circuit 10. And a control unit 54 that receives detection signals from the current sensor 52 and the voltage sensor 53 and controls switching of the switch SW5 based on the detection results.

スイッチＳＷ５としては、スイッチＳＷ１〜ＳＷ４と同様に、例えば電界効果トランジスタ（ＦＥＴ）やＩＧＢＴなどのスイッチング素子が挙げられるが、これらに限らない。電圧センサ５３としては、例えば制御ユニット５４がＡ／Ｄ入力端子を有していれば、入力電圧Ｖｉｎを分圧する単純な抵抗分圧回路としてもよい。分圧した電圧を制御ユニット５４のＡ／Ｄ入力端子に接続すれば、抵抗分圧回路の分圧比およびＡ／Ｄ変換結果から制御ユニット５４が入力電圧Ｖｉｎを検出できるからである。ただし、このような電圧検出方法に限らない。また、例えば、入力電圧Ｖｉｎの代わりに出力電圧Ｖｏｕｔを検出するような場所に電圧センサ５３を配置してもよい。   Examples of the switch SW5 include, but are not limited to, switching elements such as field effect transistors (FETs) and IGBTs, like the switches SW1 to SW4. For example, if the control unit 54 has an A / D input terminal, the voltage sensor 53 may be a simple resistance voltage dividing circuit that divides the input voltage Vin. This is because if the divided voltage is connected to the A / D input terminal of the control unit 54, the control unit 54 can detect the input voltage Vin from the voltage dividing ratio of the resistance voltage dividing circuit and the A / D conversion result. However, it is not restricted to such a voltage detection method. Further, for example, the voltage sensor 53 may be arranged at a place where the output voltage Vout is detected instead of the input voltage Vin.

制御ユニット５４によってスイッチＳＷ５がＯＮされると、並列共振インダクタンスＬｍ１に対応するコイル３３と並列にコイル５１が接続されることになるので、実質的な並列共振インダクタンスＬｍは
Ｌｍ１／／Ｌｍ２ ＝ Ｌｍ１×Ｌｍ２／（Ｌｍ１＋Ｌｍ２）
となる。 When the switch SW5 is turned on by the control unit 54, the coil 51 is connected in parallel with the coil 33 corresponding to the parallel resonance inductance Lm1, so that the substantial parallel resonance inductance Lm is Lm1 // Lm2 = Lm1 × Lm2 / (Lm1 + Lm2)
It becomes.

制御ユニット５４によるスイッチＳＷ５の切り替え制御の最も簡単な例としては、スイッチＳＷ５のＯＮ／ＯＦＦ切り替えの条件となる遷移条件電圧範囲を予め定めておいて、電圧センサ５３によって検出された入力電圧Ｖｉｎのみに基づき、この入力電圧Ｖｉｎが遷移条件電圧範囲内であればスイッチＳＷ５をＯＦＦし、遷移条件電圧範囲内でなければスイッチＳＷ５をＯＮするような制御が挙げられるが、このような制御には限らない。   As the simplest example of the switching control of the switch SW5 by the control unit 54, only the input voltage Vin detected by the voltage sensor 53 is determined in advance by setting a transition condition voltage range as a condition for switching the switch SW5 ON / OFF. On the basis of the control, the switch SW5 is turned off when the input voltage Vin is within the transition condition voltage range, and the switch SW5 is turned on when the input voltage Vin is not within the transition condition voltage range. Absent.

図２は電流共振型コンバータ１００における本発明のコンセプトを示しており、図２（ａ）は並列共振インダクタンス調整用のスイッチＳＷ５が開いている場合（ＯＦＦ）の説明図であり、図２（ｂ）はスイッチＳＷ５が閉じている場合（ＯＮ）の説明図である。   FIG. 2 shows the concept of the present invention in the current resonance type converter 100. FIG. 2 (a) is an explanatory view when the switch SW5 for adjusting the parallel resonance inductance is open (OFF), and FIG. ) Is an explanatory diagram when the switch SW5 is closed (ON).

図１を参照して説明したように、第１実施形態では、基本となる並列共振インダクタンスＬｍ１とは別に調整用並列共振インダクタンスＬｍ２を設け、スイッチＳＷ５のＯＮ／ＯＦＦを切り替えることによってこの調整用並列共振インダクタンスＬｍ２の有効／無効を制御可能にする。   As described with reference to FIG. 1, in the first embodiment, the adjustment parallel resonance inductance Lm2 is provided separately from the basic parallel resonance inductance Lm1, and this adjustment parallel is achieved by switching the switch SW5 ON / OFF. Enable / disable control of the resonance inductance Lm2.

調整用並列共振インダクタンスＬｍ２が不要な動作領域（並列共振インダクタンスＬｍ１だけで調整可能な入出力電圧条件）では、図２（ａ）に示すように、スイッチＳＷ５をＯＦＦにして調整用並列共振インダクタンスＬｍ２を無効化することで、実質的な並列共振インダクタンスＬｍを相対的に大きくする。これにより、共振電流が小さくなることで損失も小さくなるので、効率を向上させることができるが、ゲインの調整範囲は狭い。したがって、効率が重視される定格付近では、スイッチＳＷ５をＯＦＦにした状態で使用することが好ましい。   In an operating region where the adjustment parallel resonance inductance Lm2 is not required (input / output voltage conditions that can be adjusted only by the parallel resonance inductance Lm1), as shown in FIG. 2A, the switch SW5 is turned OFF to adjust the adjustment parallel resonance inductance Lm2. By invalidating, the substantial parallel resonance inductance Lm is relatively increased. As a result, since the loss is reduced by reducing the resonance current, the efficiency can be improved, but the gain adjustment range is narrow. Therefore, it is preferable to use the switch SW5 in the OFF state near the rating where efficiency is important.

一方、大きな共振電流が必要となる動作領域（並列共振インダクタンスＬｍ１だけでは調整不可能な入出力電圧条件）では、スイッチＳＷ５をＯＮにして調整用並列共振インダクタンスＬｍ２を有効化することで、実質的な並列共振インダクタンスＬｍを相対的に小さく（Ｌｍ１／／Ｌｍ２）する。これにより、共振電流が大きくなることで損失も大きくなるが、ゲインの調整範囲は広い。したがって、定格付近から大きく外れている場合など、広いゲインの調整範囲が必要な領域では、スイッチＳＷ５をＯＮにした状態で使用することが好ましい。   On the other hand, in an operating region where a large resonance current is required (input / output voltage conditions that cannot be adjusted only with the parallel resonance inductance Lm1), the switch SW5 is turned on to enable the adjustment parallel resonance inductance Lm2. The parallel resonance inductance Lm is made relatively small (Lm1 // Lm2). As a result, the loss increases as the resonance current increases, but the gain adjustment range is wide. Therefore, it is preferable to use the switch SW5 in the ON state in a region where a wide gain adjustment range is required, such as when the value is greatly deviated from the vicinity of the rating.

なお、上述した並列共振インダクタンス調整回路５０では、直列接続されたコイル５１およびスイッチＳＷ５が並列共振インダクタンスＬｍ１に対応するコイル３３に並列に接続されているが、さらに同様の直列接続されたコイルおよびスイッチを並列接続して、各スイッチを組み合わせて制御するようにしてもよい。   In the parallel resonance inductance adjustment circuit 50 described above, the series-connected coil 51 and the switch SW5 are connected in parallel to the coil 33 corresponding to the parallel resonance inductance Lm1, but the same series-connected coil and switch are also used. May be connected in parallel to control each switch in combination.

＜第１実施形態による回路設計例と効果＞
次に、本願発明の効果を確認するため、以下のような条件で回路パラメータ（Ｌｒ，Ｌｍ１、Ｃｒ）の設計を行った。 <Example of Circuit Design and Effects According to First Embodiment>
Next, in order to confirm the effect of the present invention, circuit parameters (Lr, Lm1, Cr) were designed under the following conditions.

１次側電圧： 最小＝２００Ｖ、最大＝５００Ｖ、定格＝３５０Ｖ
２次側電圧： 最小＝３５０Ｖ、最大＝３５０Ｖ、定格＝３５０Ｖ
出力： ４ｋＷ
最大出力電流： １１．４３Ａ
スイッチング周波数ｆ： 最小＝５５ｋＨｚ、最大＝１５０ｋＨｚ
入力電圧Ｖｉｎの全領域が２００〜５００Ｖなので、定格付近の限定された領域を３２０〜４４０Ｖとする条件でＬｒおよびＣｒがほぼ同等となるような解を求めたところ、以下のような解が得られた。 Primary side voltage: Minimum = 200V, Maximum = 500V, Rating = 350V
Secondary voltage: Minimum = 350V, Maximum = 350V, Rating = 350V
Output: 4kW
Maximum output current: 11.43A
Switching frequency f: minimum = 55 kHz, maximum = 150 kHz
Since the entire range of the input voltage Vin is 200 to 500V, when a solution in which Lr and Cr are almost equal under the condition that the limited region near the rating is 320 to 440V is obtained, the following solution is obtained. It was.

全領域（２００〜５００Ｖ）
トランス巻線比 １：１
Ｌｒ＝ ２３．４μＨ
Ｌｍ＝ ４６．８μＨ
Ｃｒ＝１３０．２μＦ
限定領域（３２０〜４４０Ｖ）
トランス巻線比 １：１
Ｌｒ＝ ２４．３μＨ
Ｌｍ＝１４６．０μＨ
Ｃｒ＝１３０．６μＦ
このような設計結果を電流共振型コンバータ１００でほぼ実現するには、以下のように回路パラメータを定めればよい。 All areas (200-500V)
Transformer winding ratio 1: 1
Lr = 23.4 μH
Lm = 46.8 μH
Cr = 130.2μF
Limited area (320-440V)
Transformer winding ratio 1: 1
Lr = 24.3 μH
Lm = 146.0 μH
Cr = 130.6μF
In order to substantially realize such a design result with the current resonance type converter 100, circuit parameters may be determined as follows.

Ｌｒ ＝ ２４μＨ
Ｌｍ１＝１４６μＨ
Ｌｍ２＝ ６９μＨ
Ｃｒ ＝１３０μＦ
また、比較対象として、従来技術による電流共振型コンバータで以下のような回路パラメータのものを用いた。 Lr = 24 μH
Lm1 = 146 μH
Lm2 = 69μH
Cr = 130 μF
For comparison, a current resonance converter according to the prior art having the following circuit parameters was used.

Ｌｒ ＝ ２４μＨ
Ｌｍ１＝ ４６．８μＨ
Ｃｒ ＝１３０μＦ
以上のように定めた回路パラメータを用いた第１実施形態の電流共振型コンバータ１００および従来技術による電流共振型コンバータ（調整用並列共振インダクタンスＬｍ２およびスイッチＳＷ５を設けないもの、出力電圧：３５０Ｖ、出力：４ｋＷ）それぞれについて、回路シミュレーションおよびその分析などを行った結果を次に説明する。 Lr = 24 μH
Lm1 = 46.8 μH
Cr = 130 μF
The current resonant converter 100 of the first embodiment using the circuit parameters determined as described above and the current resonant converter according to the prior art (without the adjusting parallel resonant inductance Lm2 and the switch SW5, output voltage: 350 V, output : 4 kW) The results of circuit simulation and analysis thereof will be described next.

図３は入力電圧Ｖｉｎに応じた電流共振型コンバータ１００による損失を従来設計との相対比で例示するグラフである。ここで、損失とは、スイッチの損失とコイル、トランスの損失（銅損）との合計である。   FIG. 3 is a graph illustrating the loss due to the current resonance type converter 100 according to the input voltage Vin as a relative ratio to the conventional design. Here, the loss is the sum of the loss of the switch and the loss of the coil and transformer (copper loss).

この図３に示すように、入力電圧Ｖｉｎが３２０Ｖ〜４４０Ｖの限定領域（以下では「遷移条件電圧範囲」という）内であればスイッチＳＷ５はＯＦＦされる。例えば、入力電圧Ｖｉｎが３５０Ｖ、４００Ｖでは、損失が従来設計との対比でいずれも６０％近くまで低減していることが確認された。   As shown in FIG. 3, if the input voltage Vin is within a limited region of 320V to 440V (hereinafter referred to as “transition condition voltage range”), the switch SW5 is turned off. For example, when the input voltage Vin is 350V or 400V, it has been confirmed that the loss is reduced to nearly 60% in comparison with the conventional design.

このように、電流共振型コンバータ１００では、入力電圧Ｖｉｎの遷移条件電圧範囲では並列共振インダクタンス調整用スイッチＳＷ５をＯＦＦにし、それ以外ではＯＮにすることで定格付近の効率向上が可能となる。   As described above, in the current resonance type converter 100, the efficiency near the rating can be improved by turning off the parallel resonance inductance adjustment switch SW5 in the transition condition voltage range of the input voltage Vin and turning it on in other cases.

図４は入力電圧Ｖｉｎが３５０Ｖ、出力が４ｋＷのときの電流共振型コンバータ１００による損失詳細の分析結果を従来設計と対比して例示するグラフである。   FIG. 4 is a graph illustrating an analysis result of loss details by the current resonance type converter 100 when the input voltage Vin is 350 V and the output is 4 kW, in comparison with the conventional design.

この図４に示すように、従来設計の場合と比べて、スイッチＳＷ１〜ＳＷ４およびコイル（Ｌｒ、Ｌｍ）における損失が低減していることが確認できる。   As shown in FIG. 4, it can be confirmed that the losses in the switches SW1 to SW4 and the coils (Lr, Lm) are reduced as compared with the conventional design.

図５（ａ）および図５（ｂ）は入力電圧Ｖｉｎが３５０Ｖ、出力が４ｋＷのときの電流共振型コンバータ１００におけるコイルおよびトランスの電流波形を従来設計と対比して例示しており、図５（ａ）が従来設計における各電流波形のグラフであり、図５（ｂ）が電流共振型コンバータ１００における各電流波形のグラフである。   5A and 5B illustrate the current waveforms of the coil and the transformer in the current resonance type converter 100 when the input voltage Vin is 350 V and the output is 4 kW, as compared with the conventional design. FIG. 5A is a graph of each current waveform in the conventional design, and FIG. 5B is a graph of each current waveform in the current resonance type converter 100.

図５（ａ）に示すように、従来設計では、並列共振インダクタンスＬｍ１の値が小さいため、並列共振電流が大きくなっていることがわかる。   As shown in FIG. 5A, in the conventional design, it can be seen that the parallel resonance current is large because the value of the parallel resonance inductance Lm1 is small.

一方、図５（ｂ）に示すように、並列共振インダクタンス調整方式の電流共振型コンバータ１００では実質的な並列共振インダクタンスＬｍ１が大きいため並列共振電流が小さく押さえられている。この結果、本方式の方が損失が低くなったと考えられる。   On the other hand, as shown in FIG. 5B, in the parallel resonance inductance adjustment type current resonance converter 100, the parallel resonance current Lm1 is large, so the parallel resonance current is kept small. As a result, it is considered that the loss was lower in this method.

＜第１実施形態の第１変形例＞
図６は本発明の第１実施形態の第１変形例に係る並列共振インダクタンス調整方式の電流共振型コンバータ１００Ａの概観構成図である。なお、第１実施形態と同一部分については同一の参照符号を付し、その説明の繰り返しは省略する。 <First Modification of First Embodiment>
FIG. 6 is a schematic configuration diagram of a parallel resonance inductance adjustment type current resonance converter 100A according to a first modification of the first embodiment of the present invention. The same parts as those in the first embodiment are denoted by the same reference numerals, and the description thereof will not be repeated.

上述した第１実施形態のスイッチング回路１０では、スイッチＳＷ１〜ＳＷ４によってフルブリッジ型のスイッチング回路が構成されていたが、このような回路に限らない。例えば、図６に示した電流共振型コンバータ１００Ａのように、第１実施形態のスイッチング回路１０をハーフブリッジ型のスイッチング回路１０Ａに置換するとともに、トランス３０の１次側巻線３１の一端（ＬＣ共振回路２０の共振用コイル２２側に接続されていない方）を入力端子１１ｂに接続してもよい。   In the switching circuit 10 according to the first embodiment described above, the full-bridge type switching circuit is configured by the switches SW1 to SW4, but is not limited to such a circuit. For example, like the current resonance type converter 100A shown in FIG. 6, the switching circuit 10 of the first embodiment is replaced with a half-bridge type switching circuit 10A, and one end (LC of the primary winding 31 of the transformer 30 (LC One that is not connected to the resonance coil 22 side of the resonance circuit 20) may be connected to the input terminal 11b.

＜第１実施形態の第２変形例＞
図７は本発明の第１実施形態の第２変形例に係る並列共振インダクタンス調整方式の電流共振型コンバータ１００Ｂの概観構成図である。なお、第１実施形態と同一部分については同一の参照符号を付し、その説明の繰り返しは省略する。 <Second Modification of First Embodiment>
FIG. 7 is a schematic configuration diagram of a parallel resonance inductance adjustment type current resonance converter 100B according to a second modification of the first embodiment of the present invention. The same parts as those in the first embodiment are denoted by the same reference numerals, and the description thereof will not be repeated.

上述した第１実施形態の並列共振インダクタンス調整回路５０では、コイル３３を通過する電流値およびその方向を直接検出する電流センサ５２を用いていたが、このような検出方法に限らない。例えば、図７に示した電流共振型コンバータ１００Ｂのように、第１実施形態の電流センサ５２に代えて、スイッチＳＷ５とコイル３３の間を通過する電流値およびその方向を検出する電流センサ５２ａと、トランス３０の１次側巻線３１を通過する電流値およびその方向を検出する電流センサ５２ｂとを設けてもよい。   In the parallel resonance inductance adjustment circuit 50 of the first embodiment described above, the current sensor 52 that directly detects the current value passing through the coil 33 and the direction thereof is used, but the detection method is not limited thereto. For example, like the current resonance type converter 100B shown in FIG. 7, instead of the current sensor 52 of the first embodiment, a current sensor 52a for detecting a current value passing between the switch SW5 and the coil 33 and its direction, A current sensor 52b for detecting the current value passing through the primary winding 31 of the transformer 30 and its direction may be provided.

ここで、スイッチＳＷ５とコイル３３の間を通過する電流をＩａ、トランス３０の１次側巻線３１を通過する電流をＩｂとすれば、コイル３３を通過する電流値およびその方向はＩａ−Ｉｂから算出できる。なお、このような構成変更により、第１実施形態の制御ユニット５４についても、制御内容を多少変更した制御ユニット５４Ｂに置換する必要がある。   Here, if the current passing between the switch SW5 and the coil 33 is Ia and the current passing through the primary winding 31 of the transformer 30 is Ib, the current value passing through the coil 33 and its direction are Ia-Ib. It can be calculated from Note that it is necessary to replace the control unit 54 according to the first embodiment with a control unit 54B with slightly changed control content due to such a configuration change.

＜第２実施形態＞
図８は本発明の第２実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおけるスイッチＳＷ５のＯＮ／ＯＦＦ制御の概観説明図である。 Second Embodiment
FIG. 8 is an overview explanatory diagram of ON / OFF control of the switch SW5 in the parallel resonance inductance adjustment type current resonance converter according to the second embodiment of the present invention.

上述した第１実施形態の並列共振インダクタンス調整回路５０では、制御ユニット５４は、電圧センサ５３によって検出された入力電圧Ｖｉｎのみに基づき、この入力電圧Ｖｉｎが遷移条件電圧範囲内であればスイッチＳＷ５をＯＦＦし、遷移条件電圧範囲内でなければスイッチＳＷ５をＯＮするように制御する。   In the parallel resonant inductance adjusting circuit 50 of the first embodiment described above, the control unit 54 is based on only the input voltage Vin detected by the voltage sensor 53, and if the input voltage Vin is within the transition condition voltage range, the control unit 54 switches the switch SW5. If it is turned off and not within the transition condition voltage range, the switch SW5 is controlled to be turned on.

しかしながら、例えば太陽光パネルなどに入力側を接続した場合、入力電圧Ｖｉｎは日射量に応じて変動する。もし、遷移条件電圧範囲の上限または下限を挟んで入力電圧Ｖｉｎが細かく変動したりすると、スイッチＳＷ５がその度にＯＮとＯＦＦを繰り返し、いわゆるチャタリングのような現象を生じかねない。   However, for example, when the input side is connected to a solar panel or the like, the input voltage Vin varies according to the amount of solar radiation. If the input voltage Vin fluctuates finely across the upper limit or lower limit of the transition condition voltage range, the switch SW5 repeats ON and OFF each time, and a so-called chattering phenomenon may occur.

そこで、このようなチャタリングを極力抑止するため、いわゆるヒステリシスを持たせるようなスイッチＳＷ５のＯＮ／ＯＦＦ制御が望ましい。具体例として、入力電圧Ｖｉｎの遷移条件電圧範囲が３２０Ｖ〜４４０Ｖの場合について説明する。   Therefore, in order to suppress such chattering as much as possible, ON / OFF control of the switch SW5 that has a so-called hysteresis is desirable. As a specific example, a case where the transition condition voltage range of the input voltage Vin is 320V to 440V will be described.

（１）スイッチＳＷ５がＯＦＦのとき
入力電圧Ｖｉｎが変動してこの遷移条件電圧範囲内から範囲外に変化したときは、すぐにスイッチＳＷ５をＯＦＦからＯＮに切り替える。そのとき、入力電圧Ｖｉｎの遷移条件電圧範囲を元の範囲よりも狭い範囲、例えば、（３２０＋Ｖｈ）Ｖ〜（４４０−Ｖｈ）Ｖに一時的に変更する。ここで、Ｖｈはヒステリシス用電圧である。 (1) When the switch SW5 is OFF When the input voltage Vin fluctuates and changes from within this transition condition voltage range, the switch SW5 is immediately switched from OFF to ON. At that time, the transition condition voltage range of the input voltage Vin is temporarily changed to a range narrower than the original range, for example, (320 + Vh) V to (440−Vh) V. Here, Vh is a hysteresis voltage.

これにより、入力電圧Ｖｉｎが、例えば３２０Ｖをわずかに下回ってスイッチＳＷ５がＯＦＦからＯＮに切り替わった直後に再び３２０Ｖに戻ったとしても、入力電圧Ｖｉｎが（３２０＋Ｖｈ）Ｖには達してないので、スイッチＳＷ５が再びＯＦＦに戻ることなくＯＮ状態が維持される。   As a result, even if the input voltage Vin is slightly lower than, for example, 320 V and the switch SW5 returns to 320 V immediately after switching from OFF to ON, the input voltage Vin does not reach (320 + Vh) V. The ON state is maintained without the SW5 returning to OFF again.

入力電圧Ｖｉｎがさらに大きくなって（３２０＋Ｖｈ）Ｖに達すれば、その時点でスイッチＳＷ５をＯＮからＯＦＦに切り替える。   When the input voltage Vin further increases and reaches (320 + Vh) V, the switch SW5 is switched from ON to OFF at that time.

（２）スイッチＳＷ５がＯＮのとき
上述したように、スイッチＳＷ５をＯＮに切り替えたときは、入力電圧Ｖｉｎの遷移条件電圧範囲を元の範囲よりも狭い範囲、例えば、（３２０＋Ｖｈ）Ｖ〜（４４０−Ｖｈ）Ｖに一時的に変更している。そのため、入力電圧Ｖｉｎが、例えば３２０Ｖ未満から３２０Ｖに達してもスイッチＳＷ５はＯＮ状態が維持されるが、入力電圧Ｖｉｎがさらに大きくなって（３２０＋Ｖｈ）Ｖに達すれば、その時点でスイッチＳＷ５がＯＮからＯＦＦに切り替わる。そのとき、入力電圧Ｖｉｎの遷移条件電圧範囲を元の範囲に戻す。 (2) When the switch SW5 is ON As described above, when the switch SW5 is switched ON, the transition condition voltage range of the input voltage Vin is narrower than the original range, for example, (320 + Vh) V to (440 -Vh) Temporarily changed to V. Therefore, even if the input voltage Vin reaches, for example, less than 320V to 320V, the switch SW5 is maintained in the ON state. However, if the input voltage Vin further increases and reaches (320 + Vh) V, the switch SW5 is turned on at that time. Switches from to OFF. At that time, the transition condition voltage range of the input voltage Vin is returned to the original range.

これにより、入力電圧Ｖｉｎが、例えば（３２０＋Ｖｈ）Ｖをわずかに上回ってスイッチＳＷ５がＯＮからＯＦＦに切り替わった直後に再び（３２０＋Ｖｈ）Ｖに戻ったとしても、入力電圧Ｖｉｎは３２０Ｖには達してないので、スイッチＳＷ５が再びＯＮに戻ることなくＯＦＦ状態が維持される。   As a result, even if the input voltage Vin slightly exceeds (320 + Vh) V, for example, and immediately after the switch SW5 is switched from ON to OFF, it returns to (320 + Vh) V again, the input voltage Vin does not reach 320V. Therefore, the OFF state is maintained without the switch SW5 returning to ON again.

入力電圧Ｖｉｎがさらに小さくなって３２０Ｖ未満になれば、その時点でスイッチＳＷ５がＯＦＦからＯＮに切り替わる。このとき、上述したように、入力電圧Ｖｉｎの遷移条件電圧範囲を元の範囲よりも狭い範囲に再び一時的に変更する。   When the input voltage Vin is further reduced to less than 320 V, the switch SW5 is switched from OFF to ON at that time. At this time, as described above, the transition condition voltage range of the input voltage Vin is temporarily changed again to a range narrower than the original range.

入力電圧Ｖｉｎの遷移条件電圧範囲を一時的に変更する際、必ずしもスイッチＳＷ５がＯＮのときに遷移条件電圧範囲の変更を行わなくてもよい。逆に、スイッチＳＷ５がＯＦＦのときに元の範囲よりも広い範囲に変更してもよい。つまり、スイッチＳＷ５がＯＮのときの遷移条件電圧範囲をＯＦＦのときよりも相対的に狭くすればよい。   When temporarily changing the transition condition voltage range of the input voltage Vin, it is not always necessary to change the transition condition voltage range when the switch SW5 is ON. Conversely, when the switch SW5 is OFF, the range may be changed to a range wider than the original range. That is, the transition condition voltage range when the switch SW5 is ON may be relatively narrower than when it is OFF.

また、入力電圧Ｖｉｎの遷移条件電圧範囲の下限側と上限側を必ずしも同じ電圧幅で変更しなくてもよい。例えば、（３２０＋Ｖｈ１）Ｖ〜（４４０−Ｖｈ２）としてもよい。ここで、Ｖｈ１、Ｖｈ２はいずれもヒステリシス用電圧であって、Ｖｈ１≠Ｖｈ２とする。このような制御とは異なる方法でヒステリシスを持たせるようにしてもかまわない。   Further, the lower limit side and the upper limit side of the transition condition voltage range of the input voltage Vin are not necessarily changed with the same voltage width. For example, it may be (320 + Vh1) V to (440−Vh2). Here, Vh1 and Vh2 are both hysteresis voltages, and Vh1 ≠ Vh2. Hysteresis may be provided by a method different from such control.

また、入力電圧Ｖｉｎの遷移条件電圧範囲の下限側と上限側を同時に変更しなくてもよい。例えば、入力電圧Ｖｉｎが遷移条件電圧範囲の下限側の近傍であれば、遷移条件電圧範囲の下限側のみを一時的に変更すればよい。つまり、入力電圧Ｖｉｎの遷移条件電圧範囲の下限側と上限側の少なくとも一方で変更すればよいのである。   Further, the lower limit side and the upper limit side of the transition condition voltage range of the input voltage Vin may not be changed at the same time. For example, if the input voltage Vin is near the lower limit side of the transition condition voltage range, only the lower limit side of the transition condition voltage range may be temporarily changed. That is, it is only necessary to change at least one of the lower limit side and the upper limit side of the transition condition voltage range of the input voltage Vin.

なお、スイッチＳＷ５のＯＮ／ＯＦＦ前後でＬＬＣ共振回路の共振周波数ｆｒ２が変化するため、スイッチング周波数ｆとゲインの特性は不連続に変化する。この不連続な変化に対応するため、スイッチＳＷ５の切替と合わせてスイッチング周波数ｆを補正するような制御を行うことが望ましい。   Since the resonance frequency fr2 of the LLC resonance circuit changes before and after the switch SW5 is turned on / off, the characteristics of the switching frequency f and the gain change discontinuously. In order to cope with this discontinuous change, it is desirable to perform control to correct the switching frequency f in conjunction with the switching of the switch SW5.

＜第３実施形態＞
上述した第１実施形態および第２実施形態では、並列共振インダクタンス調整回路５０の並列共振インダクタンス調整用スイッチＳＷ５をＯＦＦ（図２（ａ）参照）からＯＮ（図２（ｂ）参照）にする場合は、スイッチＳＷ５と直列にコイル５１（調整用並列共振インダクタンスＬｍ２）が存在するため、電流の変化は緩やかであり、特に問題はない。 <Third Embodiment>
In the first and second embodiments described above, the parallel resonance inductance adjustment switch SW5 of the parallel resonance inductance adjustment circuit 50 is turned from OFF (see FIG. 2A) to ON (see FIG. 2B). Since the coil 51 (adjustment parallel resonance inductance Lm2) exists in series with the switch SW5, the change in current is gradual, and there is no particular problem.

しかし、逆にスイッチＳＷ５をＯＮからＯＦＦにする場合には、コイル５１（調整用並列共振インダクタンスＬｍ２）の電流がそのまま流れ続けようとするため、電流導通時に急にＯＦＦにするとサージ電圧が発生して、スイッチＳＷ５が破損するなどの悪影響もあり得る。   However, conversely, when the switch SW5 is turned from ON to OFF, the current of the coil 51 (adjustment parallel resonance inductance Lm2) continues to flow as it is, so that a surge voltage is generated if the switch is suddenly turned OFF during current conduction. Thus, there may be adverse effects such as breakage of the switch SW5.

そこで、このようなことを極力回避できるように、並列共振インダクタンス調整用スイッチを異なる構成としたものを第３実施形態とし、以下では相違点を主に説明する。   Therefore, in order to avoid such a situation as much as possible, the parallel resonance inductance adjustment switch having a different configuration is referred to as a third embodiment, and the difference will be mainly described below.

図９は本発明の第３実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおける並列共振インダクタンス調整用スイッチＳＷ５Ｃの概観構成図である。なお、第１実施形態と同一部分については同一の参照符号を付し、その説明の繰り返しは省略する。   FIG. 9 is a schematic configuration diagram of the parallel resonance inductance adjustment switch SW5C in the parallel resonance inductance adjustment type current resonance converter according to the third embodiment of the present invention. The same parts as those in the first embodiment are denoted by the same reference numerals, and the description thereof will not be repeated.

この図９に示すように、スイッチＳＷ５Ｃは、整流素子５５ａと、この整流素子５５ａの陰極側に直列に接続されたスイッチＳａと、整流素子５５ｂと、この整流素子５５ｂの陰極側に直列に接続されたスイッチＳｂとを備えている。整流素子５５ａ、５５ｂとしては、例えばダイオードが挙げられるが、これに限らない。   As shown in FIG. 9, the switch SW5C is connected in series to the rectifying element 55a, the switch Sa connected in series to the cathode side of the rectifying element 55a, the rectifying element 55b, and the cathode side of the rectifying element 55b. The switch Sb is provided. Examples of the rectifying elements 55a and 55b include diodes, but are not limited thereto.

そして、直列接続されたスイッチＳａおよび整流素子５５ａが、直列接続された整流素子５５ｂおよびスイッチＳｂに並列に接続されており、このときの整流素子５５ａの整流方向（図９では上向き）と整流素子５５ｂの整流方向（図９では下向き）とは互いに逆になっている。   The switch Sa and the rectifying element 55a connected in series are connected in parallel to the rectifying element 55b and the switch Sb connected in series. At this time, the rectifying direction of the rectifying element 55a (upward in FIG. 9) and the rectifying element The rectification directions of 55b are opposite to each other (downward in FIG. 9).

図１０（ａ）〜（ｅ）は本発明の第３実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおける並列共振インダクタンス調整用スイッチＳＷ５Ｃの動作シーケンスの概観説明図である。   FIGS. 10A to 10E are schematic explanatory diagrams of an operation sequence of the parallel resonance inductance adjustment switch SW5C in the parallel resonance inductance adjustment type current resonance converter according to the third embodiment of the present invention.

図１０（ａ）に示すように、例えば入力電圧Ｖｉｎが遷移条件電圧範囲よりも低くてスイッチＳａおよびスイッチＳｂがいずれもＯＮ状態を想定する。電流センサ５２の検出結果によってＬＣ共振回路２０からコイル５１（調整用並列共振インダクタンスＬｍ２）の方向へ（図１０（ａ）では下向き）に電流が流れていることがわかっているときに、電圧センサ５３の検出結果によって入力電圧Ｖｉｎが高くなって遷移条件電圧範囲内に入ったことがわかったとすると、スイッチＳＷ５ＣをＯＮからＯＦＦに切り替える必要がある。   As shown in FIG. 10A, for example, it is assumed that the input voltage Vin is lower than the transition condition voltage range and both the switch Sa and the switch Sb are in the ON state. When it is known from the detection result of the current sensor 52 that current flows from the LC resonance circuit 20 toward the coil 51 (adjustment parallel resonance inductance Lm2) (downward in FIG. 10A), the voltage sensor If it is found from the detection result 53 that the input voltage Vin has increased and entered the transition condition voltage range, the switch SW5C needs to be switched from ON to OFF.

このとき、整流素子５５ａの整流方向は、コイル５１を流れている電流の向きとは逆なので、この整流素子５５ａに直列接続されているスイッチＳａには電流は流れていない。そこで、図１０（ｂ）に示すように、スイッチＳａをＯＮからＯＦＦに切り替えてもサージ電圧は発生しない。   At this time, since the rectification direction of the rectifying element 55a is opposite to the direction of the current flowing through the coil 51, no current flows through the switch Sa connected in series to the rectifying element 55a. Therefore, as shown in FIG. 10B, no surge voltage is generated even when the switch Sa is switched from ON to OFF.

コイル５１を流れている電流は共振電流であるから、図１０（ｃ）に示すように、徐々に小さくなっていく。そして、共振電流が０となった後、図１０（ｄ）に示すように、電流の向きが反転する。このとき、反転後の電流の向きとは整流素子５５ｂの整流方向が逆なので、この整流素子５５ｂに直列接続されているスイッチＳｂには電流は流れない。   Since the current flowing through the coil 51 is a resonance current, it gradually decreases as shown in FIG. Then, after the resonance current becomes 0, the direction of the current is reversed as shown in FIG. At this time, since the rectifying direction of the rectifying element 55b is opposite to the direction of the current after inversion, no current flows through the switch Sb connected in series to the rectifying element 55b.

そこで、図１０（ｅ）に示すように、スイッチＳｂをＯＮからＯＦＦに切り替えてもサージ電圧は発生しない。これにより、スイッチＳａおよびスイッチＳｂのいずれもＯＦＦとなり、スイッチＳＷ５ＣのＯＮからＯＦＦへの切り替えが完了する。   Therefore, as shown in FIG. 10E, no surge voltage is generated even when the switch Sb is switched from ON to OFF. Thereby, both the switch Sa and the switch Sb are turned off, and the switch from the ON to the OFF of the switch SW5C is completed.

このようなシーケンスでスイッチＳａおよびスイッチＳｂの切り替えを制御することで、コイル５１に電流が流れていない状態でスイッチＳＷ５ＣをＯＮからＯＦＦに切り替えるため、サージ電圧の発生を抑止できるとともに、さらに整流素子５５ａおよび整流素子５５ｂの逆回復損失も小さくすることができる。   By controlling the switching of the switch Sa and the switch Sb in such a sequence, the switch SW5C is switched from ON to OFF in a state where no current flows through the coil 51, so that generation of a surge voltage can be suppressed and further a rectifying element The reverse recovery loss of 55a and rectifying element 55b can also be reduced.

なお、図１０（ａ）の場合と同様に、例えば入力電圧Ｖｉｎが遷移条件電圧範囲よりも低くてスイッチＳａおよびスイッチＳｂがいずれもＯＮ状態ではあるものの、電流センサ５２によって検出された電流の向きが逆であれば、上述した説明のスイッチＳａとスイッチＳｂを読み替えるとともに、整流素子５５ａと整流素子５５ｂも読み替えた上で、図１０（ａ）〜図１０（ｅ）と同様の制御を行えばよい。   As in the case of FIG. 10A, for example, although the input voltage Vin is lower than the transition condition voltage range and both the switch Sa and the switch Sb are in the ON state, the direction of the current detected by the current sensor 52 Is reversed, the switches Sa and Sb described above are replaced, and the rectifier 55a and the rectifier 55b are also replaced, and the same control as in FIGS. 10A to 10E is performed. Good.

図１１は本発明の第３実施形態に係る並列共振インダクタンス調整方式の電流共振型コンバータにおいて、並列共振インダクタンスＬｍ調整用スイッチＳＷ５ＣのＯＮ／ＯＦＦ切り替えにヒステリシスも持たせるようにした場合の状態遷移を示す概観説明図である。図１２はその動作の概略フローチャートである。なお、これらの図に示すような制御を行うには、第１実施形態の制御ユニット５４の構成や制御内容を多少変更する必要がある。例えば、制御ユニットからの制御出力信号を２系統に増やし、スイッチＳａおよびスイッチＳｂを独立してＯＮ／ＯＦＦできる必要がある。それと併せて、制御内容も変更する必要がある。   FIG. 11 shows the state transition in the case where the ON / OFF switching of the parallel resonance inductance Lm adjustment switch SW5C is provided with hysteresis in the parallel resonance inductance adjustment type current resonance converter according to the third embodiment of the present invention. FIG. FIG. 12 is a schematic flowchart of the operation. In order to perform the control shown in these drawings, it is necessary to slightly change the configuration and control contents of the control unit 54 of the first embodiment. For example, it is necessary to increase the control output signal from the control unit to two systems, and to turn ON / OFF the switches Sa and Sb independently. At the same time, it is necessary to change the control contents.

図１２に示すように、まず、スイッチＳａおよびスイッチＳｂをいずれもＯＮに初期化（Ｓ１２０１）した後に、状態Ｂ（Ｓ１２０２、図１１ではＳ１１Ｂに対応）に移行する。   As shown in FIG. 12, first, both the switch Sa and the switch Sb are initialized to ON (S1201), and then the state shifts to the state B (S1202, corresponding to S11B in FIG. 11).

状態Ｂでは、電圧センサ５３によって入力電圧Ｖｉｎを検出し、（３２０＋Ｖｈ）Ｖ〜（４４０−Ｖｈ）Ｖの範囲内であるか否かを判別する（Ｓ１２０３）。判別結果がＮｏであればＳ１２０２に戻る。   In the state B, the input voltage Vin is detected by the voltage sensor 53, and it is determined whether or not it is within the range of (320 + Vh) V to (440−Vh) V (S1203). If a determination result is No, it will return to S1202.

判別結果がＹｅｓであれば、次に電流センサ５２によってコイル５１の電流（以下では「並列共振電流」という）を検出し、その向きを判別する（Ｓ１２０４）。この並列共振電流が図１０（ａ）に示したように下向きであればＳ１２０５に進み、上向きであればＳ１２０８に進む。   If the determination result is Yes, then the current of the coil 51 (hereinafter referred to as “parallel resonance current”) is detected by the current sensor 52, and the direction thereof is determined (S1204). If this parallel resonant current is downward as shown in FIG. 10A, the process proceeds to S1205, and if it is upward, the process proceeds to S1208.

Ｓ１２０５では、図１０（ｂ）に示したように、スイッチＳａをＯＮからＯＦＦに切り替える（図１１ではＳ１１Ｃに対応）。   In S1205, the switch Sa is switched from ON to OFF as shown in FIG. 10B (corresponding to S11C in FIG. 11).

次に、電流センサ５２によって検出される並列共振電流を監視し、この並列共振電流が図１０（ｄ）に示したように０になったかどうかを判別する（Ｓ１２０６）。まだ０になっていなければこのＳ１２０６の処理を繰り返し、０になっていれば、図１０（ｅ）に示したように、スイッチＳｂをＯＮからＯＦＦに切り替えた（Ｓ１２０７）後に、状態Ａ（Ｓ１２１１、図１１ではＳ１１Ａに対応）に移行する。   Next, the parallel resonance current detected by the current sensor 52 is monitored, and it is determined whether or not the parallel resonance current has become 0 as shown in FIG. 10D (S1206). If it is not yet 0, the process of S1206 is repeated. If it is 0, as shown in FIG. 10E, the switch Sb is switched from ON to OFF (S1207), and then the state A (S1211). 11 corresponds to S11A in FIG.

一方、１２０８では、スイッチＳｂをＯＮからＯＦＦに切り替える（図１１ではＳ１１Ｄに対応）。   On the other hand, at 1208, the switch Sb is switched from ON to OFF (corresponding to S11D in FIG. 11).

次に、電流センサ５２によって検出される並列共振電流を監視し、この並列共振電流が０になったかどうかを判別する（Ｓ１２０９）。まだ０になっていなければこのＳ１２０９の処理を繰り返し、０になっていれば、スイッチＳａをＯＮからＯＦＦに切り替えた（Ｓ１２１０）後に、状態Ａ（Ｓ１２１１、図１１ではＳ１１Ａに対応）に移行する。   Next, the parallel resonance current detected by the current sensor 52 is monitored, and it is determined whether or not the parallel resonance current has become 0 (S1209). If it is not yet 0, the processing of S1209 is repeated, and if it is 0, the switch Sa is switched from ON to OFF (S1210), and then the state A (S1211, corresponding to S11A in FIG. 11) is entered. .

状態Ａでは、電圧センサ５３によって入力電圧Ｖｉｎを検出し、３２０Ｖ未満または４４０Ｖより大きいか否かを判別する（Ｓ１２１２）。判別結果がＮｏであればＳ１２１１に戻る。   In the state A, the input voltage Vin is detected by the voltage sensor 53, and it is determined whether it is less than 320V or greater than 440V (S1212). If a determination result is No, it will return to S1211.

判別結果がＹｅｓであれば、スイッチＳａおよびスイッチＳｂをいずれもＯＮに切り替えた（Ｓ１２１３）後に、状態Ｂ（Ｓ１２０２、図１１ではＳ１１Ｂに対応）に移行する。   If the determination result is Yes, the switch Sa and the switch Sb are both turned ON (S1213), and then the state B (S1202, corresponding to S11B in FIG. 11) is entered.

＜その他の実施形態＞
上述した第１実施形態の制御ユニット５４、第１実施形態の第２変形例の制御ユニット５４Ｂ、第３実施形態用の制御ユニットなどを制御用集積回路（ＩＣ）としてもよい。 <Other embodiments>
The control unit 54 of the first embodiment, the control unit 54B of the second modification of the first embodiment, the control unit for the third embodiment, and the like may be used as a control integrated circuit (IC).

特に、第３実施形態用の制御ユニットを制御用集積回路（ＩＣ）としておけば、２つのスイッチを的確なタイミングで複雑に制御する必要がある第３実施形態に係るに並列共振インダクタンス調整方式の電流共振型コンバータを容易に実現することができる。   In particular, if the control unit for the third embodiment is a control integrated circuit (IC), it is necessary to control two switches in a complicated manner at an appropriate timing. A current resonance type converter can be easily realized.

また、上記の各実施形態の制御ユニットは、汎用のＣＰＵなどに適切なプログラムを組み込むことによっても実現できる。   The control unit of each of the above embodiments can also be realized by incorporating an appropriate program into a general-purpose CPU or the like.

なお、本発明は、その主旨または主要な特徴から逸脱することなく、他のいろいろな形で実施することができる。そのため、上述の実施形態はあらゆる点で単なる例示にすぎず、限定的に解釈してはならない。本発明の範囲は特許請求の範囲によって示すものであって、明細書本文にはなんら拘束されない。さらに、特許請求の範囲の均等範囲に属する変形や変更は、全て本発明の範囲内のものである。   It should be noted that the present invention can be implemented in various other forms without departing from the spirit or main features thereof. Therefore, the above-mentioned embodiment is only a mere illustration in all points, and should not be interpreted limitedly. The scope of the present invention is indicated by the claims, and is not restricted by the text of the specification. Further, all modifications and changes belonging to the equivalent scope of the claims are within the scope of the present invention.

本発明は、絶縁型ＤＣ／ＤＣコンバータが用いられるＰＶ（太陽光発電）コンバータ、蓄電池用の充電器、風力発電用コンバータなどだけでなく、絶縁型ＤＣ／ＤＣコンバータ用制御ＩＣなどにも好適である。   The present invention is suitable not only for PV (solar power generation) converters using insulated DC / DC converters, battery chargers, converters for wind power generation, but also for control ICs for insulated DC / DC converters. is there.

１００ 電流共振型コンバータ
１００Ａ 電流共振型コンバータ
１００Ｂ 電流共振型コンバータ
１０ スイッチング回路
１１ａ、１１ｂ 入力端子
１２ コンデンサ
ＳＷ１〜ＳＷ４ スイッチ
１３ａ、１３ｂ 接続点
２０ ＬＣ共振回路
２１ 共振用コンデンサ
２２ 共振用コイル
３０ トランス
３１ １次側巻線
３２ ２次側巻線
３３（Ｌｍ１） コイル
４０ 整流回路
４１ａ〜４１ｄ 整流素子
４２ａ、４２ｂ 接続点
４３ コンデンサ
４４ａ、４４ｂ 出力端子
５０ 並列共振インダクタンス調整回路
５０Ｂ 並列共振インダクタンス調整回路
５１（Ｌｍ２） コイル
ＳＷ５ スイッチ
ＳＷ５Ｃ スイッチ
Ｓａ スイッチ
Ｓｂ スイッチ
５２ 電流センサ
５２ａ、５２ｂ 電流センサ
５３ 電圧センサ
５４ 制御ユニット
５４Ｂ 制御ユニット
５５ａ 整流素子
５５ｂ 整流素子
Ｖｉｎ 入力電圧
Ｖｏｕｔ 出力電圧 100 current resonance converter 100A current resonance converter 100B current resonance converter 10 switching circuit 11a, 11b input terminal 12 capacitor SW1 to SW4 switch 13a, 13b connection point 20 LC resonance circuit 21 resonance capacitor 22 resonance coil 30 transformer 31 1 Secondary winding 32 Secondary winding 33 (Lm1) Coil 40 Rectifier circuits 41a to 41d Rectifier elements 42a and 42b Connection point 43 Capacitors 44a and 44b Output terminal 50 Parallel resonance inductance adjustment circuit 50B Parallel resonance inductance adjustment circuit 51 (Lm2) ) Coil SW5 Switch SW5C Switch Sa Switch Sb Switch 52 Current sensor 52a, 52b Current sensor 53 Voltage sensor 54 Control unit 54B Control unit 55a Rectifier element 55b Rectifier element Vi n Input voltage Vout Output voltage

Claims (6)

直流の入力電圧から交流電圧を生成するスイッチング部と、
共振用キャパシタンスおよび第１共振用インダクタンスに前記交流電圧が印加されて共振する共振部と、
この共振部に１次側が直列接続された変圧器と、
この変圧器と並列に存在する第２共振用インダクタンスと、
前記変圧器の２次側に現れる電流を整流して直流の出力電圧を生成する整流部と、
前記変圧器の前記１次側に第２スイッチを介して並列接続された調整用インダクタンスを有し、前記第２共振用インダクタンスを変化させる並列共振インダクタンス調整部と、
前記入力電圧もしくは前記出力電圧を検出する電圧検出手段と、
この電圧検出手段で検出された前記入力電圧もしくは前記出力電圧に基づいて前記第２スイッチの開閉を制御する第２スイッチ制御部と
を備え、
前記第２スイッチ制御部では、前記第２スイッチの開閉状態遷移の条件となる遷移条件電圧範囲が定められており、前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の内であれば前記第２スイッチを開くとともに、前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の内でなければ前記第２スイッチを閉じるように制御し、さらに、前記第２スイッチが開いているときの前記遷移条件電圧範囲に対して、前記第２スイッチが閉じているときの前記遷移条件電圧範囲を変化させて、前記第２スイッチの開閉制御にヒステリシスを持たせることを特徴とする電流共振型直流電圧変換器。 A switching unit that generates an AC voltage from a DC input voltage;
A resonance unit that resonates when the AC voltage is applied to the resonance capacitance and the first resonance inductance;
A transformer having a primary side connected in series to the resonating unit;
A second resonance inductance present in parallel with the transformer;
A rectifying unit that rectifies a current appearing on the secondary side of the transformer to generate a DC output voltage;
A parallel resonance inductance adjustment unit having an adjustment inductance connected in parallel to the primary side of the transformer via a second switch, and changing the second resonance inductance ;
Voltage detection means for detecting the input voltage or the output voltage;
A second switch control section for controlling opening and closing of the second switch based on the input voltage or the output voltage detected by the voltage detection means;
Equipped with a,
In the second switch control unit, a transition condition voltage range that is a condition for the open / close state transition of the second switch is determined, and if the input voltage or the output voltage is within the transition condition voltage range, the second switch controller 2 switch is opened, the second switch is controlled to be closed if the input voltage or the output voltage is not within the transition condition voltage range, and the transition condition when the second switch is opened the voltage range, the second and the changing the transition condition voltage range when the switch is closed, the opening and closing control a current resonance type DC voltage conversion, wherein Rukoto a hysteresis of the second switch vessel. 請求項１に記載の電流共振型直流電圧変換器において、
前記第２スイッチ制御部は、
前記第２スイッチが開いているときの前記遷移条件電圧範囲に対して、前記第２スイッチが閉じているときの前記遷移条件電圧範囲をその両端の少なくとも一方で相対的に狭めることを特徴とする電流共振型直流電圧変換器。 The current resonance type DC voltage converter according to claim 1 ,
The second switch controller is
The transition condition voltage range when the second switch is closed relative to the transition condition voltage range when the second switch is open is relatively narrowed at least at one of both ends thereof. Current resonance type DC voltage converter. 請求項１または２に記載の電流共振型直流電圧変換器において、
前記第２スイッチは、
第１整流手段と、
この第１整流手段に直列接続された第１方向スイッチと、
第２整流手段と、
この第２整流手段に直列接続された第２方向スイッチと
を備え、
前記第１整流手段の整流方向を第１方向とし、前記第２整流手段の整流方向を第２方向とすると、前記第１方向と前記第２方向とが互いに逆向きとなるように、前記第１整流手段および前記第１方向スイッチが前記第２整流手段および前記第２方向スイッチに並列接続されていることを特徴とする電流共振型直流電圧変換器。 In the current resonance type DC voltage converter according to claim 1 or 2 ,
The second switch is
First rectifying means;
A first direction switch connected in series to the first rectifying means;
A second rectifying means;
A second direction switch connected in series to the second rectifying means,
When the rectifying direction of the first rectifying means is the first direction and the rectifying direction of the second rectifying means is the second direction, the first direction and the second direction are opposite to each other. A current resonance type DC voltage converter, wherein one rectifying means and the first directional switch are connected in parallel to the second rectifying means and the second directional switch. 請求項３に記載の電流共振型直流電圧変換器において、
前記変圧器の並列共振インダクタンスに流れる並列共振電流値および並列共振電流方向を検出する並列共振電流検出手段をさらに備え、
前記第２スイッチ制御部は、
前記第１方向スイッチおよび前記第２方向スイッチがいずれも閉じている場合に
前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の外から内へと変化すると、
前記並列共振電流方向が前記第１方向であれば、まず前記第２方向スイッチを開き、次に前記並列共振電流値が０になった後で前記第１方向スイッチを開くとともに、
前記並列共振電流方向が前記第２方向であれば、まず前記第１方向スイッチを開き、次に前記並列共振電流値が０になった後で前記第２方向スイッチを開くように、前記第１方向スイッチおよび前記第２方向スイッチを制御することを特徴とする電流共振型直流電圧変換器。 In the current resonance type DC voltage converter according to claim 3 ,
A parallel resonance current detecting means for detecting a parallel resonance current value flowing in a parallel resonance inductance of the transformer and a parallel resonance current direction;
The second switch controller is
When both the first direction switch and the second direction switch are closed, the input voltage or the output voltage changes from outside to inside the transition condition voltage range,
If the parallel resonant current direction is the first direction, first open the second direction switch, then open the first direction switch after the parallel resonant current value becomes 0,
If the parallel resonance current direction is the second direction, the first direction switch is first opened, and then the second direction switch is opened after the parallel resonance current value becomes zero. A current resonance type DC voltage converter for controlling a direction switch and the second direction switch. 請求項４に記載の電流共振型直流電圧変換器の前記第２スイッチ制御部を内蔵する制御用集積回路。 5. A control integrated circuit including the second switch control unit of the current resonance type DC voltage converter according to claim 4 . 直流の入力電圧から交流電圧を生成するスイッチング工程と、
共振用キャパシタンスおよび第１共振用インダクタンスを有する共振部に前記交流電圧が印加されて共振する共振工程と、
前記共振部に１次側が直列接続された変圧器で変圧を行う変圧工程と、
前記変圧器の２次側に現れる電流を整流して直流の出力電圧を生成する整流工程と、
前記変圧器と並列に存在する第２共振用インダクタンスを変化させる共振インダクタンス調整工程と、
前記入力電圧もしくは前記出力電圧を検出する電圧検出工程と、
この電圧検出工程で検出された前記入力電圧もしくは前記出力電圧に基づいて、前記変圧器の前記１次側への調整用インダクタンスの並列接続を介する第２スイッチの開閉を制御する第２スイッチ制御工程と
を備え、
前記第２スイッチ制御工程では、前記第２スイッチの開閉状態遷移の条件となる遷移条件電圧範囲が定められており、前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の内であれば前記第２スイッチを開くとともに、前記入力電圧もしくは前記出力電圧が前記遷移条件電圧範囲の内でなければ前記第２スイッチを閉じるように制御し、さらに、前記第２スイッチが開いているときの前記遷移条件電圧範囲に対して、前記第２スイッチが閉じているときの前記遷移条件電圧範囲を変化させて、前記第２スイッチの開閉制御にヒステリシスを持たせることを特徴とする電流共振型直流電圧変換方法。 A switching process for generating an AC voltage from a DC input voltage;
A resonance process in which the AC voltage is applied to a resonance part having a resonance capacitance and a first resonance inductance to resonate;
A transformation step of performing transformation with a transformer in which a primary side is connected in series to the resonance unit;
A rectifying step of rectifying a current appearing on the secondary side of the transformer to generate a DC output voltage;
A resonance inductance adjusting step of changing a second resonance inductance existing in parallel with the transformer ;
A voltage detection step of detecting the input voltage or the output voltage;
A second switch control step for controlling the opening and closing of the second switch through a parallel connection of the adjustment inductance to the primary side of the transformer based on the input voltage or the output voltage detected in the voltage detection step and comprising a <br/>,
In the second switch control step, a transition condition voltage range as a condition for switching the open / close state of the second switch is determined, and if the input voltage or the output voltage is within the transition condition voltage range, 2 switch is opened, the second switch is controlled to be closed if the input voltage or the output voltage is not within the transition condition voltage range, and the transition condition when the second switch is opened the voltage range, the second and the changing the transition condition voltage range when the switch is closed, the opening and closing control a current resonance type DC voltage conversion, wherein Rukoto a hysteresis of the second switch Method.
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EP15761619.4A EP3118983B1 (en) 2014-03-13 2015-02-19 Current resonant type dc voltage converter, control integrated circuit, and current resonant type dc voltage conversion method
PCT/JP2015/054523 WO2015137069A1 (en) 2014-03-13 2015-02-19 Current resonant type dc voltage converter, control integrated circuit, and current resonant type dc voltage conversion method
KR1020167022258A KR101865488B1 (en) 2014-03-13 2015-02-19 Current resonant type dc voltage converter, control integrated circuit, and current resonant type dc voltage conversion method
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EP3118983A4 (en) 2017-11-22
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US20170054379A1 (en) 2017-02-23
KR20160107323A (en) 2016-09-13
CN105981279A (en) 2016-09-28
EP3118983B1 (en) 2019-04-10
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US10079545B2 (en) 2018-09-18
WO2015137069A1 (en) 2015-09-17

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