JP4657062B2 - Resonant type converter - Google Patents

Description

  The present invention relates to a resonant converter used in a switching power supply device and a control method thereof.

  In the DC-DC converter, power loss occurs due to the overlap of noise, current and voltage when the switching element is turned on and off. As a means for improving this problem, various resonant converters have been studied. A typical resonance type converter is a series resonance type converter. The original form of this converter is well known as shown in FIG. When this is operated under certain conditions, the above problems can be improved. However, when the output is controlled under the conditions of input voltage fluctuation and load fluctuation, improvement is difficult at all operating points.

  FIG. 11 is an example of operation waveforms when the input voltage of the circuit shown in FIG. 10 is high and the output voltage is low. 11, a is the current of the resonance circuit, b is the current of D1 on the high side, c is the total current of the low side capacitor C2, diode D2, and the second switch element Q2, and d is the current of the second switch element Q2. Voltage. C1 and D1 are parasitic elements of the first switch element Q1, and C2 and D2 are parasitic elements of the second switch element Q2.

  In FIG. 11, at time t0, the switch element Q2 is turned on, the first switch element Q1 is turned off, and the resonance current ir increases in the direction in which the capacitor Cr is discharged (opposite to the arrow), the capacitor Cr and the inductor Lr. began to decrease resonant current ir by resonance of, becomes zero at t1, the discharge of the capacitor Cr is continued, the resonant current ir increases reversed (arrow) direction. At this time, if the ON voltage of the second switch element Q2 is higher than the forward voltage of the diode D2, the diode D2 becomes conductive. At t2, the first switch element Q1 is turned on, and the second switch element Q2 is turned off. At this moment, the reverse recovery current of the diode D2 and the charging current of the capacitor C2 flow, the switching loss is large, and noise is generated. In a so-called lead phase, zero volt switching (hereinafter referred to as “ZVS”) is also difficult. At t3, the next cycle starts again, but a similar problem occurs at this time.

  Many proposals have been made for this countermeasure (see Patent Document 1 as an example). There is a capacitor voltage clamped series resonant converter. FIG. 12 shows an example thereof, and FIG. 13 shows an example of operation waveforms. The operation will be described with reference to FIGS. In FIG. 12, capacitors C1 and C2 and diodes D1 and D2 indicate parasitic elements of the switch elements Q1 and Q2. In FIG. 13, a is the current of the inductor Lr, b is the voltage of the capacitor Cr, c is the total current of the capacitor C1, the diode D1, and the switch element Q1, and d is the voltage of the switch element Q1.

  The switch element Q1 is turned off at t0. Charging of the capacitor C1 by transformer magnetizing current, the discharge of the capacitor C2 is carried out in a short time, the voltage of the switching element Q1 and the voltage of the switching element Q2 to the input voltage becomes zero volts. At that time, the switching element Q2 is turned on to obtain ZVS. Thereafter, due to resonance between the capacitor Cr and the inductor Lr, the capacitor Cr is discharged to zero volts at t1, and the diode D4 becomes conductive. Since the voltage of the inductor Lr is fixed to a voltage determined by the output voltage and the turns ratio of the transformer, the current of the inductor Lr decreases linearly and becomes zero at t2. Since the voltage of the capacitor Cr is clamped by the diode D4, the current of the inductor Lr does not flow on the negative side and maintains zero after t2, so that no current flows through the diode D2. When the switch element Q2 is turned off at t3, the capacitor C2 is charged and the capacitor C1 is discharged in a short time by the exciting current of the transformer, and the voltage of the switch element Q1 becomes zero volts. Immediately after that, ZVS is obtained by turning on the switching element Q1. The operation is the same until t4, and the next cycle is started. Thus, since no current flows through the body diode when the switching elements Q1 and Q2 are switched, there is no switching loss due to the reverse recovery current of the body diode, and ZVS is easy.

  By the way, in the series resonance type converter, a resonance current is output every cycle. In FIG. 12, the charging / discharging current of the capacitor Cr is determined by the input voltage, the output voltage, the inductor Lr, and the capacitor Cr. For example, when the output voltage is controlled to a constant value, the frequency is greatly reduced when the load is light. The fluctuation range of the operating frequency with respect to the fluctuation of the input voltage and the load becomes very large. Large fluctuations in operating frequency cause problems such as increasing the size of the transformer and deteriorating the frequency characteristics of the output.

As an example of a measure for reducing the fluctuation of the operating frequency, there is one in which a parallel resonant circuit is connected in series with a series resonant circuit, as introduced in Patent Document 1. An example is shown in FIG. FIG. 15 shows the principle of reducing the fluctuation of the operating frequency. The parallel circuit of the resonant inductor Lp and the resonant capacitor Cp in FIG. 14 is a parallel resonant circuit and is connected in series with the inductor Lr. Others are the same as FIG. The resonance frequency fp of the parallel resonance circuit is set lower than the series resonance frequency fr composed of the inductor Lr and the capacitor Cr. When the operating frequency is close to the series resonance frequency fr, the current in the parallel resonance circuit is small, and the current in the series resonance circuit flows through the resonance capacitor Cp, and the operation is almost the same as in FIG. When the operating frequency becomes substantially equal to the resonance frequency fp of the parallel resonance circuit, the voltage of the parallel resonance circuit increases rapidly, the voltage applied to the series resonance circuit is limited, and the current of the series resonance circuit decreases. That is, the power per cycle decreases and the output decreases. Since the output power is highest when the operating frequency is equal to the series resonant frequency fr, the output can be controlled by changing the operating frequency between the series resonant frequency fr and the resonant frequency fp of the parallel resonant circuit.
JP-A-3-103070

  However, when the operating frequency approaches the resonant frequency fp of the parallel resonant circuit, the current in the parallel resonant circuit increases rapidly. In addition, since the parallel resonant frequency is set low, the inductance and capacitance of the resonant inductor Lp and the resonant capacitor Cp are increased, which overlaps with the large maximum passing current, and the resonant inductor Lp and the resonant capacitor Cp have a large volume. Become. In addition, the current of the parallel resonant circuit increases at light loads, which increases the loss and decreases the efficiency.

  The present invention has been made in view of the above-mentioned problems. While taking advantage of low loss and low noise, which are inherent advantages of a resonant converter, the present invention has been made to maintain a lagging phase during control, and there is little variation in operating frequency. The series resonance converter with a small current is provided, which contributes to the miniaturization and high performance of the power supply device.

  In order to solve the above-described problems, a resonant converter according to the present invention includes a first series circuit of a first switch element and a second switch element connected to a DC power source, and one junction point is the first switch circuit. Consists of a first series resonant circuit consisting of a first inductor and a third capacitor, and a second series resonant circuit consisting of a second inductor and a fourth capacitor, connected to the connection point of the series circuit. A diode bridge that rectifies the current of the two series resonant circuits, the parallel circuit, one of the AC terminals is connected to the other junction of the parallel circuit, the other of the AC terminals is connected to one end of the DC power supply, The fifth capacitor for smoothing the output of the diode bridge, and the first switch element and the second switch element are alternately turned on and off alternately for a short period of time when both are turned off simultaneously. A circuit for controlling the voltage of the fifth capacitor by changing the operating frequency, and setting the resonance frequency of the first series resonance circuit and the resonance frequency of the second series resonance circuit to be different values. It is characterized by that.

  A first series circuit of a first switch element and a second switch element connected to a DC power source, and first and second primary terminals having one terminal connected to one terminal of the DC power source A transformer provided with a coil and a secondary coil, and a first inductor and a third capacitor connected between a connection point of the first series circuit and the other terminal of the first primary coil. A first series resonant circuit; and a second series resonant circuit comprising a second inductor and a fourth capacitor connected between a connection point of the first series circuit and the other terminal of the second primary coil. A diode bridge that rectifies the current flowing through the secondary coil, the fifth capacitor that smoothes the output of the diode bridge, the first switch element, and the second switch element. Both at the same time And a circuit for controlling the voltage of the fifth capacitor by changing the operating frequency alternately and turning on and off over a short period of time, the resonance frequency of the first series resonance circuit, The resonance frequency of the two series resonance circuits is set to have different values.

The transformer is characterized in that first and second primary coils are wound around a monopod of a bipod or tripod core, and a secondary coil is wound around the monopod or other monopod.
The first switch element is connected in parallel with the first diode and the first capacitor, respectively, and the second switch element is connected with the second diode and the second capacitor, respectively. And a second parallel circuit connected in parallel.
Further, the first and second inductors are characterized in that the leakage inductance of the transformer is part or all of the transformer.
The first and / or second diode and the first and / or second capacitor may be included as parasitic elements of the first and / or second switch element.

  According to the present invention, by controlling the operating frequency fs between the resonance frequency fr1 of the first series resonance circuit and the resonance frequency fr2 of the second series resonance circuit, an output is obtained with respect to input and load fluctuations. Control such as keeping constant is possible, and the frequency fluctuation range can be narrowed. In particular, as will be described later, the resonance current is added by controlling between the frequency fv and fr1 and fr2, which are near the midpoint between fr1 and fr2, and the lower resonance frequency of fr1 and fr2. The phase is delayed with respect to the switching cycle, and the ZVS of the switching element is facilitated.

  FIG. 1 shows the best mode for carrying out the present invention. The resonant converter shown in FIG. 1 is a non-insulated resonant converter, and a first switch element SW1 and a second switch element SW2 are connected in series to form a first series circuit. Connected to Vi. The first switch element SW1 is connected in parallel with the first diode D1 and the first capacitor C1, respectively, and the second switch element SW2 is connected in parallel with the second diode D2 and the second capacitor C2, respectively. is there.

  The first inductor L1 and the third capacitor C3 constitute a first series resonance circuit, and the second inductor L2 and the fourth capacitor C4 constitute a second series resonance circuit, which are connected in parallel. One junction of the parallel circuit is connected to a connection point between the first switch element SW1 and the second switch element SW2. The resonant converter according to the present embodiment includes diode bridges D3, D4, D5, and D6, one of the AC terminals of the diode bridges D3 to D6 is connected to the other junction of the parallel circuit, and the AC terminal. Is connected to one end of the DC power source Vi. The DC terminals of the diode bridges D3 to D6 are connected to a fifth capacitor C5 that smoothes the outputs of the diode bridges D3 to D6.

  The resonant converter according to the present embodiment alternately turns on and off the first switch element SW1 and the second switch element SW2 with a short period during which both are turned off at the same time, and changes the operating frequency to change the fifth. A circuit for controlling the voltage of the capacitor. The resonance frequency of the first series resonance circuits L1 and C3 is fr1, the resonance frequency of the second series resonance circuits L2 and C4 is fr2, and fr1 and fr2 are set to different values such that fr1 <fr2. is there. Note that fr1 ÷ fr2 = 1.3-3 is preferable.

  The resonant converter configured as described above operates as follows. First, since the two series resonant circuits are in parallel, the sum of the currents of these circuits is output from the input. When the operating frequency is fs, when fs = fr1, the currents of the first series resonant circuits L1 and C3 are significantly larger than the currents of the second series resonant circuits L2 and C4, and the output is the first series resonant circuit. The characteristics of the circuits L1 and C3 dominate. In the case of fs = fr2, on the contrary, the characteristics of the second series resonance circuits L2 and C4 dominate. When the operating frequency is changed between fr1 and fr2, the output changes depending on the current amplitude and phase difference of the first and second series resonant circuits. For example, the sum of the currents of the two series resonant circuits if substantially equal phase difference amplitude within 180 ° in order to approach zero, it is possible to reduce the output substantially zero. According to this principle, it is possible to perform control such as keeping the output voltage constant against input fluctuation and load fluctuation.

  Although not shown in FIG. 1, showing detecting an output voltage, a waveform example in which the output voltage is controlled by the control circuit unit for controlling the frequency to be constant in Figure 2, Figure 3. Figure 2 is a low input voltage, if the load current is large, the operating frequency fs is operating in a value slightly higher than the resonance frequency fr1 of the first series resonant circuit L1, C3. 2a is the current of the first inductor L1, b is the current of the second inductor L2, c is the input current of the diode bridges D3 to D6, d is the total current of the first parallel circuits SW1, D1 and C1, e shows the voltage waveform of the first switch element SW1.

  Since the amplitude of b is very small compared to FIG. 2 shown a, c in a state approximately equal to a, the output is dominated by the first series resonant circuit L1, C3. At t0, the first switch element SW1 is turned off. The current of the resonance circuit is in a lagging phase with respect to the switching timing, and this current charges the first capacitor C1 and discharges the second capacitor C2. Discharge of the second capacitor C2 is completed at t1, after the voltage becomes zero, the ZVS by turning on the second switching element SW2. t2 in the resonance current to turn off the second switching element SW2 is charged in the second capacitor C2, spans, to discharge the first capacitor C1. Immediately after the voltage of the first capacitor C1 becomes zero, the first switch element SW1 is turned on to obtain ZVS. Transition to the next cycle at t4.

  FIG. 3 shows a waveform when the input voltage is high and the load current is 10% of the rating. a to e show waveforms at the same site as in FIG. The operating frequency fs becomes higher and the time axis becomes finer. Since the operating frequency fs changes in a direction higher than the resonance frequency fr, a has a smaller amplitude and a larger phase delay than the state of FIG. b is the operating frequency fs is some amplitude so becomes closer to the resonance frequency fr2 of the second series resonant circuit L2, C4 is increased, but still leading phase. Approaches the amplitude of a and b, the phase difference is reduced c is the sum of the increases and a and b toward 180 °, which corresponds to a decrease in the output current. c is in the lag phase, and ZVS is the same as in FIG.

  FIG. 4 shows the output voltage when the input / output conditions are fixed and the operating frequency fs is changed. The operating frequency fs changes from the resonant frequency fr1 of the first series resonant circuits L1 and C3 to the high frequency side. go to the output voltage decreases, started to increase from a certain point, indicating that also become maximum at the resonance frequency fr2 of the second series resonant circuit L2, C4. If the frequency at which the output voltage becomes a valley point is fv, control is possible on the low frequency side and high frequency side where fs is fv, but on the high frequency side, current c advances and is not preferable in terms of ZVS. Therefore, control is performed in the region of fr1 <fs <fv.

  In this way, by controlling the operating frequency fs between the first resonance frequency fr1 and the frequency fv at which the output voltage becomes a trough point, it is possible to control the output to be constant with respect to input and load fluctuations. The frequency fluctuation range can be narrowed. Returning to FIG. 3, FIG. 3 shows a state in which the operating frequency fs approaches the frequency fv at which the output voltage becomes a trough point, but the amplitude of a decreases and the amplitude of b is still small. Therefore, the problem can be solved without increasing the size of a transformer, a capacitor and the like as in the embodiment according to the existing technology.

  The total current in the resonant circuit by controlling at fs <fv is in phase lag, ZVS is easy reduction of switching loss, it is possible to reduce the noise. In addition, as can be seen in FIG. 4, the output voltage can be reduced to near zero under the condition that the operating frequency fs is substantially equal to the frequency fv at which the output voltage becomes a trough point. It is possible to protect up to.

  As described above, according to the present invention, it is possible to take measures against the problems of the series resonance type converter without enlarging the winding components and the capacitor, and since it is possible to easily protect the ZVS and the output overcurrent, the switching power supply device Can be effective in reducing the size, cost and noise.

  FIG. 5 shows a first modification of the resonant converter shown in FIG. This resonant converter is characterized by having the following configuration. First, a transformer T provided with first and second primary coils N1, N2 whose one terminal is connected to one terminal of the DC power source Vi and a secondary coil N3 is provided.

  The first switch element SW1 and the second switch element SW2 connected in series are connected between the connection point of the first series circuit and the other terminal of the first primary coil N1. A first series resonant circuit comprising an inductor L1 and a third capacitor C3; a second inductor L2 connected between the connection point of the first series circuit and the other terminal of the second primary coil N2; And a second series resonant circuit comprising a fourth capacitor C4. Further, an input secondary coil N3, are provided with a diode bridge D3, D4, D5, D6 for rectifying the current flowing in the secondary coil N3. About another structure, it is substantially the same as embodiment of FIG. 1 illustration.

  With the above configuration, the leakage inductance of the transformer T can be all or part of the inductors L1 and L2 of the resonance circuit. The operation is substantially the same as in the embodiment shown in FIG. The same applies to the following modifications.

  FIG. 6 shows a second modification of the resonant converter shown in FIG. The resonant converter is substantially the same as the modification of FIG. 5, this modification is characterized in that the switch with MOSFET, Q1, Q2. The first and second diodes D1 and D2 and the first and second capacitors C1 and C2 are also characterized as having parasitic elements of MOSFETs Q1 and Q2, respectively. Incidentally, the first and second diodes D1, D2, and the first and second capacitors C1, C2 may be used external components MOSFET, Q1, Q2.

  FIG. 7 shows a third modification of the resonant converter shown in FIG. This resonance type converter uses a center tap rectification method for the output of the transformer, and is effective when the output voltage is low.

  FIG. 8 shows a first embodiment of a transformer structure usable in the first to third modifications. This transformer has a tripod iron core, and two primary coils N1, N2 and a secondary coil N3 are wound around a central monopod and resonates the leakage inductance between the primary coils N1, N2 and the secondary coil N3. Can be used for

  FIG. 9 shows a second embodiment of a transformer configuration usable in the first to third modifications. This transformer has a tripod iron core, in which a first primary coil N1 and a secondary coil N3 are wound around a first leg at one end, and a second primary coil N2 is wound around a second leg at the other end. It is. Since the leakage inductance between the second primary coil N2 and the secondary coil N3 can be adjusted by the central third leg, it is effective when the two resonance circuits having different resonance frequencies are configured by the leakage inductance as in the present invention. .

  Although the transformers shown in FIGS. 8 and 9 are proved to be useful for the present invention in an equivalent circuit by magnetic circuit / electric circuit conversion. These transformers are merely examples, and the configuration of the transformer is not limited, for example, a two-leg iron core is used. Further, the present invention is not limited to the resonant converter according to the embodiment described with reference to FIGS. For example, a plurality of secondary coils may be provided and applied to a multi-output power supply device. The switch element is not limited to the MOSFET, but can be applied to other switch elements such as IGBT and BiTr. It is also possible to replace the half bridge circuit with a full bridge circuit or replace the output rectifier diode with SRMOS. In the case of having a transformer, ZVS can be caused by the action of an exciting current of the transformer in addition to the resonance current.

  According to the present invention, the operating frequency fs is controlled between the first series resonant frequency fr1 and the frequency fv at which the output voltage becomes a valley point in this way, thereby taking advantage of the series resonant converter such as ZCS. In addition, it is possible to control such as keeping the output constant with respect to input and load fluctuations, making it possible to narrow the frequency fluctuation range, and making ZVS of switching elements easy, etc. Is available.

It is a circuit block diagram of the best embodiment of the present invention. 1 is an operation waveform diagram at the time of full load in the embodiment shown in FIG. 1 is an operation waveform diagram at light load in the embodiment shown in FIG. 1 is a diagram showing an output voltage when the operating frequency in the embodiment shown in FIG. 1 is changed. It is the circuit block diagram which showed the 1st modification. It is the circuit block diagram which showed the 2nd modification. It is the circuit block diagram which showed the 3rd modification. 1 is a configuration diagram illustrating a first embodiment of a transformer according to the present invention. It is the block diagram which similarly showed the 2nd Example of the transformer. It is a circuit block diagram which shows an example of the conventional resonance type converter. FIG. 11 is an operation waveform diagram in the conventional example shown in FIG. 10. It is a circuit block diagram which shows the prior art example different from the prior art example shown in FIG. FIG. 13 is an operation waveform diagram at full load in the conventional example shown in FIG. 12. It is a circuit block diagram which shows the prior art example different from the said prior art example. It is a wave form diagram which shows the principle which reduces the fluctuation | variation of an operating frequency.

Explanation of symbols

Vi DC power supply C1, C2, C3, C4, C5, Cr capacitor Cp capacitor L1, L2, Lr inductor Lp inductor SW1, SW2 switching elements Q1, Q2 MOSFET
T transformer N1 coil N2 coil N3, N4 coil D1, D2, D3, D4, D5, D6 diode D3-D6, DB diode bridge

Claims (6)

A first series circuit of a first switch element and a second switch element connected to a DC power source;
A first series resonance circuit comprising a first inductor and a third capacitor, one junction point connected to a connection point of the first series circuit, and a second series comprising a second inductor and a fourth capacitor. A parallel circuit composed of two series resonant circuits;
One of the AC terminals is connected to the other junction of the parallel circuit, the other of the AC terminals is connected to one end of the DC power supply, and a diode bridge that rectifies the current of the two series resonant circuits;
A fifth capacitor for smoothing the output of the diode bridge;
A circuit for alternately turning on and off the first switch element and the second switch element at the same time, with a short period of time being off, and changing the operating frequency to control the voltage of the fifth capacitor; Have
A resonance type converter characterized in that the resonance frequency of the first series resonance circuit and the resonance frequency of the second series resonance circuit are set to be different values. A first series circuit of a first switch element and a second switch element connected to a DC power source;
A transformer having a first and a second primary coil, one of which is connected to one of the terminals of the DC power source, and a secondary coil;
A first series resonant circuit comprising a first inductor and a third capacitor connected between a connection point of the first series circuit and the other terminal of the first primary coil;
A second series resonant circuit comprising a second inductor and a fourth capacitor connected between the connection point of the first series circuit and the other terminal of the second primary coil;
A diode bridge that receives the secondary coil as an input and rectifies a current flowing through the secondary coil;
A fifth capacitor for smoothing the output of the diode bridge;
A circuit for alternately turning on and off the first switch element and the second switch element at the same time, with a short period of time being off, and changing the operating frequency to control the voltage of the fifth capacitor; Have
A resonance type converter characterized in that the resonance frequency of the first series resonance circuit and the resonance frequency of the second series resonance circuit are set to be different values. 3. The resonance according to claim 2, wherein the transformer is formed by winding a first and second primary coil on a monopod of a bipod or tripod core and a secondary coil on the monopod or another monopod. Shape converter. A first switch element is connected in parallel with the first diode and the first capacitor, respectively, and a second switch element is connected in parallel with the second diode and the second capacitor, respectively. 4. The resonant converter according to claim 1, further comprising a second parallel circuit connected thereto. 5. 5. The resonant converter according to claim 1, wherein the first and second inductors have part or all of leakage inductance of the transformer. 6. The first and / or second diode and the first and / or second capacitor are included as parasitic elements of the first and / or second switch element. A resonant converter according to any one of the above.

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