JPH05176532A - Power circuit - Google Patents

Abstract

PURPOSE:To realize stabilization of output by a simple construction in a power circuit wherein all switching operations are executed by a zero voltage or a zero current by utilizing current and voltage resonance operations alternately. CONSTITUTION:By turning a switching element S1 ON and OFF, parallel resonance is executed by the primary self-inductance of a transformer T1 and a capacitor C1 and series resonance is executed by leakage inductances L2 between the primary and secondary sides and by capacitors C2a to C2c. Thereby zero-voltage ON and zero-current OFF are realized. Constant-voltage circuits composed of switching elements S3a to S3c, inductances L4a to L4c, diodes D3a to D3c and capacitors C6a to C6c are connected to the capacitors C2a to C2c and thereby constant voltages are attained.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply circuit of a switching inverter type which utilizes both voltage resonance and current resonance to reduce switching loss to the limit to improve conversion efficiency. Has been realized.

[0002]

BACKGROUND OF THE INVENTION Japanese Patent Application No. 3-166383, filed by the applicant of the present invention, discloses a power supply circuit that utilizes both voltage resonance and current resonance to reduce switching loss to the limit and improves conversion efficiency. And the ones shown in the drawings. This power supply circuit will be described. This power circuit
By using both voltage resonance and current resonance to reduce switching loss to the limit and improving conversion efficiency, the operating waveform of each voltage and each current in the circuit was made closer to a sine wave to reduce noise. It is a thing.

As shown in FIG. 2, this power supply circuit includes a DC power supply 1 and a switching element which can be turned on and off at arbitrary timings, and switching means 2 for switching the input DC power supply to convert it into an alternating current and output it. When,
A series resonance means 4 formed in series with a current flowing through the output terminal of the switching means, a parallel resonance means 5 formed in parallel with a voltage generated at the output terminal of the switching means, a series resonance means, and Parallel resonance means 5
DC output means 3 for full-wave rectifying an AC input supplied via the DC output and smoothing it with a capacitor to obtain a DC output, and timing control means 6 for controlling the switching elements of the switching means to be turned on and off periodically. And.

FIG. 3 is a basic principle block diagram showing the block of FIG. 2 a little more structurally. The operation of the basic principle configuration shown in FIG. 3 will be described with reference to FIG. 4 showing the operation timing of each part. In FIG. 3, the switching elements S1 and S2 are
When the power supply voltage + VI and -VI are repeatedly turned on and off at the timings of FIG. It is rectified by the diode D2, smoothed by the capacitors C3 and C4, becomes a direct current of ZV1, and a current flows through the load RL.

When S1 is on, D1 is in the forward direction and the charge current flows into C3. However, assuming that the impedance of S1 and D1 is sufficiently small, C3 >> C2 is set. It becomes a sinusoidal series resonance current due to C2 (see (a) in FIG. 4). This resonance current becomes a reverse voltage and turns off when a half wave passes and the direction of the current is reversed. Therefore, series resonance cannot be performed and resonance stops. That is, the resonance automatically stops when the resonance current ends in half wave and the current returns to zero.

At this time, in C2, charges corresponding to the flowing resonance current are accumulated and a voltage remains at both ends (see (e) in FIG. 4). This charge QC2 = C2 · VC2 is released to the load in the cycle in which the next S2 is turned off, so that it does not cause energy loss. Since the energy stored in the inductance is proportional to the current, the energy of L2 is zero when the resonance stops at zero current. This means that the generation of harmful noise here is extremely small, and is a major point in which the voltage resonance mode is established in the final circuit.

In order to completely reduce the magnetic energy of L2 to zero, it is necessary to turn on S1 until the resonance current returns to zero. After the resonance current becomes zero, nothing happens even if S1 is kept turned on, but it is inefficient because the time during which energy is not transmitted becomes long, so it may be turned off with some margin. Since the resonance time due to L2 and C2 is constant, the ON time of S1 may be a constant value.

When S1 is turned off, the current resonance ends and the current becomes zero. Therefore, the current flowing through S1 is only the current flowing through the inductance L1. The value of L1 is L2,
It can be set independently of C2, and by setting L1 >> L2,
Since the current flowing through L1 can be set to a value sufficiently smaller than the resonance currents of L2 and C2, S1 is almost zero current off, and the loss at the time of off is extremely small. When S1 is turned off (since S2 has not been turned on yet, both S1 and S2 are turned off). Since D1 and D2 are also turned off, the operation here is only L1 and the capacitor C1.

The magnetic energy (current) stored in L1 while S1 is on becomes energy for operating parallel resonance with C1, and the voltage at point A is reduced in a sine wave form.
It goes beyond zero and approaches -VI. The operation during this period is the voltage resonance mode. In principle, the voltage resonance waveform is symmetrical with respect to a point intersecting with the voltage reference potential (potential expressed as zero in FIG. 4A) and is formed as shown in FIG. 4A. However, depending on the circuit configuration (specifically, it may be considered that the timing control circuit or the like partially consumes the energy of this voltage resonance through its coupling winding), the waveform deformation may occur. Can happen.

When the potential at the point A becomes close to -VI (lower than the potential at one end of C4), D2 is turned on and the remaining energy (current) of L1 is released to C4 through L2, C2, D2, but L1. Since the current of is originally set to be small, it does not change greatly in terms of current and the potential at point A is-
It will be stopped near VI. As it is, S1, S2
If you keep turning off, the magnetic energy (current) of L1 becomes zero in about half the time that S1 was on and L1
The voltage across (C1) drops from a potential near -VI toward zero. Conversely, for about half the on-time of S1, the magnetic energy of L1 causes point A to be -VI.
Since the potential can be maintained at a near potential, if S2 is turned on during that time, S2 becomes a zero voltage ON operation in which the voltage across the S2 is very small, and the loss at the time of ON is extremely small.

The voltage across S2 is turned on (-V above).
Strictly speaking, the difference between the value expressed as near I and −VI is not zero, and VC2 that remains after the current resonance when S1 is on is mainly
And so on. However, VC2 has different values depending on the value of C2. L2 at the same resonance frequency
In consideration of this, there is a degree of freedom in the setting of C2, and generally, in the range where series resonance normally occurs, the larger C2 is and the smaller L2 is, the smaller the loss is. Compared to, the voltage is almost negligible. When S2 is turned on, current resonance on the negative side occurs and C
A charge current flows through 4. Thereafter, as shown in FIG. 4, the above-described operation is repeated while switching the positions of S1 and S2.

From the time when S1 is turned off to the time when S2 is turned on, point A is -VI due to voltage resonance caused by L1 and C1.
It only takes a little longer than the time it takes to get closer, and this too is just inefficient if taken too long. This time does not require a strict setting and may be a fixed value.

As a precaution, the ON period of S1 and S2 and the time from when S1 or S2 is turned off to when S2 or S1 is turned on will be examined a little now. It can be said that the ON period of the switch element is longer than the resonance half cycle of the series resonance means, and the OFF period of both switch elements is shorter than 1/2 of the resonance cycle of the parallel resonance means. Also, L
It should be noted that the amount of energy given to the voltage resonance circuit by C1 and C1 in advance is examined, and additionally, how to set each value of L1 and C1 even when the same parallel resonance frequency is set. That is, the ON period of each switch element determines the applied energy, and the OFF period is naturally restricted by the given energy (that is, the ON period equivalent value). According to the analysis, in practice, if the on period and the off period are determined, the switching frequency is determined at that time, and the parallel resonance (voltage resonance) frequency that satisfies the operation of the present invention and the portion of use of the parallel resonance waveform are determined. Has been determined to be unique. For example, when the ON period is set to a finite small value (nearly zero), the voltage resonance waveform in that case appears to change substantially on a sine wave at the same frequency as the switching frequency. It should be noted that in some cases, even when the voltage peak value of the voltage resonance is reached, the desired change in the output terminal potential of 2VI may not be realized yet.

Further, as is clear from the above description, L1 >> L2, C2 >> are set as conditions for setting the value of each resonance circuit.
> C1 is desirable, the rectification method needs to be a full-wave rectification method, the smoothing method is a capacitor input method for current resonance, and the smoothing capacitor has a capacitance considerably larger than that of the series resonance means. Make it big,
It is necessary to prevent the Q of current resonance from decreasing.

When the above-described principle configuration is to be embodied as an actual circuit, as is clear from the above description of the principle configuration, the actual setting condition of the value of each resonance circuit is as follows.
Since L1 >> L2 and C2 >> C1 are desirable, L1 is the primary self-inductance of the transformer, L2
Can be effectively used by using independent inductance or by using leakage inductance between the primary and secondary sides of the transformer. Since the rectifier circuit is located on the secondary side of the transformer, either a center tap method or a bridge method may be used, but it is necessary to use a full-wave rectification method because it is necessary to perform current resonance with positive and negative currents. The smoothing method is
Capacitor input method for current resonance, C
3 >> C2 is set so that the Q of current resonance is not lowered.

When the transformer is viewed from the primary side, it looks like FIG. Since the transformer originally has a self-inductance and a leakage inductance, it can be used in place of L1 and L2 in FIG. 3 by setting them to appropriate values at the time of design. Also, in general transformers originally L1
> L2.

FIG. 6 is a modification of the principle configuration circuit shown in FIG.
become that way. In FIG. 6, current resonance is performed by L2 and C2 divided into two, and voltage resonance is performed by C1 divided into two.
And L1. In the loop of voltage resonance, L2, C
2 seems to be different from the one shown in FIG. 3, but since L2 << L1, C2 >> C1, L2, C
The presence of 2 does not affect the voltage resonance, and substantial voltage resonance is performed by C1 and L1 as in the configuration of FIG.

FIG. 7 is a more specific implementation circuit using a transformer T1 having a self-inductance L1 and a leakage inductance L2. The output circuit is of center tap type. The reason for adopting the center tap method is to reduce the number of diodes on the rectification path in each rectification cycle and minimize the loss due to these diodes to contribute to the improvement of the efficiency of the entire circuit. Then, the bases of S1 and S2 are driven at a fixed timing by a drive circuit having a timing as shown in FIGS. With such an extremely simple circuit, a low noise and highly efficient power supply circuit can be realized.

The effects of the configuration of the power supply circuit described above can be summarized as follows. The effect of current resonance is to reduce current noise. A large amount of current noise is generated when a sudden change in current is generated especially in a large amount of current, but noise is extremely generated because the current that has changed sinusoidally due to current resonance automatically stops when it reaches zero. Few. Next is efficiency improvement, but S1 and S
2 is turned off at zero current, and the voltages of D1 and D2 are also inverted after the current is zero. Therefore, the influence of recovery time is small, and the deterioration of efficiency due to this is eliminated. The effect of voltage resonance is to reduce noise and improve efficiency.
Components such as semiconductors used in the power supply circuit are attached to a chassis or the like via an insulator for heat dissipation, and thus the component electrodes and the chassis have an electric capacity. Therefore, current flows through this capacity where the component electrode has an AC signal,
It is the main cause of common mode noise. In addition, the semiconductor itself has a junction capacitance, and the inductance and the transformer also have a line capacitance. Although these capacitances are not shown in the circuit diagram, they actually exist in each component or circuit board, so that current flows through all these capacitances when the circuit is operating. Since this current is a current flowing through the capacitor, the larger the voltage change (dV / dT), the larger the current becomes, and when switching with a square wave, it becomes a pulsed current, causing current noise, or the current flowing in the chassis. Causes pulsed common mode noise. Further, since this pulsed current is supplied from the switching transistor, it naturally causes a loss and reduces efficiency. Since a large voltage of dV / dT contains a high frequency component, a radio wave (unnecessary secondary radiation) directly emitted from the circuit naturally becomes large.

These improvements can be realized by making the waveform part of a sine wave by utilizing voltage resonance and reducing dV / dT. However, in this power supply circuit, this voltage resonance is S1,
When both S2 are off, it is made up of only L1 and C1, so switching element loss does not occur and L1 and C1
The current flowing through 1 is also only a transfer of energy between each other,
There is only reactive power, and the loss due to voltage resonance is extremely small (it is zero in principle).

What is important here is that in order to reduce voltage noise, dV / of voltage waveforms of all terminals in the circuit is reduced.
It is necessary that dT is small. If there is a square waveform even at one place, it will become a noise source. In a general voltage resonance type power supply circuit, although a certain point in the circuit (for example, a transformer output) has a sine wave shape, a square waveform exists (in another circuit portion) in many cases. This power supply circuit has the most important goal of practical noise reduction, and is characterized in that all voltage waveforms are similar to the voltage resonance waveforms of L1 and C1. The reason for satisfying this point is that the voltage resonance is used separately from the current resonance. By current resonance, the currents of S1, S2 and D2 are set to zero,
After the magnetic energy of L2 is also zeroed, it is brought into the voltage resonance mode and S1, S2, D1, and D2 are kept in the OFF state, so that the current movement of L2 and C2 in the voltage resonance mode is zeroed. The waveforms at points A and A'are the same.
As a result, the terminal voltage waveforms of L1 and C1 and S1, S2,
The waveforms of all terminals of L2, C2, D1, and D2 become the same (similar), and the square waveform disappears from the circuit.

When using a transformer, current resonance can occur on either the primary side or the secondary side. FIG. 8A shows an example in which current resonance occurs on the primary side. The current resonance is realized by the primary-secondary leakage inductance L2 of the transformer T1 (value seen from the primary side) and the capacitor C2 arranged on the primary side. The voltage resonance is realized by the primary self-inductance L1 of the transformer T1 and the capacitor C1 divided into two.

However, in a multi-output power supply having a plurality of secondary windings, the leakage inductance between the primary and the secondary of the transformer often differs from winding to winding, so that the current resonance capacitor C2 is used for each secondary winding. In the current resonance on the primary side that is commonly used in the windings, the resonance frequency with the capacitor C2 differs for each secondary winding. Therefore, the current resonance does not end at the same timing, and the current resonance takes too long to satisfy the zero current off, or the power resonance in which the current resonance ends too early and the peak value of the resonance current increases every time. There is a problem that a grid is created and loss is increased. Therefore, it is desirable to carry out current resonance on the secondary side.

An example of current resonance on the secondary side is shown in FIG.
It shows in (C). In FIG. 8B, the current resonance is represented by the transformer T.
It is realized by the primary-secondary leakage inductance L2 (value seen from the secondary side) and the capacitor C2 arranged on the secondary side. In a multi-output power supply having a plurality of secondary windings, a current resonance capacitor is arranged for each secondary winding, and the primary and secondary leakage inductance of each secondary winding is adjusted so that the same resonance frequency can be obtained. Set the values of these capacitors individually according to the value of.

FIG. 8C shows that the switching means is composed of one stone. Current resonance is the primary of transformer T1, 2
Secondary leakage inductance L2 (value seen from the secondary side) and 2
It is realized by the capacitor C2 arranged on the next side. Also,
The voltage resonance is the primary self-inductance L1 of the transformer T1.
And the capacitor C1. Switching element S
1 is connected in parallel to the capacitor C1.

The operation of this power supply circuit will be described with reference to FIG. When the switching element S1 is turned on, the diode D1 is in the forward direction and the charge current I _D1 changes to the capacitor C2.
Flow into. This current I _D1 becomes a sinusoidal series resonance current due to the inductance L2 and the capacitor C2 (FIG. 9).
(See (a)). This resonance current becomes a reverse voltage and turns off when a half wave passes and the direction of the current is reversed. Therefore, series resonance cannot be performed and resonance stops. That is,
The resonance automatically stops when the resonance current ends half-wave and the current returns to zero.

At this time, since the current of the inductance L2 is zero and the energy is completely discharged, spike-like noise is not generated due to the turning off of the diode D1. The voltage stored in the capacitor C2 due to the current resonance is smoothed by the inductance L3 and the capacitor C3 and is output as DC.

After the current resonance is completed, the switching element S1 is turned off. Current resonance time (see Fig. 8 (a))
Is a constant value determined by the values of the inductance L2 and the capacitor C2, the on time of the switching element S1 (see FIG. 8C) may be a constant value, and this on time is a value slightly longer than the current resonance time. To be turned off after the current resonance is completely completed.

While the switching element S1 is on, the voltage V _E is also applied to the inductance L1 and the exciting current I _L1 flows through the inductance L1. When the switching element S1 is turned off, the current I _L1 flowing in the inductance L1 flows in the capacitor C1 and voltage resonance is started.
In voltage resonance, the voltage across the inductance L1 (see FIG.
(A) is a sine wave due to resonance with the capacitor C1, drops from the voltage V _E , passes through zero, becomes a reverse voltage, and returns to the voltage V _E again.

When the switching element S1 is turned on at the timing when the voltage of the inductance L1 returns to V _E , the current resonance mode is entered again, and the above operation is repeated thereafter. The voltage resonance time (see FIG. 8D) is the time when the switching element S1 is off, and this time is until the voltage across the inductance L1 (FIG. 8A) returns to the voltage V _E. Set to time (determined by the inductance L1 and the capacitor C1). As described above, the zero-voltage ON and the zero-current ON are realized with the one-stone configuration. ,by the way,
When current resonation is performed on the secondary side of the transformer T1 as shown in FIGS. 8B and 8C, the output is taken from both ends of the resonance capacitor C2, so that a voltage generated with the resonance current is generated in the capacitor C2. This will be output as ripples, and a removal filter will be required. Further, since the load impedance is connected across the capacitor C2, this affects the current resonance. In order to solve these problems, it is necessary to use a choke input filter having an impedance sufficiently larger than the resonance impedance of L2 and C2. This is shown in FIG.
The smoothing inductance L3 and the smoothing capacitor C3 in (C).

[0031]

2. Description of the Related Art The power supply circuit which combines the current resonance and the voltage resonance described above does not have a special output stabilizing (constant voltage) function. A series switching regulator should be attached on the rear side.

An example of the circuit configuration of the series switching regulator is shown in FIGS. Each of these switches the DC voltage (unstable output) stored in the capacitor C5 so that the switching element S3 is turned on and off in a variable ratio of off time, and the energy transferred during the on period or off period is changed to the inductance L4 (or L4 , L5)
And the capacitor C6 (or C6, C7) smoothes it to a constant voltage and supplies it to the load RL. The solid-state arrows indicate the flow of current when the switching element S3 in each circuit is on, and the dotted-line arrows indicate the flow of current when it is off.

[0033]

The problem to be solved by the invention is shown in FIG.
In the power supply circuit of (C), the output can be stabilized by connecting any of the series switching regulators shown in FIG. 10 at the subsequent stage, but if it is simply connected, the circuit configuration becomes complicated and the device becomes large. I will end up. As an example of simply connecting, the configuration in which the series switching regulator of FIG. 10 (a) is connected to the latter stage of FIG. 8 (C) is shown in FIG.
Shown in 1. This circuit compares the values of the three inductances L2, L3, and L4. The transformer primary-secondary inductance L2 current-resonates with the capacitor C2 and sends energy to the capacitor C2 with a half-wave current. Since this current is terminated in each switching cycle, the current flowing through the inductance L2 becomes a current in a mode in which it is intermittent with a half wave. At this time, the ripple voltage generated in the capacitor C2 is applied to the inductance L3 connected to this capacitor C2. Since the inductance L3 removes this ripple voltage, it is common to set the inductance value so that the peak-to-peak value of the ripple current at the rated output is ½ or less of the output current. For the same ripple voltage, the inductance L2 functions in the intermittent mode, while the inductance L3
In order to function as a ripple filter in the continuous mode, it is necessary to set L3 = 4 · L2 or L3> 4 · L2.

On the other hand, a voltage close to the input voltage (voltage of the capacitor C5) is applied to the inductance L4 of the constant voltage circuit due to the on / off operation of the switching element S3. This voltage is naturally higher than the ripple voltage of the capacitor C2. Since the inductance L4 is for removing this ripple voltage, the condition that the inductance value is set so that the peak-to-peak value of the ripple current at the rated output becomes 1/2 or less of the output current has been described above. It is the same as the inductance L3. Therefore, it is necessary to satisfy L4> L3. Therefore, the inductances L2, L
It is necessary to set L4>L3> L2 for L3 and L4, and two inductances L3 and L4 larger than L2 are required. Also, three electrolytic capacitors, C3, C5 and C6, are required, and simply C3
Even if you combine C5 into one, you still need two. The present invention has been made in view of the above points, and an object of the present invention is to provide a power supply circuit capable of stabilizing the output with a simple configuration.

[0035]

According to the present invention, there is provided a DC power source and a switching element capable of ON / OFF control, switching means for switching the DC power source to convert into AC, and outputting the AC power. A transformer applied to the secondary side, a primary self-inductance of the transformer, and a capacitor disposed on the primary side of the transformer are used at least and are connected in parallel to the voltage generated at the output terminal of the switching means. A parallel resonance means formed in the transformer, a leakage inductance between the primary and secondary sides of the transformer, and a capacitor arranged on the secondary side of the transformer, and a current flowing to the output terminal of the switching means. A series resonance means formed in series with respect to the constant resonance circuit, and a switching circuit for a constant voltage circuit at both ends of the capacitor of the series resonance means. A smoothing inductor connected via a child and a smoothing capacitor to which a current flowing in the smoothing inductance is supplied, and the voltage across the smoothing capacitor is taken out as an output. Is.

[0036]

In the constant voltage circuit without the capacitor C5, the impedance Z _in seen from the input side is the inductance L4 itself as shown in FIG. 12A regardless of whether the switching element S1 is on or off. .. Moreover, as described above, L4> L3 and L3> 4.L2 (or L
3 = 4 · L2), the inductance L4
Is directly connected to the inductance L2, L4> 4 · L2 is satisfied, and there is no problem in the operation of current resonance. In that case, the inductance L4 and the output capacitor C6 naturally function as a smoothing inductance and a capacitor,
No problem. Therefore, the output capacitor C3 of the switching inverter circuit and the input capacitor C of the constant voltage circuit
4 and the smoothing inductance L3 of the switching inverter circuit can be removed.

Therefore, the circuit of FIG. 11 is the same as that of FIG.
It can be simplified as in (b), the configuration is simplified, and the device can be downsized.

As a constant voltage circuit, the CUK of FIG.
Even when the SEPIC of (e) is used, the BOO of (a)
Since the input impedance Z _in is the inductance L4 itself as in ST, it can be directly connected. In BUCK of (b) and BUCK-BOOST of (c), the switching element S3 is inserted before the inductance L4, but the switching element S in which the input and the inductance L4 are connected.
Since the diode D3 is turned off at the timing of 3 on, the impedance Z _in seen from the input becomes the inductance L4, which can be directly connected in the same manner. (F)
In ZETA, the parallel impedance of the inductances L4 and L5 is visible from the input when the switching element S3 is on, but this parallel impedance is set to operate in continuous mode as a ripple filter. It is satisfied that the parallel impedance> L3, which can also be directly connected. Therefore, the present invention is at least (a) to (f) of FIG.
It can be applied to all constant voltage circuits.

[0039]

(Embodiment 1) The one-stone secondary current resonance type switching inverter circuit of FIG. 8 (C) and FIG. 10 (a)
FIG. 1 shows an embodiment in which the present invention is applied to a combination of the BOOST type constant voltage circuit of FIG. Here, a transformer T1 including a plurality of secondary windings 11 to 13 is used to configure a multi-output power source including power source systems A to C.

The DC power source 1 is, for example, a rectified commercial AC power source,
It is composed of a power source or a battery for smoothing to obtain a DC voltage. The parallel resonance means 5 is composed of the primary mutual inductance L1 of the transformer T1 and the capacitor C1. Switching means 2 including a switching element S1 is connected to both ends of the capacitor C1.

The series resonance means 4a to 4c include the secondary winding 11
Each of the transformers has a different primary-secondary leakage inductance L2a to L2c (value seen from the secondary side) and capacitors C2a to C2c. Capacitors C2a-C
The value of 2c is set in relation to the values of the leakage inductances L2a to L2c so that the same resonance frequency is obtained in each system A to C.

Rectifying diodes D1a to D are provided between the transformer secondary windings 11 to 13 and the capacitors C2a to C2c.
1c is inserted. Smoothing inductances L4a to L4c are connected to both ends of the capacitors C2a to C2c via switching elements S3a to S3c. The switching elements S3a to S3c are the timing control means 1
4, the switching is performed in a variable on / off time ratio.
Smoothing capacitors C6a to C6 are provided at both ends of the switching elements S3a to S3c via diodes D3a to D3c.
c is connected, and the voltages across the capacitors C6a to C6c are supplied to the loads RLa to RLc as output voltages, respectively.

The operation of the switching element S1 by the timing control means 6 is performed by the switching element S shown in FIG.
This is the same as the operation (1) (FIG. 9). That is, when the switching element S1 is turned on, the diodes _{D1a to D1c} become forward and the charge currents I _{D1a to} I _D1c flow into the capacitors C2a to C2c. The currents I _{D1a to} I _D1c are the inductances L2a to L2c and the capacitors C2a to C.
It becomes a sinusoidal series resonance current due to 2c (see (a) in FIG. 9). This resonance current becomes a reverse voltage and turns off when a half wave passes and the direction of the current is reversed. Therefore, series resonance cannot be performed and resonance stops.
That is, the resonance automatically stops when the resonance current ends in half wave and the current returns to zero. At this time, the inductance L
Since the currents 2a to L2c are zero and the energy is completely released, spike-like noise is not generated when the diodes D1a to D1c are turned off.

After the current resonance is completed, the switching element S1 is turned off. Current resonance time (see Fig. 8 (a))
Are inductances L2a to L2c and a capacitor C2a.
Since it is a constant value determined by the values of C2c, the ON time of the switching element S1 (see FIG. 8C) may be a constant value, and the ON time is set to a value slightly longer than the current resonance time. Set to turn off after the current resonance is completely completed.

While the switching element S1 is on, the voltage V _E is also applied to the inductance L1 and the exciting current I _L1 flows through the inductance L1. When the switching element S1 is turned off, the current I _L1 flowing in the inductance L1 flows in the capacitor C1 and voltage resonance is started.
In voltage resonance, the voltage across the inductance L1 (see FIG.
(See (a)) is a sine wave due to resonance with the capacitor C1, drops from the voltage V _E , passes through zero, becomes a reverse voltage, and returns to the voltage V _E again.

When the switching element S1 is turned on at the timing when the voltage of the inductance L1 returns to V _E , the current resonance mode is entered again, and the above operation is repeated thereafter. The voltage resonance time (see FIG. 8D) is the time when the switching element S1 is off, and this time is until the voltage across the inductance L1 (FIG. 8A) returns to the voltage V _E. Set to time (determined by the inductance L1 and the capacitor C1).

Control of the constant voltage switching elements S3a-S3b by the timing control means 14 is performed as follows. When the switching elements S3a to S3c are turned off, the energy stored in the capacitors C2a to C2c is changed to the inductances L4a to L4c and the diodes D3a to D.
It is transmitted to the output capacitors C6a to C6c via 3c. When the switching elements S3a to S3c are turned on, this transmission is cut off. Therefore, the timing control means 14 outputs the output voltage of each system A to C (capacitors C6a to C6).
6c) is detected, and the switching elements S3a to S3c are individually turned on and off according to the change, and switching is performed (the off time is lengthened when the output voltage decreases and the on time increases when the output voltage increases). To stabilize the output. It should be noted that the timing control means 14 may be constructed completely independently of the timing control means 6, but for example, the timings of both are synchronized so as to prevent generation of a beat sound due to the interference of their switching frequencies. You can also do it.

Although the case where the BOOST of FIG. 10 (a) is applied to the power supply circuit of FIG. 8 (A) as the constant voltage circuit has been shown above, the circuit of FIGS. 10 (b) to (f) is also shown in FIG. It can be similarly applied to the circuit of (A). An example of the circuit configuration of the secondary side of the transformer T1 when applied is shown in FIG.
It shows in (a)-(e). The same parts as those in FIG. 10 are designated by the same reference numerals.

(Embodiment 2) The two-stone secondary side current resonance type switching inverter circuit of FIG. 8 (B) and FIG.
FIG. 14 shows an embodiment in which the present invention is applied to a combination of the BOOST type constant voltage circuit of (a). Although the transformer T1 has a single output here, it can have a plurality of outputs as in FIG. The parallel resonance means 5 is composed of the primary self-inductance of the transformer T1 and the capacitor C1. The secondary side of the transformer T1 is configured by a center tap method, and the voltage induced in the secondary winding is applied to the diodes D1 and D2.
Is full-wave rectified and stored in the capacitor C2. The series resonance means 4 is a leakage inductance L between the primary and secondary sides of the transformer T1.
2 and a capacitor C2. A smoothing inductance L4 is connected to both ends of the capacitor C2 via a switching element S3. The switching element S3 is
The timing control means 14 switches the ON / OFF time ratio in a variable manner. A smoothing capacitor C6 is connected to both ends of the switching element S3 via a diode D3, and the voltage across the capacitor C6 is supplied to the load RL as an output voltage.

The operation of the switching element S1 by the timing control means 6 is the same as that shown in FIG. 4, whereby zero voltage on and zero current off are realized.

Control of the constant voltage switching element S3 by the timing control means 14 is performed as follows. When the switching element S3 is turned off, the energy stored in the capacitor C2 is transferred to the inductance L4 and the diode D.
3 is transmitted to the output capacitor C6. When the switching element S3 is turned on, this transmission is cut off. Therefore, the timing control unit 14 switches the switching element S3 by varying the on / off time ratio in accordance with the output voltage (voltage of the capacitor C6) (when the output voltage decreases, the off time is lengthened, and when the output voltage increases, the switching time increases). The output is stabilized by increasing the ON time.).

Although the case where the BOOST of FIG. 10 (a) is applied to the power supply circuit of FIG. 8 (B) as a constant voltage circuit has been described above, the circuit of FIGS. 10 (b) to (f) is also shown in FIG. The same can be applied to the circuit of (B). That is, FIG.
13 (a) to 13 (e) show the part after the capacitor C2 of FIG.
It may be replaced with the structure after the capacitor C2.

[0053]

As described above, according to the present invention, when the switching inverter circuit and the series switching regulator are combined, the number of smoothing regulators and smoothing inductances can be reduced as compared with the case where they are simply combined. The configuration is simplified and the device can be downsized.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a schematic block diagram showing the basic principle of a power supply circuit described in Japanese Patent Application No. 3-166383.

FIG. 3 is a configuration circuit diagram showing a basic principle configuration of the power supply circuit of FIG.

FIG. 4 is a timing diagram illustrating the operation of the principle configuration shown in FIG.

FIG. 5 is an explanatory diagram illustrating an equivalent circuit of a transformer.

FIG. 6 is a circuit diagram illustrating a modification of the principle configuration shown in FIG.

7 is a circuit diagram showing a configuration obtained by modifying the principle configuration shown in FIG.

8 is a circuit diagram showing a specific configuration example of the principle configuration of FIG.

9 is an operation waveform diagram of the power supply circuit of FIG. 8C.

FIG. 10 is a circuit diagram showing various examples of a constant voltage circuit.

11 is a circuit diagram showing a configuration in which the constant voltage circuit of FIG. 10 (a) is simply added to the power supply circuit of FIG. 8 (A).

12 is a circuit diagram showing a state in which a part of the element is removed from the constant voltage circuit of FIG.

13 is a circuit diagram showing a state in which various other constant voltage circuits are applied to the power supply circuit of FIG.

FIG. 14 is a circuit diagram showing another embodiment of the present invention.

[Explanation of symbols]

1 DC Power Supply 2 Switching Means 3 DC Output Means 4 Series Resonance Means 5 Parallel Resonance Means 6 Timing Control Means S1, S2 Switching Elements S3 Constant Voltage Circuit Switching Elements T1 Transformers L1 Primary Self-Inductance of Transformers C1 On Primary Side of Transformers Disposed capacitors L2, L2a, L2b, L2c Transformer primary and secondary leakage inductance C2, C2a, C2b, C2c Capacitors disposed on the secondary side of the transformer L4, L5 Smoothing inductance C6, C7 Smoothing capacitor

Claims (1)

[Claims] 1. A switching means, which includes a direct current power source and a switching element capable of on / off control, switches the direct current power source to convert into an alternating current and outputs the alternating current, and a transformer to which an output of the switching means is applied to a primary side. A parallel resonance means which is configured by using at least a primary self-inductance of the transformer and a capacitor arranged on the primary side of the transformer and which is formed in parallel with a voltage generated at an output terminal of the switching means, A leakage inductance between the primary and secondary sides of the transformer and a capacitor arranged on the secondary side of the transformer at least, and is formed in series with a current flowing through an output terminal of the switching means. The series resonance means and a flat resonator connected to both ends of the capacitor of the series resonance means through a constant voltage circuit switching element. And use the inductance, it comprises a smoothing capacitor current is supplied to flow to the smoothing inductance, the power supply circuit formed by the taken out as an output voltage across the smoothing capacitor.