JP2781602B2 - Power converter control device and system thereof - Google Patents

Description

Description: BACKGROUND OF THE INVENTION The present invention relates to a control device for a power converter such as a converter having an AC motor variable speed driving inverter or the like and a converter for converting AC to DC, and a power supply voltage phase. And a converter control method and system for flowing a sinusoidal power supply current at a power factor of 1.

[Conventional technology]

Conventional power converter control devices include, for example,
o.4 (1987) pp. 14-17. In this method, a deviation between the DC voltage command value V _DC * of the inverter and the DC voltage detection value V _DC is output via an amplifier, and this output is used as an amplitude command I _q * of a power supply current command. Further, a phase voltage of the power supply voltage is detected via a transformer, and an output obtained by multiplying the phase voltage detection signal and an amplitude command _Iq * by a multiplier is used as a power supply current command. That is, the power supply current command has a power factor of 1 synchronized with the phase voltage. The current control circuit outputs a PWM signal to the converter via the hysteresis comparator so that the actual power supply current follows the AC current command.

[Problems to be solved by the invention]

In the above prior art, a highly reliable current command is obtained by multiplying the amplitude command _Iq * of the current command by the phase voltage itself by an analog multiplier, but it is considered that the digitalization of the control circuit is difficult. There is also a problem that if the phase voltage itself is distorted, the current command is also distorted. It should be noted that a three-phase transformer is used for the detector, and the size of the detector is slightly larger than that of the photocoupler, so that it is difficult to reduce the size and weight of the control device. SUMMARY OF THE INVENTION An object of the present invention is to provide a converter control device and a system thereof which are small, light, and inexpensive and are suitable for digitalization, while allowing a sinusoidal power supply current with a small power factor and a power factor to flow accurately.

[Means for solving the problem]

To achieve the above object, first, in order to provide a small, lightweight and low-cost control device, the phase of the power supply voltage is detected using a photocoupler without using a transformer. As means for detecting the phase of the power supply voltage, a reference power supply frequency command ω
_By integrating digitally a primary frequency command ω ₁ * to which ₀ * and a frequency command correction amount Δω ₀ * have been added, a phase command θ _RS * of the line voltage is obtained. Was calculated. Note that the frequency command correction value Δω ₀
* Indicates that the line voltage polarity is detected by a photocoupler and the zero phase point of the line voltage is determined in consideration of the operation delay time of the photocoupler and the like, and the phase command value Δθ _RS of the line voltage at this point is obtained.
, The correction amount Δω ₀ * is determined.

Furthermore, based on the phase command θ _RS * corresponding to the actual line voltage phase in order to accurately supply a sine wave power supply current with a power factor of 1,
Obtains the actual active current I _q and the reactive current I _d of the power supply current taking into consideration the delay of such current detection filter compensates proceeds power current command of the phase of the AC as I _d is zero , To compensate for the delay of the current control system.

Further, in order to convert the control circuit of the converter to a low-cost digital circuit using a one-chip microcomputer, the actual reactive component current _Id and the active component current _Iq are considered in consideration of the delay of the power supply current detection.
Detects a magnitude of the converter input voltage vector as I _d is zero, yet effective current command I _q * on the active current I _q matches, calculates the phase instantly, space vector PWM
A gate signal is output to the converter by processing.

Furthermore, by simplifying the software processing of the one-chip microcomputer, the switching frequency of the converter is increased to reduce the power supply current ripple, and in order to provide a high-speed current control system, the AC A comparison is made between the command value and the detected value to compare the magnitudes, and the comparison result is output as a gate signal of the converter.

[Action]

 Next, the operation of the power supply voltage phase detecting means will be described.

The commercial power frequency is based on 50Hz or 60Hz, usually ±
It fluctuates within 0.1Hz. Therefore, the reference frequency command ω ₀ *
Is set to 50Hz or 60Hz, and the frequency correction amount Δω ₀ * is about ± 0.1H
It will change within z. Then (ω ₀ * + Δω
₀ *) as a primary frequency command ω ₁ * and is digitally integrated at regular intervals to obtain a power supply voltage phase command θ _RS *. If the actual zero phase of the line voltage is detected using a photocoupler and the phase command θ _RS * at this point is held, the difference between the actual power supply voltage phase and the current phase command (corresponding to the phase error) is obtained. Can be detected. Therefore, this phase error Δθ
_In the direction in which _RS converges to zero, a frequency correction amount Δω ₀ * is output corresponding to Δθ _RS . As a result, since the phase command of the power supply voltage normally corresponds to the actual phase, an accurate power supply voltage phase can be detected. Furthermore, even after a momentary power failure, the phase command is output by integrating ω ₁ * at that time. Therefore, even when the power failure is restarted, the operation is performed so that almost no phase error occurs, and the accurate power supply voltage phase can be digitally determined. Can be detected.

Next, in the current control system, the actual power supply current flows with a delay due to a delay in the switching cycle of the converter in response to the AC power supply current command. As a result, a reactive component current _Id is generated and the power factor does not become 1. Therefore, based on the accurate power supply voltage phase, the actual power supply current is input by the A / D converter of the one-chip microcomputer in consideration of the delay of the current detection filter, etc., and the reactive component current I _{d is obtained} by software processing of the microcomputer. look, to compensate for advancing the phase of the supply current command as I _d becomes zero. In this way, a power supply current command whose phase is advanced by a delay of the current control system from the phase voltage phase of the power supply is output.

In addition, the operation of the converter control means, which has been converted to digital, converts the three-phase power supply current into two rotating coordinate axes (dq axes) in consideration of the current detection delay, and detects I _d and I _q . .
Therefore, d-axis voltage correction amount ΔV _d * of converter input voltage vector command ₁ * is output such that reactive component current I _d becomes zero. On the other hand, the q-axis voltage correction amount ΔV _q * is output so that the detected value I _q matches the effective component current command I _q *, and ΔV _d *, Δ
The magnitude and rotation phase of the instantaneous converter input voltage vector command V ₁ * are calculated from V _q *, the magnitude setting value V _R * of the power supply voltage, and the voltage drop amount of the AC reactor. Based on this, the three-phase switching pattern and its conduction time are calculated and set to the one-chip microcomputer built-in timer.
M signal is output. In this manner, the operation is performed by the software processing of the one-chip microcomputer until the PWM signal is generated, and an instantaneous voltage vector is output.

The operation of the other fully digitalized converter control means calculates the AC current command value having the power supply phase voltage phase with the effective current command value _Iq * as the amplitude. Similarly, the power supply current is detected, the magnitudes of the two are compared, and the result is given as a converter gate signal at regular intervals.

That is, when the current command is larger than the actual current, the positive arm of the converter is turned off. As a result, the converter input voltage decreases and the power supply current increases. on the other hand,
When the actual current is larger than the current command, the positive arm is turned on to increase the converter input voltage. As a result, the power supply current decreases. As described above, the software processing is performed only for comparing the magnitudes of the power supply currents, and the calculation time can be shortened, so that high-speed switching can be performed.

〔Example〕

 Hereinafter, an embodiment of the present invention will be described with reference to FIG.

The main circuit consists of three-phase power supply 1, voltage polarity detector 2, current detector 3,
AC reactor 4, converter 5, smoothing capacitor 6, load 7
The power consumed by the load 7 is supplied from the commercial power supply 1 via the converter 5. in this case,
The converter is PWM-controlled so that the power supply current of each phase flowing through the AC reactor 4 is a sine wave and has the same phase (power factor 1) as the power supply voltage of each phase. On the other hand, the control device of the converter includes a power supply voltage phase detecting means 8, a current amplitude command calculating section 9,
It comprises a current command generating means 10 and a current control circuit 11. The power supply voltage phase detecting means 8 which is a main part of the present invention includes:
A zero point detection unit 12 for detecting a zero phase of the line voltage v _RS , an addition unit 13 for adding a reference frequency command ω ₀ * of 50 Hz or 60 Hz and a frequency correction amount Δω ₀ * corresponding to a fluctuation amount of the power supply frequency,
An integrator 14 for digitally integrating the added primary frequency command ω ₁ *, and a power supply voltage phase command θ as an integrated output
Hold processing unit that holds _RS * at the zero phase of the line voltage
15, a (proportional + integral) compensating unit 16, and a limiter processing unit 17.

Also, the power supply voltage phase command θ _RS * is 3
When it exceeds 60 °, it returns to 0 and changes in a sawtooth wave. The output Δθ _{RS of the} hold unit is the difference between the actual power supply voltage phase and the power supply voltage phase command value, and is a phase error having a positive or negative value. Therefore, Δθ _{RS is set} to (proportional + integral) compensator 16,
The frequency correction amount Δω ₀ * is set via the limiter 17,
Δω ₀ * is corrected by feed bark control so that the phase error Δθ _RS converges to zero. Next, FIG. 2 shows a specific hardware configuration of the control circuit of the present invention. Photocoupler 18, resistor 19, diode as voltage polarity pole output 2
20, comprising an inversion circuit 21. Next, as shown in the block diagram of FIG. 1, the processing of the power supply voltage phase detecting means 8, the current command generating means 10 and the current amplitude calculating section 9 is performed by software processing of the one-chip microcomputer 22 as shown in FIG. Do with. First, the process of the current amplitude calculation unit 9 is as follows. First, a voltage detection value V _DC (converter output voltage) of the smoothing capacitor 6 is input through an A / D converter 23 built in a microcomputer, and a DC voltage command is input by a subtractor 24. The deviation from V _DC * is set as an amplitude command I _q * of the current command via a (proportional + integral) compensation process 25 and a limiter process 26. Note that I _q * is an effective current command. Therefore, V _DC
In the case of *> V _DC , since the effective power is supplied from the power supply so as to charge the smoothing capacitor 6, I _q * becomes positive. On the other hand, when V _DC * <V _DC , a power supply current command is given to regenerate the power of the capacitor to the power supply side, and I _q * becomes negative.

Next, the power supply phase detection processing 8 performs the software processing of the block diagram shown in FIG. 1 and outputs a phase command θ _RS * of the line voltage. Further, the current command generating means 10 outputs the effective component current I _q ,
It consists of a reactive component current _Id calculation process 27, a current control system delay compensation process 28, and a phase current command generation process 29.
The output is output to the D / A converter 30, and the AC power supply current command is output via the filter 31 for three phases. The current control circuit 11 is configured with a hysteresis comparator, compares the current command with the actual current, and outputs a PWM signal on / off signal so as to fall within the deviation when a deviation exceeding the hysteresis width occurs. , And the gate signal of the converter.

Next, a method of detecting a power supply voltage phase and a method of outputting a current command of a converter, which are main parts of the present invention, will be described in detail. FIG. 3 shows the overall configuration of the one-chip microcomputer software processing. In the interrupt processing (IRQ1) for each fixed period T _S , first,
The equation (1) is used to output the line voltage phase command θ _RS *.

θ _RS * = θ _RS * (n−1) + ω ₁ * (1) In the equation (1), 1 is added to the previous phase command θ _RS * (n−1).
Digital integration is performed by adding the next frequency command ω ₁ * to calculate the sawtooth-shaped power supply voltage phase command θ _RS * shown in FIGS. 4 and 5.

Next, the active component current _Iq and the reactive component current _Id are obtained, the phase of the current command is calculated, and the current command is output. Next, FIG. 2 shows a means for determining the power supply voltage phase. An interrupt process (IRQ2) that operates at the edge of the photocoupler output is provided, and its contents will be described with reference to the operation waveforms in FIGS. First, FIG. 4 shows an S1 signal obtained by detecting an approximate positive signal of the power supply voltage using the photocoupler shown in FIG. In other words, when the current flows through the primary side light emitting diode of the photocoupler to a certain extent or more, the line voltage v _RS
Since the output voltage operates at V _SLV or more, the S1 signal rises with a delay of ΔT time from the zero phase of v _RS , and this rise time is set to t ₁
At this point, an IRQ2 interrupt occurs. Next, the primary frequency command ω ₁ * immediately before t ₁ is integrated. Set the start time of IRQ1 to t _n
From this time, the value of θ _RS * output in the T _S section is represented by θ _n
* To, theta _RS * value at the zero cross point of the power supply voltage v _RS (t _z time) can be calculated by the phase error [Delta] [theta] _RS next (2) between the actual power supply voltage v _RS and phase command value. That is,
Since θ _RS * value at time t _n is θ _{n *} value t _z time in the θ _RS *
The value is given by equation (2).

Δθ _RS = (t _z −t _n ) · K + θ _n * (2) where t _z is t ₁ −ΔT. Further, the constant K is a power supply voltage phase change amount in one cycle section at the reference power supply frequency (50 Hz or 60 Hz). Next, the primary frequency correction amount Δω ₁
* Is given by equation (3).

Here, K _P is a proportional gain, and K _I is an integral gain. The primary frequency command ω ₁ * is given by the equation (4).

ω ₁ * = ω ₀ * + Δω ₁ * (4) Here, ω ₀ * is a reference frequency of 50 Hz or 60 Hz.

Next, the operation of power supply phase detection will be described with reference to FIG. θ _RS * at point A where v _RS = 0 is Δθ _A , and when Δθ _A is positive, ω ₁ * is controlled to be small from equations (3) and (4) to obtain v _RS in the next cycle. And the zero phase of θ _RS * are substantially synchronized. On the other hand, when Δθ _B is negative like the point _B , ω ₁ * is obtained from the equations (3) and (4).
Is increased, feedback control is performed such that θ _RS * always follows the power supply voltage phase, and θ _RS * can be detected as the power supply voltage phase.

As described above, the power supply voltage phase detection method of the present invention provides a method in which the phase command value at the zero cross point of the power supply voltage is Δθ _RS
Since the primary frequency command is corrected so that θ _RS converges to zero and can be configured with a hard circuit including only a photocoupler, there is an effect that an extremely low-cost, small-sized and lightweight control device is obtained. Furthermore, the operation delay time ΔT of the photocoupler
Also, the phase error Δθ _RS is calculated by compensating the synchronization deviation time (t ₁ −t _n ) from the digital integration time t _n of the primary frequency from the zero cross point t ₁ of the power supply voltage, and accurately calculating the power supply. There is also an effect that the phase can be digitally detected. Next, the current command generation method of the present invention will be described. The current command generation method is performed by software processing of the microcomputer as shown in the IRQ1 software processing of FIG. 3. First, the active component current _Iq and the reactive component current _Iq
It calculates the I _d, so that I _d = 0 (power factor 1), by advancing the phase of the current command, and performs delay compensation in the current control system by the PWM switching-like.

Therefore, the method for detecting I _d and I _q will be described first. FIG. 6 shows a vector diagram of the power supply voltage and the power supply current. The phase voltage vector _{R of the} R phase is delayed by 30 ° from the line voltage vector _R of the RS phase. Therefore, the _R vector axis is defined as the q axis, and q
If the axis delayed by 90 ° from the axis is the d-axis, the d-axis component of the R-phase power supply current vector _R becomes the reactive current _Id , and the q-axis component becomes the active component current _Iq . Also, the actual power supply current is subject to current ripple due to the PWM switching of the converter,
As shown in FIG. 2, the filter 32 and the microcomputer A / D converter 2
Enter through three. As a result, if the phase is delayed with respect to the actual current and the total phase delay is Δθ _f , I _d and I _{q are} detected as shown in the block diagram of FIG. 7 in consideration of Δθ _f . I'm wearing The phase of the d-axis in consideration of the filter delay and the like is given by the following equation (5).

θ _d * = θ _RS * −2 / 3 π−Δθ _f (5) In addition, I _q and I _d are expressed by equations (6) and (7).

As described above, θ _d * is determined from the equation (5) in consideration of the phase delay Δθ _f obtained by adding the delay of the filter 23 and the delay of the input of the A / D converter, and based on the equation (6), From equation (7)
Since _Iq and _{Id are} detected, there is an effect that accurate _Id and _Iq can be detected.

 Next, an output method of a current command will be described.

When an AC power supply current command is given, the D / A shown in FIG.
Due to delays of the converter 30, the filter 31, the hysteresis comparator 11, etc., the actual current is delayed with respect to the current command. Therefore, in order to match the phases of the power supply voltage vector _R and the power supply current vector i _R and operate at a power factor of 1, it is necessary to advance the phase of the current command before. Therefore, in the present invention, I _q detection value as shown in FIG. 7 is at a positive, I _d detection value is determined from the [Delta] [theta] _{R *} to be zero (8).

Here, K _I is an integral gain.

Therefore, the current command phase of the R phase is given by equation (9), and Δθ
By performing the _R- lead compensation, it is possible to compensate for the delay of the current control system, so that there is an effect that a sinusoidal power supply current can be accurately passed at a power factor of 1. Even if Δθ _R * is previously measured and set as a constant, almost the same effect is obtained.

_{θ R * = θ RS * -π} / 6 + Δθ R * ... and (9), each phase of the current command is given by (10) (11) (12).

_{i R * = i q * ·} sinθ R * ... (10) i S * = i q * · sin (θ R * -2 / 3π) ... (11) i T * = i q * · sin (θ R * -4 / 3 [pi]) ... (12) where, i _q *, as shown in FIG. 2, the deviation of the converter output voltage (proportional + integral) is compensated output.

Next, another embodiment using a power supply current control method will be described with reference to FIGS.
This will be described with reference to the drawings. The power supply current i _R flows via the AC reactor 4 due to the difference voltage between the power supply voltage _R and the converter input voltage ₁ . Therefore, ₁ by the vector diagram of Figure 8
The dq-axis components V _d and V _q of the vector are given by equations (13) and (14).

Here, ω ₁ * is the power supply frequency, L is the inductance of the AC reactor, and r ₁ is the resistance between the power supply and the converter.
It can be approximated by r ₁ ≒ 0. From equation (13), in the steady state, r ₁ · I _d = ω ₁ * · L · I _q −V _d, so if I _d increases, I _d needs to be decreased, and if V _d is increased, I _d Becomes smaller. The same applies to I _q , and it is necessary to increase V _q as I _q increases. As a result, control may be performed as shown in FIG. 9 when r ₁ ≒ 0 and steady. In addition, by setting I _d * = 0, operation with a power factor of 1 is performed. FIG. 9 shows that (I _d * −I _d ) is compensated by (proportional + integral) compensation 33
The d-axis correction amount ΔV _d * is calculated via a and the limiter processing 34a.

Therefore, the d-axis voltage command V _d * is given by equation (15). Similarly, (I _q * −I _q ) is subjected to (proportional + integral) compensation processing 33b and limiter processing 34b to calculate the q-axis voltage correction amount ΔV _q *, and the q-axis voltage command V _q * is (16) It is given by the formula.

_{V d * = I q · ω} 0 · L-ΔV d * ... (15) V q * = V R * -I d · ω 0 · L-ΔV q * ... (16) In addition, the current response at the time of sudden load change 9 to improve
V _d * and V _q * in the figure may be calculated using (15) ′ and (16) ′.

Also, the magnitude of the input voltage vector ₁ of the converter | V
₁ | * and the phase θ _P * are obtained by the voltage vector calculation processing 35 (17),
Calculated from equation (18).

Therefore, space vector PWM processing 3 based on | V ₁ | * and θ _P *
The gate signal of the converter is given by calculating the pulse width of the PWM signal by 6 and setting the pulse change time in the microcomputer built-in timer 37.

Next, the space vector PWM processing will be described with reference to FIGS.

As shown in FIG. 10, when the three-phase converter, the converter input voltage vector is located 8 kinds 2 ^3, _R direction vector Vector. here The vector is a state where the positive arm of the converter is on in the R phase, off in the S phase, and off in the T phase. Therefore, the conduction time is controlled by selecting the voltage vector shown in FIG. 11 according to the rotation of the converter input voltage vector ₁ . The conduction time of the voltage vector is given by the equations (19) and (20).

Here, K [alpha is a constant determined by the PWM carrier cycle T _C, V _DC is the converter output voltage. Therefore, the 11 theta _{P *} is 0 to 60 ° interval as shown in FIG. Giving the switching pattern of T _A time, Given the switching pattern of T _B time, T _C −T _A −T _B time, zero voltage vector give. In the section where θ _P * is between 60 ° and 120 °, Vector is T _A time, Vector is T _B time.

In the other embodiments described above, since the control device of the converter can be entirely digitalized with only one chip microcomputer, there is an effect that the control device is small, light, and inexpensive. In addition, since the magnitude and phase of the converter input voltage vector ₁ can be instantaneously controlled, there is an effect that the response of the power supply current is fast. Further, by performing the space vector PWM processing, the converter input voltage can be increased as compared with a general sine wave and triangular wave comparison PWM method. So, conversely,
When the power supply voltage V _R is constant, the lower limit of the set value V _DC * of the converter output voltage (smoothing capacitor voltage) can be reduced, which has the effect of widening the control range of the converter output voltage. Next, another embodiment for digitally comparing an AC power supply current command and a detected value will be described with reference to FIG. That is, a three-phase power supply current command is calculated from the equations (21), (22), and (23) by software processing started every fixed period T _S.

i _R * = I _q * · sin (θ _RS * −π / 6)… (21) i _S * = I _q * · sin (θ _RS * −5 / 6π)… (22) i _T * = − ( i _R * + i _S *) also ... (23), the supply current is detected through a direct microcontroller built a / D converter without passing through filter 32. Next, when the power supply command value is larger than the detected value, the positive arm of the converter is turned off to reduce the converter input voltage. On the other hand, when the current command value is smaller than the detection value, the positive arm is turned on to increase the converter input voltage. Thus, this method
As shown in the software processing of Fig. 12, the sampling period is simple
T _S can be made very short. As a result, the switching frequency of the converter can be increased, and the current ripple can be reduced. Also, because of the on / off control, there is also an effect that all the digitalization can be performed with a one-chip microcomputer without the need for a timer element. FIG. 13 shows a general-purpose inverter system to which the above-described converter control device is applied.

This is a system in which an induction motor 39 controlled at a variable speed by an inverter 38 is connected as the load in FIG. Here, the inverter receives the primary frequency command ₁ *, and the inverter control device 40 controls the inverter output frequency and output voltage. The converter control device 41 is a control device using the one-chip microcomputer described in the above embodiments. A general system of a general-purpose inverter uses a three-phase full-wave rectifier circuit as a substitute for the converter shown in FIG. 13, and the low-order harmonic of the power supply current is extremely large, which is a source of noise to other devices. Also, the DC voltage of the smoothing capacitor 6 becomes constant at 270 V when receiving 200 V, and the maximum inverter output voltage becomes
It is as small as 180V. For this reason, in order to increase the maximum torque of the motor, it is necessary to increase the motor current, and the motor efficiency deteriorates. In addition, it is necessary to shift from a PWM inverter to a square wave inverter in order to increase the inverter output voltage to 200 V in a high-speed region, and the motor current waveform is distorted. Also, the switching frequency is reduced, so that the noise is also increased. Thus, by using a converter in a general-purpose inverter system as shown in FIG. 13, the power supply current can be made sinusoidal, and low-order harmonics can be reduced. In addition, since the DC voltage _VDC can be controlled to 270 V or more, the maximum inverter output voltage can be increased. As a result, the motor current can be reduced, thereby improving the efficiency. Furthermore, inverter output voltage up to 200V or more
PWM control can be performed, which has the effect of reducing torque ripple and noise even in a high-speed region.

Further, the converter control device controls the DC voltage _{VDC to} be constant. By increasing the speed of the current control system of the converter, the capacity of the smoothing capacitor 6 can be reduced, resulting in an inexpensive system. Further, by converting the converter control device 41 into a microcomputer as in the present invention, communication between the converter and the inverter is simplified. As a result, there is an effect that both parties can easily cooperate in protection.

〔The invention's effect〕

Since the present invention is configured as described above, it has the following effects.

Since the power supply voltage phase can be detected with high accuracy using only the photocoupler and the one-chip microcomputer, the control device is small, light, and inexpensive. In addition, the reactive component current and the active component current can be accurately detected in consideration of the delay of the power supply current detection, and the AC current command is advanced and compensated so that the reactive component current becomes zero.
A sinusoidal power supply current with a low power factor and a small low-order harmonic can be passed accurately with a power factor of one. In addition, a high-speed current control system is realized by instantaneously controlling the magnitude and phase of the converter input voltage vector such that the reactive component current is zero and the active component current command follows the active component current command using only one chip microcomputer. And the fluctuation of the converter output voltage can be reduced. As a result, the capacity of the smoothing capacitor can be reduced. In addition, the converter control circuit can be completely digitalized, resulting in a compact and low-cost device. In addition, in the all digital method in which the magnitude of the AC power supply current command value and the detected value are compared at regular intervals and the result is used as a gate signal, the switching frequency of the converter can be increased, and simple on / off control is performed. It becomes a control system and the control device becomes inexpensive. Further, in the system in which the inverter for variable speed AC motor is connected as a load to the converter described above, the maximum output voltage of the inverter can be increased, so that the motor can have high efficiency, low torque ripple, and low noise. In addition, by using a microcomputer for the converter control device, communication processing with the inverter control device is simplified, and protection and coordination between the two are facilitated.

[Brief description of the drawings]

FIG. 1 is a control block diagram showing one embodiment of the present invention, and FIG.
FIG. 3 is a hardware configuration diagram of one embodiment of the present invention, FIG. 3 is a flowchart of software processing of the microcomputer shown in FIG. 2, FIGS. 4 and 5 are explanatory diagrams of power supply voltage phase detection of the present invention,
FIG. 6 is a power supply voltage and power supply current vector diagram, FIG. 7 is a detailed block diagram of active component current and reactive component current detection shown in FIG. 2, and delay compensation processing of the current control system, and FIG. Voltage vector diagram, FIG. 9 is a control block diagram of another embodiment, and FIGS. 10 and 11 are space vectors PW shown in FIG.
FIG. 12 is an explanatory diagram of the M process, FIG. 12 is a flowchart of a soft process showing another embodiment, and FIG. 13 is a diagram of a general-purpose inverter system to which the present invention is applied. 1 ... AC power supply, 4 ... AC reactor, 5 ... Converter, 6 ... Smoothing capacitor, 8 ... Power supply voltage phase detecting means, 18 ... Photo coupler, 22 ... 1 chip microcomputer,
27: Effective component current / inactive component current calculation processing, 28: Current control system delay compensation processing, 32: Filter, 35 ... Converter input voltage vector calculation processing.

──────────────────────────────────────────────────続 き Continuation of the front page (72) Inventor Hiroshi Fujii 7-1-1, Higashi-Narashino, Narashino-shi, Chiba Pref. 62-95972 (JP, A) JP-A-58-79478 (JP, A) (58) Fields investigated (Int. Cl. ⁶ , DB name) H02M 7/00-7/40

Claims (4)

(57) [Claims] 1. A converter for converting an AC power supply to a DC power.
A control device for a power converter that controls a power supply power factor to be 1 according to a power supply voltage phase command value, wherein a power supply voltage phase error is detected from a power supply voltage phase command value at a zero cross point of a power supply voltage of the AC power supply. Then, a correction amount of the frequency command is obtained in accordance with the phase error, a primary frequency obtained by adding the correction amount and a reference power supply frequency command is integrated, and an output thereof is used as the power supply voltage phase command value. Power converter control device. 2. A method of digitally obtaining a power supply voltage phase command value at a power supply voltage zero crossing point according to claim 1, wherein an approximate power supply voltage zero crossing point is detected from a power supply via a photocoupler. The power supply voltage phase command value at the power supply voltage zero crossing point is determined in consideration of the operation delay time ΔT of the photocoupler and the difference between the power supply voltage phase command output time output at regular intervals and the change time of the photocoupler output. A control device for a power converter, comprising: 3. A control apparatus of a power conversion system according paragraph 1 claims, based on the power supply voltage phase command value, determine the active current I _q and the reactive current I _d of the power supply current, the converter output The effective component current command compensated so that the voltage command and the converter output voltage coincide with each other, and the phase of the AC power supply current command created based on the power supply voltage phase command is used as the reactive component current detection value.
I _d is the controller of the power converter is characterized in that lead compensation and to become zero. 4. The reactive current I _{d according to} claim 3.
As a method of detecting the effective component current _Iq, the phase obtained by subtracting the delay phase amount of the current detection filter from the power supply phase voltage phase command value and the delay phase amount from the power supply voltage phase calculation time to the current detection time is used as a reference phase. A power converter control device characterized in that a power supply current is decomposed into two rotational coordinate axes (dq axes) and detected.