DE102019208559A1 - Operation of switching elements of an inverter - Google Patents

Abstract

The invention relates to a method for operating switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) of an inverter (12) to which a polyphase stator winding of an electrical machine (14 ) is connected, with switching signals being determined as a function of an operating state indicator (84), the operating state indicator for a number of directly dependent on a period duration of a fundamental oscillation of phase voltages and the number of phases (U1, U2, V1, V2, W1, W2) successive clock periods (T) is determined in such a way that for a period of time corresponding to a control angle range (v) one of the phases is electrically coupled to one of two intermediate circuit potentials of the DC voltage intermediate circuit (46) Modulator phase angle (γ) is set within an adjustment range (κ), for which purpose a load on an intermediate circuit capacitance of the G light voltage intermediate circuit (46) is determined and the modulator phase angle (γ) is determined depending on the determined stress.

Description

The invention relates to a method for operating switching elements of an inverter to which a polyphase stator winding of an electrical machine is connected, the inverter having at least one series connection of the switching elements for each of the phases of the stator winding in order to electrically connect the stator winding with a DC voltage intermediate circuit connected to the inverter to be coupled by applying switching signals for the respective switching elements, which are dependent on the respective phase voltages of the respective phases, to the switching elements of the series connections, the switching signals being determined based on a clock signal using a discontinuous pulse width modulation, a clock period of the clock signal being smaller than one oscillation period of a fundamental oscillation of the phase voltages, the switching signals being determined as a function of at least one operating state indicator, which in turn depends on du The switching state vectors that can be achieved by the switching elements of the inverter are determined in a corresponding space vector system, the operating state vector being determined for a number of immediately consecutive clock periods dependent on the period duration of the fundamental oscillation and the number of phases in such a way that a control angle range with respect to the Period duration of the fundamental oscillation corresponding time duration of these clock periods at least one of the phases is electrically coupled by means of at least one respective one of the switching elements to one of at least two intermediate circuit potentials of the DC voltage intermediate circuit. The invention also relates to a control device for operating switching elements of an inverter to which a polyphase stator winding of an electrical machine is connected, the inverter having at least one series circuit of the switching elements for each of the phases of the stator winding to connect the stator winding to a DC voltage intermediate circuit connected to the inverter to be electrically coupled, the control device being designed to apply switching signals for the respective switching elements that are dependent on the respective phase voltages of the respective phases to the switching elements of the series circuits, to determine the switching signals based on a clock signal using a discontinuous pulse width modulation, with a clock period of the clock signal being smaller as an oscillation period of a fundamental oscillation of the phase voltages is to determine the switching signals as a function of at least one operating state indicator, which the control device in turn determines in a corresponding space vector system as a function of the switching states of the inverter that can be achieved by the switching elements, in that the operating state indicator is determined for a number of immediately consecutive clock periods that is dependent on the period of the fundamental and the number of phases in order to provide a To electrically couple at least one of the phases by means of at least one respective of the switching elements with one of at least two intermediate circuit potentials of the DC voltage intermediate circuit in relation to the period of these clock periods corresponding to the period duration of the fundamental oscillation. Finally, the invention relates to an inverter for connection to a polyphase stator winding of an electrical machine, the stator winding being designed according to the number of phases of the electrical machine, with at least one series connection of the switching elements for each of the phases of the stator winding to the electrical machine with to electrically couple a DC voltage intermediate circuit that can be connected to the inverter by providing respective phase voltages for the respective phases of the stator winding depending on switching signals for the respective switching elements in normal operation by means of the series connections, and a control device for providing the switching signals.

Methods of the generic type, control devices of the generic type and also inverters of the generic type are known extensively in the prior art, so that there is basically no need for separate printed evidence for this. In the prior art, these are used for a large number of applications in which a respective specific drive function is to be implemented by means of a multi-phase electrical machine. The use of three-phase electrical machines, which - depending on the application - can even be directly coupled to a three-phase public power supply network in order to be able to implement the respectively specified drive function, is particularly widespread.

The control options are limited in such a case, which is why multi-phase electrical machines are now operated via respective inverters, which are electrically coupled to a DC voltage intermediate circuit for this purpose. The DC voltage intermediate circuit, for its part, can be electrically coupled to the public energy supply network or to another energy source or energy sink. This makes it possible to obtain the electrical energy required for the intended operation of the electrical machine not only from the public energy supply network, but also from another energy source such as an electrical one Provide energy storage, for example a battery, but also a fuel cell, a wind generator, a photovoltaic module and / or the like. In addition, it can be achieved that drive systems can be implemented which are not only suitable for stationary applications, but which are also suitable for mobile use, for example in motor vehicles or the like.

The use of an inverter makes it possible to control the multiphase electrical machine in a flexible manner, even if the electrical machine has more than two or three phases, for example four phases, five phases, six phases or even more. Even if a predominant application of three-phase electrical machines is currently widespread, it is provided for specific applications to use electrical machines with more than three phases, for example in ship propulsion systems or the like. In specific applications, however, it can also be provided that the stator winding of the electrical machine is only designed in two phases.

The use of the inverter makes it possible to operate the multiphase electrical machine in a very dynamic manner.

For this purpose, provision is generally made for phase voltages to be provided for the respective phases, which serve as setpoint values for controlling the inverter. Depending on these phase voltages, the switching signals for the respective switching elements can then be determined by means of the discontinuous pulse width modulation. Operating conditions of the electrical machine can be set with the phase voltages. For this purpose, for example, a frequency, an amplitude, harmonics and / or the like can be specified for the phase voltages. To provide the phase voltages, a phase voltage generator can be provided, for example, which specifies the phase voltages according to an operating state to be achieved by the electrical machine. This can be designed as a pure control. In addition, a control can also be provided that preferably takes into account one or more operating parameters of the electrical machine, for example an instantaneous speed, a current angle of rotation, in particular during start-up, an electrical power, an electrical current of the stator winding and / or the like.

A frequently used option to provide the phase voltages, for example, uses a vector control, which - depending on the design - is sometimes also referred to as vector control and which for the three-phase case is based on a Clarke transformation and also a d / q transformation, also Park -Transformation called, based. The Clarke transformation is used to transform three-phase variables of the electrical three-phase machine into a two-axis coordinate system which has reference axes α and β. In this way, vector control or vector control can be implemented in which space vector representations can be used to set the operating states of the inverter. The Park transformation is related to the Clarke transformation and differs from the latter in particular in that the two-axis coordinate system has reference axes d and q, whereby in the Park transformation the d / q coordinate system is based on a machine size in a stationary operating case rotating three-phase electrical machine. In this case, a respective pair of values composed of values d, q can represent a variable that is constant over time for this operating state.

For multi-phase electrical machines that have more than three phases, corresponding vector controls or vector controls can be used, as they are, for example, by the US 2011/0221365 A1 are disclosed.

In addition, the phase voltages can also be provided by means of other methods and / or devices, for example by specifying an external multiphase AC voltage network on which the phase voltages are sampled and used as a reference for the discontinuous pulse width modulation, by digital voltage specification of the phase voltages, in particular using tables or using computer units that, for example, carry out a sine calculation themselves, or also via analog hardware circuits, such as an oscillator, a digital specification via a system model of the electrical machine or a simulation of a system model with an analog hardware circuit, a Direct torque control, in particular with or without hysteresis, as specified, for example, at https://de.mathworks.com/help/phvsmod/sps/powersvs/ref/directtorquecontroller.ht ml, or the like.

Overall, the state of the art has proven itself. However, there remain disadvantages. This is because it has been shown that, in particular with regard to the DC voltage intermediate circuit, a ripple current load can be unfavorable, particularly with regard to the stress on an intermediate circuit capacitor.

In order to be able to achieve an improvement here, it is, for example, from the JP 2001-197779 A known to provide a stator winding of a three-phase electrical machine with two electrically separated winding systems and to connect this machine to a correspondingly designed inverter and operate it. The inverter can apply a respective phase voltage to each of the phases of each of the winding systems separately. The inverter is in turn connected to a DC voltage intermediate circuit.

For the two winding systems, according to JP 2001-197779 A the switching signals for the switching elements are determined by means of a vector control. For one of the two winding systems, however, the switching signals for the switching elements are delayed by a fixed predetermined period of time. As a result, a reduction in a ripple current amplitude at the DC voltage intermediate circuit should be able to be achieved.

However, this teaching shows that the implementation of a carrier signal shift or switching signal shift according to JP 2001-197779 A requires at least one carrier signal per winding system, which leads to a significant increase in the complexity of hardware and software resources. In addition, a combination of carrier signal shift and variable switching frequency or variable clock period, which is used in modern inverters, for example to optimize the efficiency of the overall system, is very difficult.

It also proves to be problematic that the winding systems are inductively coupled via the common laminated core, so that the DC intermediate circuit can be relieved through undesired coupling when the carrier signal shift is used, but at the same time significantly more losses are introduced into the electrical machine due to an increased distortion component of the phase currents which, depending on the coupling, can lead to increased overall losses.

The invention is therefore based on the object of improving a control option for a polyphase electrical machine with a polyphase stator winding, which preferably has at least two winding systems that are electrically separated from one another.

As a solution, the invention proposes a method, a control device and an inverter according to the independent claims.

Advantageous developments result from the features of the dependent claims.

With regard to a generic method, the invention proposes in particular that the control angle range is set as a function of a modulator phase angle within an adjustment range, for which purpose a load on an intermediate circuit capacitance of the DC voltage intermediate circuit is determined and the modulator phase angle is determined as a function of the determined stress.

With regard to a generic control device, it is proposed in particular that the control device is designed to set the control angle range as a function of a modulator phase angle within a setting range, to determine a load on an intermediate circuit capacitance of the DC voltage intermediate circuit, and to determine the modulator phase angle as a function of the determined stress.

With regard to an inverter of the generic type, it is proposed in particular that the control device is designed according to the invention.

The invention is based, among other things, on the idea that with discontinuous pulse width modulation (DPWM) at least one of the phases of the electrical machine is electrically coupled to one of the at least two intermediate circuit potentials for the number of immediately successive clock periods, which results in the setting range that is determined by the Number of phases depends. The setting range usually denotes an angle range within which the control angle range can be positioned. The adjustment range is therefore usually larger than the control angle range. The position of the control angle range within the setting range can be determined by means of the modulator phase angle or the control angle range can be positioned within the setting range as a function of the modulator phase angle. The setting range is particularly dependent on the number of phases or, in the case of several winding systems, dependent on the number of phases of a respective one of the winding systems.

The invention uses the advantage that there is no need to intervene in the provision of the phase voltages for the control or regulation of the polyphase electrical machine. With regard to vector control, this means that only a single vector control or vector control needs to be used. With this vector control or vector regulation, the phase voltages can first be determined in relation to the respective phases, on the basis of which the voltages determined by means of the discontinuous pulse width modulation can be determined Switching signals could in principle already be used for the operation of the switching elements, but then, depending on the structure of the winding or the winding systems, may result in a high level of stress on the intermediate circuit capacitor.

The invention thus makes it possible, in particular, to use only a single clock signal, as a result of which all of the required currents can be recorded essentially simultaneously with the aid of current sensors in the manner of a low-side shunt current measurement. This saves effort and costs.

In particular, an adjustment period can be assigned to the adjustment range depending on the period duration of the fundamental oscillation. The greater the number of phases, the smaller the adjustment range or the adjustment period. The immediately consecutive clock periods are positioned in time within the setting period. In the case of three phases, the setting range can for example include a range from 0 ° to approximately 120 ° or from approximately -60 ° to approximately + 60 °.

The adjustment period is usually greater than the number of immediately consecutive clock periods, which results in degrees of freedom with regard to the positioning of the immediately consecutive clock periods within the adjustment period. The position in time can be determined by the modulator phase angle. This makes it possible to influence the actual temporal position as a function of the stresses on the DC voltage intermediate circuit through individual phase stresses, whereby the stress on the DC voltage intermediate circuit, in particular the intermediate circuit capacitor, can be influenced. With the invention it is thus possible to act on the stress on the DC voltage intermediate circuit by suitable determination of the modulator phase angle, so that, for example, a ripple current load can be reduced.

Therefore, according to the invention, the load on the intermediate circuit capacitance is determined. The modulator phase angle is then determined depending on the load on the intermediate circuit capacitance. As a result, the load on the intermediate circuit capacitance can be adjusted by suitable determination of the modulator phase angle. It can be provided that a comparison value for the load on the intermediate circuit capacitance is specified and the modulator phase angle is determined as a function of a comparison result. As a result, the invention does not need to be activated permanently, but can be activated as required, for example if the load on the intermediate circuit capacitance is greater than the comparison value. The modulator phase angle is preferably determined in such a way that the stress on the intermediate circuit capacitance is as small as possible, in particular minimal. For this purpose, for example, certain circuit topologies and / or control methods of the inverter and / or the electrical machine can be taken into account.

The modulator phase angle does not therefore need to be constant, but can, for example, be changed or also varied as required. For example, the modulator phase angle can be determined differently for each clock period. Adjustment patterns for the modulator phase angle with respect to successive clock periods or the like can also be provided, by means of which the modulator phase angle can be changed, preferably as a function of a control variable.

In addition, the modulator phase angle can of course also be dependent on a degree of modulation or a modulation index. This makes it possible to take into account that an intervention according to the invention is only required for a predeterminable range of the degree of modulation or the modulation index. For example, it has been shown that an intervention with a degree of modulation or a modulation index of approximately 0.15 to approximately 0.85 can be expedient. Outside of this range, the process control or control according to the invention can be deactivated.

The modulator phase angle is preferably determined to be the same for all phases. However, if necessary, for example in the case of specific operating states, this can also be deviated from, so that the modulator phase angle can be determined differently for one or more phases. For example, it can be determined differently, in particular slightly, between different phases. Provision can be made for the modulator phase angle to toggle around a predetermined mean value or the like in relation to successive clock periods.

Overall, however, it has been shown that a duty cycle, that is to say a ratio of an active switching state vector to the clock period, in particular for a respective individual clock period, can be maintained. With the invention it can consequently be achieved that - without influencing the intended operation of the electrical machine - it is possible to act on the ripple current and consequently also on the loading of the intermediate circuit capacitance.

Since no change needs to be made to the inverter, in particular its power unit with the switching elements, for the invention, the invention can also be retrofitted in a simple manner in inverters that have already been designed, in particular taking into account the respective electrical machines. The invention can therefore be implemented in a particularly simple manner, even in the case of control devices which are already completely constructed. The invention is therefore also suitable, inter alia, for retrofitting in already existing control devices or inverters. As a result, the invention can be implemented in a particularly flexible and cost-effective manner.

The invention can use the advantage that, for the control or regulation of the multiphase electrical machine, basically only a single control variable needs to be acted upon, namely the modulator phase angle.

Furthermore, the invention is based in particular on the use of discontinuous pulse width modulation. The discontinuous pulse width modulation is described in, for example "A High-Performance Generalized Discontinuous PWM Algorithm" by Ahmet M. Hava in IEEE Transactions on Industry Applications, Vol. 34, No. 5, September / October 1998 .

The invention therefore makes it possible, in particular, to use only a single clock signal, as a result of which all of the required currents can be recorded essentially simultaneously with the aid of current sensors in the manner of a low-site shunt current measurement. This saves effort and costs.

In addition, the control unit can also be implemented cost-effectively because, for example, only a single timer unit needs to be provided for a computer unit or a microcontroller. In particular, essentially simultaneous detection and processing of phase currents of the respective phases of the stator winding can be achieved in a simple manner and with little effort, because preferably both machine-related and coupling-related harmonics can be compensated.

The invention also makes it possible to be implemented in a simple manner, for example by means of a computer program which can be executed by a computer unit of the control device. Finally, the invention can make it possible in particular to optimize stress on the DC voltage intermediate circuit, voltage ripple and also system efficiency for almost every single clock period, which can result in particularly great advantages, particularly in dynamic systems such as electric drives.

The invention can therefore be provided for inverters that operate electrical machines with a stator winding with at least two phases. The invention is not limited to electrical machines with two phases. It can of course also be used in particular in electrical machines with at least three phases. The stator winding can be star-shaped, triangular or n-corner-shaped or the like. The stator winding is preferably only subjected to electrical potentials from the inverter. For the invention it can also be noteworthy that individual windings of the stator winding can be magnetically coupled to one another by the electrical machine. The above statements preferably also apply to stator windings in a case in which the stator winding comprises two or more winding systems.

Specific applications can also be found in the field of energy supply, for example in the electrical charging of electrically driven motor vehicles, further particularly in almost any type of electrical drive, in particular in the field of aviation, seafaring and / or the like or in the field of industry, for example be provided in the case of a conveyor device such as conveyor belts or the like.

The invention also makes it possible to provide great flexibility with regard to changes in the clock period or the clock rate or the clock frequency of the clock signal for an optimization of efficiency and also with regard to undesirable noise development in vehicles or noise vibration harshness (NVH) which could otherwise be problematic, especially with different clock signals.

In addition, the invention makes it possible, in particular, to reduce the stress on the DC voltage intermediate circuit, especially that of the intermediate circuit capacitor, in that, for example, in the case of star-shaped winding systems, zero point voltages of the individual respective winding systems can be set in relation to one another as required. Of course, this is not limited to stator windings or winding systems embodied in a star shape.

The phase voltages are alternating voltages that also occur indirectly at the respective phase connections of the inverter or the electrical machine and consequently relate to voltages at the respective phase connections of the inverter to which the stator winding of the electrical machine is connected. For each of the phases of each of the stator winding of the polyphase electrical machine, the inverter provides at least one phase connection to which the respective phase of the stator winding of the electrical machine can be connected. For each such phase connection, the inverter provides at least one series circuit of the switching elements, which in turn is usually connected to the DC voltage intermediate circuit to which the inverter is also connected. The respective center taps of the series connections provide the phase connections of the inverter. The series connections are therefore connected in parallel to the DC voltage intermediate circuit.

As a rule, respective phase currents of the stator winding are set by respective PWM patterns of the switching signals provided by the control device, with which the respective switching elements of the respective series connections are applied. The phase voltages can now be determined from these PWM patterns by, for example, filtering the respective PWM pattern using a suitable low-pass filter, determining a Fourier transformation, determining the respective phase voltage using a power meter, and / or the like. The phase voltage therefore does not correspond to the immediate PWM pattern which is provided directly at the respective phase connection of the inverter. For this reason, the frequency of the fundamental oscillation of a respective one of the phase voltages is generally lower than the clock rate or the clock frequency of the switching signals according to which the respective PWM patterns are provided at the respective phase connections.

The inverter is a form of an electrical energy converter that couples the DC voltage intermediate circuit with an AC voltage network in terms of energy, so that electrical energy can be exchanged between the AC voltage network and the DC voltage intermediate circuit. The AC voltage network is formed here by the phase voltages. For this purpose, provision is generally made for at least one series connection of at least two switching elements, which are electrically coupled to the DC voltage intermediate circuit, to be provided for a respective phase of the AC voltage network. The respective phase of the AC voltage network can be connected to the center taps of the series connections, in this case the respective phase of a respective one of the winding systems of the electrical machine.

In the case of coupling the DC link via the inverter to the at least two-phase electrical machine, the respective phase connections of the electrical machine, called phases for short, are connected to the respective center taps of the series circuits that provide the phase connections of the inverter. By suitably controlling the switching elements by means of the switching signals provided by the control device, the desired energy conversion function of the inverter can be achieved, so that the desired drive function of the electrical machine can also be achieved.

For this purpose, the switching elements are operated with the switching signals on the basis of the clock signal in a switching mode, with a clock rate of the clock signal generally being considerably greater than the frequency of the fundamental oscillation of the phase voltages or the alternating phase currents of the electrical machine. In the present case, the phase voltages are therefore alternating voltages. The desired energy-related coupling can then be established by means of certain control methods such as pulse width modulation (PWM) or the like. The control device provides the switching signals for this operation. The switching signals can preferably be provided individually for each of the switching elements. A predetermined exchange of energy can thus be achieved by means of the inverter by means of suitable pulse patterns of the switching signals for the switching elements.

The clock rate or the clock frequency can be essentially constant in normal operation. However, depending on requirements, it can also vary or be changed as a function of one or more parameters.

A switching element within the meaning of this disclosure is in particular a controllable electronic switching element, for example a controllable electronic semiconductor switch such as a transistor that is operated in a switching mode, a thyristor, combination circuits thereof, preferably with parallel-connected inverse diodes, a gate turn-off thyristor (GTO ), a
Insulated gate bipolar transistor (IGBT), combinations thereof or the like. Basically, however, the switching element can also be a field effect transistor, in particular a
Metal oxide semiconductor field effect transistor (MOSFET) be formed.

To provide the desired conversion function by the inverter, the switching elements are operated in switching mode. With regard to a semiconductor switch using a transistor, switching operation means that, when the switching state is switched on, a very small electrical resistance is provided between the terminals of the transistor that form a switching path, so that a high current flow is possible with a very small residual voltage. In a switched-off switching state, on the other hand, the switching path of the transistor is highly resistive, that is, it provides a high electrical resistance so that there is essentially no or only a very low, in particular negligible, current flow even when there is a high electrical voltage applied to the switching path. Linear operation differs from this in transistors, but this is generally not used in inverters of the generic type.

To implement the desired function, the switching elements each have at least one control connection in addition to the switching path, at which they can be acted upon with the switching signal provided by the control device, so that the desired switching function of the switching element can be achieved. The switching signal can be a binary switching signal which can assume two status values in order to be able to achieve the desired switching functions of the switching element. For example, the switching element can be formed by a pulse train by means of which the control connection is acted upon. This is particularly useful with thyristors such as GTO or the like. In addition, in the case of transistors, provision can be made for the switching signal to be a square-wave signal, with a respective switching state of the switching element being able to be assigned to one of the potentials of the square-wave signal. Such a signal is useful, for example, for transistors, in particular for bipolar transistors, field effect transistors or the like.

In addition to generating the switching signals, the control device can also provide further functions, in particular with regard to the inverter, for example monitoring functions, safety functions and / or the like. For this purpose, the control device can comprise a hardware circuit and / or a program-controlled computer unit or the like. Of course, the control device can be designed as a separate assembly. The control device can, however, also be at least partially comprised by a higher-level controller of an application for a drive system. The control device can also be at least partially encompassed by the inverter.

The inverter is an electronic hardware circuit which, in accordance with the number of phases of the winding systems, has the series connection of the preferably two switching elements. The series connections are generally connected in parallel to the at least one DC voltage intermediate circuit. The center taps of the series connections provide the phase connections to which the respective phases of the electrical machine can be connected. For example, it can be provided that the inverter has precisely one series connection of the switching elements for each phase, so that respective half-bridge circuits can be implemented. In addition, it can also be provided, for example, that the inverter has two series connections made up of the switching elements for each phase, whereby respective full-bridge connections can be implemented. The invention can also be provided for other topological structures on the inverter side without losing the effect of the invention.

The at least two-phase electrical machine can be designed as a synchronous machine, as an asynchronous machine, as a double-fed asynchronous machine or the like. Combinations of these can of course also be provided in order, for example, to be able to implement certain possible applications that make it desirable to influence the magnetic flux.

The DC voltage intermediate circuit can be coupled to an electrical energy source, which can be formed by an electrical energy store, such as an accumulator or the like. In addition, it is of course possible for the DC voltage intermediate circuit to be electrically coupled to a public power supply network, which is designed as an AC voltage network, via a rectifier unit. Combinations of these can also be provided. In addition, further electrical energy sources can also be coupled to the DC voltage intermediate circuit in a predefinable manner, for example photovoltaic systems, wind turbines and / or the like.

A space vector system is used in the present case to represent the phase voltages occurring when processing corresponding data by the control device, so that the phase voltages of the at least two-phase system can be represented. Depending on the number of phases, a Clarke transformation can be used for this purpose if the number of phases is three or, in the case of a higher-phase electrical machine, a Fortescue transformation or a derivative thereof can be used. When using a Fortescue transformation, one dimension can be a corresponding Matrix can be chosen depending on the number of phases.

The space vector system is preferably mapped in the complex plane. In order to determine the operating state indicator, certain switching state vectors are mapped in the space indicator system by the switching elements of the inverter. The operating state pointer is then determined as a function of a respectively set operating state using the switching state vectors. The switching signals can then be determined on the basis of the operating status indicator. For this purpose, the control device provides corresponding signal processing units that can also implement mathematical transformations and the like, for example.

In a first step, the load on the DC link capacitance can be determined. Furthermore, for example, a degree of modulation can be determined or also set. The modulation phase angle is then determined as a function of this, on the basis of which the switching signals can then be determined. This can be provided for each individual clock period or also for a predetermined plurality of clock periods.

Overall, it can be achieved that the modulation phase angle causes a time offset or an angular offset of the currents on the DC voltage intermediate circuit, so that the ripple currents individually caused by the phases of the stator winding on the DC voltage intermediate circuit can at least partially compensate for each other. This can also reduce the load on the intermediate circuit capacitor.

The determination of a stress on the intermediate circuit capacitor is disclosed in, for example "Analytical calculation of the RMS current stress on the DC-link capacitor of voltage-PWM converter systems" by JW Kolar and SD Round in IEE Proc.-Electr. Power Appl., Vol. 153, No. 4, July 2006 in the case of a three-phase electrical machine. The stress can be determined accordingly for a higher number of phases. This disclosure is intended to be encompassed by this document.

The control device can be designed to implement vector control or vector control based on discontinuous pulse-width modulation (DPWM). The control unit can preferably be designed to implement vector control or vector control based on generalized discontinuous pulse-width modulation (GDPWM), as indicated above by the description by Ahmet M. Hava.

Depending on the application, the clock rate can be, for example, about 5 kHz, about 10 kHz, preferably about 20 kHz or even more. The clock rate can be chosen to be constant. Depending on the application, however, it can also be provided that the clock rate is variable and can be dependent, for example, on a phase position of a respective phase voltage. In addition, other manipulated variables can of course also be taken into account for setting the clock rate.

It is also proposed that the switching signals for all switching elements are based on the same clock signal. This makes it possible to avoid undesirable reactions, as can occur in the prior art, in particular in the case of inductive coupling of the windings of the winding systems. In addition, a simple setting option with regard to the clock frequency is possible because the individual switching signals only need to be determined in relation to the current clock frequency. As a result, any adaptation of the switching signals to a changed clock frequency can be implemented very easily.

wobei m der Modulationsgrad, U_DC die Zwischenkreisgleichspannung und û_ph eine Amplitude einer der Phasenspannungen ist. Die Phasenspannung wird vorliegend in Bezug auf die halbe Zwischenkreisgleichspannung ermittelt. Dem Grunde nach kann der Modulationsgrad natürlich auch anders in geeigneter Weise definiert sein. Beispielsweise kann vorgesehen sein, dass zum Bestimmen des Modulatorphasenwinkels der Modulationsgrad nur in einem vorgegebenen Bereich des Modulationsgrads berücksichtigt wird.It is also proposed that the modulator phase angle be determined as a function of a degree of modulation. The degree of modulation can preferably be defined as follows: $$\text{m}=\frac{\widehat{\text{u}}\text{ph}}{U_{\mathrm{D.}\mathrm{C.}}/2}\frac{\pi }{\mathrm{4th}}$$

where m is the degree of modulation, U _{DC is} the _DC link _voltage and û _{ph is} an amplitude of one of the phase voltages. In the present case, the phase voltage is determined in relation to half the DC link voltage. Basically, the degree of modulation can of course also be defined differently in a suitable manner. For example, it can be provided that, in order to determine the modulator phase angle, the degree of modulation is only taken into account in a predetermined range of the degree of modulation.

According to a development, it is proposed that the modulator phase angle be determined as a function of the fundamental oscillation. For this purpose, for example, an amplitude and / or a period duration of the fundamental oscillation can be taken into account in order to determine the modulator phase angle.

It is also proposed that the modulator phase angle be dependent on a ripple current amplitude of an intermediate circuit capacitor of the DC link is determined. This makes it possible to regulate the ripple current amplitude at the DC voltage intermediate circuit in that the modulator phase angle is specified as a function of the amplitude of the ripple current. This makes it possible to keep the ripple current amplitude in a predetermined range even with changing operating conditions, in particular to reduce the ripple current amplitude as much as possible. This has a particularly advantageous effect on the intermediate circuit capacitor of the DC voltage intermediate circuit, the load on which can be reduced accordingly. This makes it possible to reduce effort and costs with regard to the intermediate circuit capacitor. A capacitor current, for example, can be recorded as the ripple current. Alternatively, however, an alternating current of the current can also be detected as the ripple current, which is caused by the superposition of the currents of the series connections of the inverter on the intermediate circuit side, or the like.

In a further development, it is proposed that the modulator phase angle be determined as a function of an amplitude of at least one harmonic with respect to the fundamental oscillation. This makes it possible to take into account that the modulator phase angle can also be used to influence at least one harmonic with respect to the fundamental oscillation. As a result, the modulator phase angle can also be determined in such a way that not only the stress on the intermediate circuit capacitance is reduced, but at the same time an increase in the amplitude of the at least one harmonic is also limited. Overall, the function can be further improved.

A further development proposes that the stator winding has at least two electrically separate winding systems, each of the winding systems being designed according to the number of phases of the electrical machine, the inverter having at least one series connection of the switching elements for each of the phases of the winding systems. In the case of a polyphase electrical machine or an electrical machine with at least two phases, the winding systems of which can be acted upon by means of the series connections of the inverter, in addition to reducing a ripple current amplitude, a more uniform load on the switching elements of the series connections can also be achieved.

This further development can therefore be provided for inverters that operate electrical machines with at least two electrically separated winding systems as stator windings. The invention is not limited to electrical machines with two winding systems. It can of course also be used in particular in electrical machines with at least two phases, in which three or more winding systems that are electrically separated from one another are present. The winding systems can be star-shaped, triangular or n-corner-shaped or the like. For the invention, it can also be noteworthy that individual windings, in particular from different winding systems, can be magnetically coupled to one another through the electrical machine.

To implement the invention, the vector controls or vector controls, which are preferably assigned by means of the winding systems, can be processed again, specifically now taking into account the fact that at least two winding systems are present. The switching signals for a respective one of the series connections can be changed according to the invention, which are assigned to a respective phase of a first of the winding systems. The modulator phase angle can be selected differently for the different winding systems. The modulator phase angle can thus have a positive value for a first one of the winding systems and a corresponding negative value for a second one of the winding systems. This is preferably carried out the same for all phases of the winding systems. In the case of more than two winding systems, it can be provided, for example, that with an odd number of winding systems for one of the winding systems, the switching signals are provided without using the invention. The invention does not therefore need to be used in each of the winding systems. In addition, the switching signals for the winding systems can be based on the same clock signal.

In addition, it can be achieved in particular that stress on the DC voltage intermediate circuit, in particular the intermediate circuit capacitor, can be reduced by, for example, in star-shaped winding systems, zero point voltages of the individual respective winding systems can be set in relation to one another as required. This is of course not restricted to winding systems designed in a star shape.

According to a further development of the invention, it is proposed that individual winding system modulator phase angles are determined for the at least two winding systems depending on the load on the intermediate circuit capacitance and that the respective control angle range is set individually within the setting range depending on the respective winding system modulator phase angles. With a multi-phase electrical Machine or an electrical machine with at least three phases, the winding systems of which can be acted upon by means of the series connections of the inverter, in addition to reducing a ripple current amplitude, an improved, more uniform load on the switching elements of the series connections can also be achieved.

To implement this development, the switching signals, preferably determined by means of respective vector controls or vector controls, can be processed in addition, taking into account the fact that at least two winding systems are present. The switching signals for a respective one of the series connections can be changed according to the invention, which are assigned to a respective phase of a first of the winding systems. The winding system modulator phase angles can preferably be selected to be different for the at least two winding systems. The winding system modulator phase angles are preferably determined individually for each of the at least two winding systems. The winding system modulator phase angles can be determined independently of one another. However, they are preferably determined to be mutually dependent.

By appropriately determining the winding system modulator phase angle, the effect provided by the invention can be further improved. For example, in the case of two winding systems, the winding system modulator phase angles can be shifted opposite to one another in order to be able to achieve at least partial compensation for the stress on the intermediate circuit capacitance through appropriate operation of the winding systems. In addition, the switching signals for further winding systems can be changed according to further individual winding system modulator phase angles.

This is preferably carried out the same for all phases of the winding systems. With more than two winding systems, it can be provided, for example, that the invention does not need to be used in each of the winding systems. In addition, the switching signals for both winding systems can be based on the same clock signal. This is particularly suitable, for example, for reducing the stress on the DC voltage intermediate circuit because, for example, a reliable compensation can be achieved at least partially with regard to the ripple current.

The winding systems can in principle be arranged offset from one another, for example by a predetermined angle of rotation or the like. This can be taken into account accordingly for the determination of the winding system modulator phase angle. Alternatively, of course, they can also be arranged without an offset to one another.

It is further proposed that the individual winding system modulator phase angles are changed by changing between the at least two winding systems. It can thus be provided that the switching signals for the first winding system are changed in accordance with a first winding system modulator phase angle. For the second winding system, the switching signals can be changed in accordance with a second winding system modulator phase angle. Depending on a predefined condition, an inversion can be provided so that the function is swapped when the condition occurs. This allows specific loads, particularly in the area of the switching elements, to be better distributed over a plurality of the switching elements. Overall, a better utilization of the inverter can be achieved. The change can be repeated cyclically.

It is also proposed that the change take place after a predetermined number of clock periods of the clock rate. In this way it can be achieved that loads as well as possibly occurring reactions can be distributed more evenly. For example, it can be provided that a change takes place after a single clock period. However, it can also be provided that the specified number is greater than 1, for example 5, 10, 25 or even more. The specified number of clock periods can also be varied as required and, for example, also be dependent on a phase position of the fundamental oscillation.

Alternatively or in addition, it can also be provided that the change takes place at a predetermined time interval. For this purpose it can be provided that a period of time is determined after which a respective change is to be carried out. For example, it can be provided that after a change has been carried out, the period of time is waited again and a further change then takes place. This can be continued cyclically. This makes it possible to better take into account other specific properties of the inverter and its switching elements as well as the DC voltage intermediate circuit.

It is also proposed that the time interval be specified as a function of thermal properties of the switching elements. This makes it possible to identify specific properties of the Switching elements to be taken into account particularly well and thus to better distribute a load on the switching elements. As a result, the overall reliability of the inverter can be improved.

In addition, it is proposed that the modulator phase angle, in particular the winding system modulator phase angle, be specified as a function of an instantaneous value of the clock period. This makes it possible to use the effect of the modulator phase angle or the winding system modulator phase angle particularly advantageously. Flexibility in terms of effect can be improved.

The advantages and effects specified for the method according to the invention apply equally to the control device according to the invention and to the inverter according to the invention and vice versa. Consequently, process features can also be formulated as device features and vice versa.

The invention also includes developments of the method according to the invention which have features as they have already been described in connection with the developments of the control device according to the invention or the inverter according to the invention. For this reason, the corresponding developments of the method according to the invention are not described again here.

The invention also includes the combinations of the features of the described embodiments.

• 1 eine schematische Prinzipschaltbilddarstellung eines elektrischen Antriebssystems eines elektrisch antreibbaren Kraftfahrzeugs mit einer Asynchronmaschine, die eine Ständerwicklung mit zwei voneinander elektrisch getrennten dreiphasigen Wicklungssystemen aufweist, die an einen entsprechend ausgebildeten Wechselrichter angeschlossen sind, der seinerseits über einen Gleichspannungszwischenkreis mit elektrischer Energie von einer Hochvoltbatterie versorgt wird;
• 2 eine dreidimensionale schematische Diagrammdarstellung für eine Beanspruchung eines Zwischenkreiskondensators des Gleichspannungszwischenkreises abhängig von einem Modulationsgrad beim Betrieb des Wechselrichters gemäß 1 sowie einer Phasenverschiebung zwischen einer Phasenspannung und einem Phasenstrom einer jeweiligen der Phasen;
• 3 eine schematische Diagrammdarstellung für Phasenspannungen gemäß einem SVPWM-Modulationsverfahren für den Betrieb des Wechselrichters gemäß 1;
• 4 eine schematische Diagrammdarstellung für Schaltsignale für die jeweiligen Phasen gemäß dem in 3 dargestellten SVPWM-Modulationsverfahren bei einem Winkel von etwa 90°;
• 5 eine schematische Diagrammdarstellung wie 3 für Phasenspannungen gemäß einem GDPWM-Modulationsverfahren bei einem Modulatorphasenwinkel von -15° für beide Wicklungssysteme gemäß 1;
• 6 eine schematische Diagrammdarstellung für Schaltsignale für die jeweiligen Phasen gemäß dem in 5 dargestellten GDPWM-Modulationsverfahren zu vier in 5 dargestellten Winkeln;
• 7 eine schematische Diagrammdarstellung wie 5, wobei für das erste der Wicklungssysteme ein Modulatorphasenwinkel von -15° und für das zweite der Wicklungssysteme ein Modulatorphasenwinkel von +15° eingestellt ist;
• 8 eine schematische Diagrammdarstellung für Schaltsignale für die jeweiligen Phasen gemäß dem in 7 dargestellten GDPWM-Modulationsverfahren zu vier in 7 dargestellten Winkeln;
• 9 eine schematische Diagrammdarstellung wie 7, wobei für das erste der Wicklungssysteme ein Modulatorphasenwinkel von +15° und für das zweite der Wicklungssysteme ein Modulatorphasenwinkel von -15° eingestellt ist;
• 10 eine schematische Diagrammdarstellung für Schaltsignale für die jeweiligen Phasen gemäß dem in 9 dargestellten GDPWM-Modulationsverfahren zu vier in 9 dargestellten Winkeln;
• 11 eine schematische Darstellung eines Raumzeigerdiagramms für Schaltzustandsvektoren eines jeweiligen der Wechselrichterteile des Wechselrichters gemäß 1;
• 12 ein schematisches Blockdiagramm einer Steuereinrichtung für den Wechselrichter gemäß 1;
• 13 ein schematisches Blockdiagramm für eine Vektorregelung der Steuereinrichtung gemäß 12;
• 14 ein schematisches Blockdiagramm für einen Optimierungsblock für einen Modulationsblock der Steuereinrichtung gemäß 13;
• 15 eine schematische Diagrammdarstellung für eine der Phasenspannungen eines dreiphasigen Systems gemäß 1, wobei mit Graphen drei unterschiedliche DPWM-Modulationsverfahren dargestellt sind; und
• 16 bis 27 schematische Diagrammdarstellungen von vergleichenden Messergebnissen zu unterschiedlichen PWM-Modulationsverfahren.

An exemplary embodiment of the invention is described below. This shows:

• 1 a schematic block diagram representation of an electrical drive system of an electrically drivable motor vehicle with an asynchronous machine, which has a stator winding with two electrically separated three-phase winding systems, which are connected to a correspondingly designed inverter, which in turn is supplied with electrical energy from a high-voltage battery via a DC intermediate circuit;
• 2 a three-dimensional schematic diagram representation of a loading of an intermediate circuit capacitor of the DC voltage intermediate circuit depending on a degree of modulation during operation of the inverter according to FIG 1 and a phase shift between a phase voltage and a phase current of a respective one of the phases;
• 3 a schematic diagram representation for phase voltages according to an SVPWM modulation method for the operation of the inverter according to FIG 1 ;
• 4th a schematic diagram representation for switching signals for the respective phases according to the in 3 SVPWM modulation method shown at an angle of approximately 90 °;
• 5 a schematic diagram representation of how 3 for phase voltages according to a GDPWM modulation method with a modulator phase angle of -15 ° for both winding systems according to 1 ;
• 6th a schematic diagram representation for switching signals for the respective phases according to the in 5 GDPWM modulation method shown to four in 5 illustrated angles;
• 7th a schematic diagram representation of how 5 wherein a modulator phase angle of -15 ° is set for the first of the winding systems and a modulator phase angle of + 15 ° is set for the second of the winding systems;
• 8th a schematic diagram representation for switching signals for the respective phases according to the in 7th GDPWM modulation method shown to four in 7th illustrated angles;
• 9 a schematic diagram representation of how 7th wherein a modulator phase angle of + 15 ° is set for the first of the winding systems and a modulator phase angle of -15 ° is set for the second of the winding systems;
• 10 a schematic diagram representation for switching signals for the respective phases according to the in 9 GDPWM modulation method shown to four in 9 illustrated angles;
• 11 a schematic representation of a space vector diagram for switching state vectors of a respective one of the inverter parts of the inverter according to FIG 1 ;
• 12 a schematic block diagram of a control device for the inverter according to FIG 1 ;
• 13th a schematic block diagram for a vector control of the control device according to 12 ;
• 14th a schematic block diagram for an optimization block for a modulation block of the control device according to FIG 13th ;
• 15th a schematic diagram representation for one of the phase voltages of a three-phase system according to FIG 1 , with Three different DPWM modulation methods are shown in the graph; and
• 16 to 27 schematic diagrams of comparative measurement results for different PWM modulation methods.

The exemplary embodiments explained below are preferred embodiments of the invention. In the exemplary embodiments, the described components of the embodiments each represent individual features of the invention that are to be considered independently of one another, which also develop the invention independently of one another and are therefore also to be regarded as part of the invention individually or in a combination other than the one shown. Furthermore, the described embodiments can also be supplemented by further features of the invention already described.

In the figures, functionally identical elements are each provided with the same reference symbols.

1 shows a drive system in a schematic circuit diagram 10 an electric vehicle (not shown) with an inverter 12 , to which a stator winding of a six-phase asynchronous machine 14th is connected, the stator winding having two winding systems that are electrically separated from one another 42 , 44 having. Any of the winding systems 42 , 44 has three of the phases, namely the first has the winding systems 42 the phases U1 , V1 , W1 and the second of the two winding systems 44 the phases U2 , V2 and W2 on. Here are the winding systems 42 , 44 each executed in star connection. In alternative configurations, the winding systems 42 , 44 however, it can also be designed in a delta connection without impairing the function of the invention.

For each of the phases U1 , U2 , V1 , V2 , W1 , W2 is provided by the inverter 12 a respective series connection of MOSFETs 18th to 40 ready as switching elements, with two of the MOSFETs 18th to 40 are interconnected to form a respective series circuit. The series connections for their part are related to the winding systems 42 , 44 grouped into a first and a second inverter part in the inverter 12 arranged. The first inverter part comprises the switching elements 18th to 28 , whereas the second inverter part is the switching elements 30th to 40 includes.

All of the series connections are connected in parallel to a DC voltage intermediate circuit 46 with an intermediate circuit capacitor 50 connected. The DC link 46 is in turn connected to a high-voltage battery 16 connected, which in the present case provides a DC voltage of approximately 450 V. In alternative configurations, the direct voltage can of course also differ from this.

The MOSFETs 18th to 40 are to a control device 52 connected that for each of the mosfets 18th to 40 provides a respective individual switching signal.

Each of the series connections is connected to a respective low-site MOSFET 20th , 24 , 28 , 32 , 36 , 40 a respective current sensor 48 connected in series. The current sensors 48 are also to the control device 52 connected.

The inverter 12 serves the asynchronous machine 14th with the DC voltage intermediate circuit 46 to couple electrically. For this purpose, in normal operation by means of the series connections, switching signals in accordance with corresponding switching patterns are dependent on the respective phase voltages Uu , Uv , Etc. for the respective phases U1 , U2 , V1 , V2 , W1 , W2 the winding systems 42 , 44 for the respective MOSFET 18th to 40 provided. The phase voltages are alternating voltages. For this purpose, the switching signals are provided in accordance with a pulse width modulation.

To this end, the control device determines 52 the switching signals based on a clock signal using a discontinuous pulse width modulation, which in the present case uses a space vector pulse width modulation (DPWM), a clock period T of the clock signal being less than an oscillation period of a fundamental oscillation of the phase voltages. The control device 52 determines the switching signals for this purpose depending on an operating status indicator 84 ( 11 ) the control device 52 depending on by the MOSFET 18th to 40 achievable switching states of the inverter 12 certain switching state vectors 86 , 88 , 90 , 92 , 94 , 96 , 100 , 102 in a corresponding space vector system 98 determined. First of all, the respective phase voltages are determined by means of a vector control, which in the present case uses a Clarke and Park transformation Uu , Uv , Etc. determined as target values. An operating status indicator is dependent on this 84 ( 11 ) determined.

The control device 52 determines the switching signals depending on the operating status indicator 84 ( 11 ) the control device 52 depending on by the MOSFET 18th to 40 achievable switching states of the inverter 12 certain switching state vectors 86 , 88 , 90 , 92 , 94 , 96 , 100 , 102 in a corresponding space vector system 98 determined.

The control device 52 determines the individual control signals for the MOSFETs 18th to 40 such that the MOSFETs 18th , 20th , 22nd , 24 , 26th , 28 the phase voltages U _U1 , U _V1 , U _W1 for the phases U1 , V1 and W1 provide as well as that the MOSFETs 30th , 32 , 34 , 36 , 38 , 40 the phase voltages U _U2 , U _V2 , U _W2 for the phases U2 , V2 and W2 of the second winding system 44 provide.

11 shows a schematic representation of the space vector diagram 98 for the switching state vectors 86 , 88 , 90 , 92 , 94 , 96 , 100 , 102 of each of the inverter parts of the inverter 12 according to 1 . In the present case, there is therefore a separate space vector diagram for each of the two inverter parts 98 intended. In 11 is also an example of an instantaneous operating status indicator 84 for a respective one of the winding systems 42 , 44 shown as he was by the control device 52 using selected switching state vectors 86 , 88 , 90 , 92 , 94 , 96 , 100 , 102 is determined to the switching signals for the switching elements for each of the winding systems 42 , 44 to create.

The switching state vectors 86 , 88 , 90 , 92 , 94 , 96 are presently active switching state vectors in which an electrical coupling with the DC voltage intermediate circuit 46 will be produced. The active switching state vectors 86 , 88 , 90 , 92 , 94 , 96 are therefore also referred to as active pointers. The switching state vectors 100 , 102 however, are passive switching state vectors in which the phase connections of each of the inverter parts are electrically connected to one another. The passive switching state vectors 100 , 102 are therefore also referred to as zero pointers. The operating status indicator 84 is made by a combination of the switching state vectors 86 , 88 , 90 , 92 , 94 , 96 , 100 , 102 determined. A duration of each of the for determining the operating state pointer 84 switching state vectors used 86 , 88 , 90 , 92 , 94 , 96 , 100 , 102 can be assigned to a respective length of the respective vector.

2 shows a three-dimensional schematic diagram representation for a loading of the intermediate circuit capacitor serving as intermediate circuit capacitance 50 of the DC link 46 depending on a degree of modulation m when operating the inverter 12 according to 1 as well as a phase shift between a phase voltage Uu , Uv , Etc. and an associated phase current of a respective one of the phases U1 , U2 , V1 , V2 , W1 , W2 . The load is shown as the DC link load factor kd with a graph 54 depending on the degree of modulation m and a phase shift for the drive system 10 , in which the switching signals for the respective phase voltages of a respective phase of the first and the second winding system 42 , 44 are essentially provided at the same time. The degree of modulation is defined as follows: $$\text{m}=\frac{\widehat{\text{u}}\text{ph}}{U_{\mathrm{D.}\mathrm{C.}}/2}\frac{\pi }{\mathrm{4th}}$$

In particular, with a degree of modulation m in a range of approximately 0.5 and with a small phase shift, the intermediate circuit load factor kd is large. The DC link load factor kd is here, for example, about 0.63. This also results in a corresponding load on the intermediate circuit capacitor 50 .

wobei Ic dem Strom im Zwischenkreiskondensators 50 und I_ph,sum der Summe aller Phasenströme entspricht.The DC link load factor kd is determined as follows: $$kd=3\ast {\mathrm{I.}}_c/{\mathrm{I.}}_{pH,sum}$$

where Ic is the current in the intermediate circuit capacitor 50 and I _{ph, sum} corresponds to the sum of all phase currents.

This measure can achieve that through the phases U1 , U2 , V1 , V2 , W1 , W2 can at least partially compensate for individual ripple currents, so that overall the stress on the intermediate circuit capacitor 50 can be reduced.

In the present case, the control device 52 a discontinuous PWM (DPWM) is used to carry out the vector control, based on a GDPWM as it is based on the 12 to 15th will be further explained. It is provided in the present case that for a respective clock period T the operating state indicator 84 for one of the period of the fundamental and the number of phases U1 , U2 , V1 , V2 , W1 , W2 dependent number of immediately consecutive clock periods T is determined in such a way that for a time duration of these clock periods T, which correspond to a control angle range v ( 15th ) corresponds to one of the phases in relation to the period of the fundamental U1 , U2 , V1 , V2 , W1 , W2 by means of a respective one of the switching elements 18th , 20th , 22nd , 24 , 26th , 28 , 30th , 32 , 34 , 36 , 38 , 40 with a respective one of the two intermediate circuit potentials of the DC voltage intermediate circuit 46 is electrically coupled. In this exemplary embodiment, the control angle range is v for each of the winding systems 42 , 44 about 60 °. The control angle range v can be within one of the number of phases U1 , U2 , V1 , V2 , W1 , W2 dependent setting range κ can be set. In this exemplary embodiment, the adjustment range κ is approximately 120 ° ( 15th ). Depending on requirements, the setting range κ can of course also be specified smaller. A larger setting range κ is im In contrast, the present exemplary embodiment is essentially not appropriate.

The position of the control angle range v within the setting range κ can now be determined by means of a modulator phase angle γ, which can be approximately -30 ° to approximately + 30 ° in relation to a reference angle α for the present constellation. Basically, the control angle range can of course also be chosen differently, in particular it can be chosen to be smaller.

For this refinement, the reference angle α is positioned approximately in the middle in the adjustment range κ. The reference angle α does not necessarily have to be positioned centrally in the adjustment range κ. In other configurations, it can therefore also be positioned differently from this in relation to the setting range κ, for example at an edge point of the setting range κ or the like. The modulator phase angle γ must then be adjusted accordingly.

Out 15th it can be seen that by varying the modulator phase angle γ on the control of the inverter 12 can be acted upon, whereby a current i _d to the DC voltage intermediate circuit 46 , in particular to a current ic of the intermediate circuit capacitor 50 can be acted upon. The invention makes use of this in order to reduce the stress on the intermediate circuit capacitor 50 to act.

12 shows a schematic block diagram of a section of the control device 52 . A block 110 is assigned to a vector control using GDPWM. At an input N, a speed sensor, not shown, is a speed signal of a speed of a rotor, also not shown, of the asynchronous machine 14th provided. At an input connection M a torque signal is made available, which in the present case corresponds to a momentary torque of the asynchronous machine 14th corresponds. The block 110 determines duty cycle signals U ' _U1,2 , V ' _V1.2 , W ' _W1,2 that a switching signal generator 112 are supplied, from which the respective switching signals for the MOSFET 28 to 40 determined and made available.

13th shows a schematic block diagram for the vector control of the block 110 according to 12 . A signal for a target torque M _Soll and a signal for an actual speed of the rotor of the asynchronous machine 14th become a block 114 supplied, which includes a machine model. The block 114 determines a signal I _{d / q_soll} . A difference maker 116 represents a difference between the signal I _{d / q_soll} and a signal I _{d / q_ist} for a control block 118 ready. The rule block 118 determines a signal U _{d / q} , which is a transformation block 120 is fed. The transformation block 120 carries out an inverse Clarke transformation and an inverse Park transformation and determines the phase voltages from the signal U _{d / q} Uu , Uv , Etc. . The phase voltages Uu , Uv , Etc. then become a block 122 which, among other things, executes a GDPWM and the duty cycle signals for the switching signal generator 112 determined.

In the embodiment described above, it has so far been assumed that the winding systems 42 , 44 are not offset from one another in the stand. In specific embodiments, however, this may be different.

In the following exemplary embodiment, which is based on the exemplary embodiment described above, there is an offset for the arrangement of the two winding systems 42 , 44 of about 30 °. With regard to a reference angle, for example, for the first winding system 42 an angle ϑ ₁ , and for the second winding system 44 an angle ϑ _{2 can be} provided. The two winding systems 42 , 44 are thus arranged offset from one another in the stator by a difference angle ϑ ₁ - ϑ ₂ . This can supplement the determination of the actual switching signals for the MOSFET 28 to 40 be used. 14th FIG. 11 shows a more detailed schematic block diagram based on FIG 13th , the spatial arrangement of the winding system 42 , 44 can also be taken into account in the stand. The offset can of course also be selected differently in alternative configurations.

Out 14th it can be seen that for each of the two winding systems 42 , 44 a respective block 120 is provided, the respective block 120 a respective angle signal corresponding to the respective angle ϑ ₁ , ϑ _{2 of} the respective winding system 42 , 44 receives. The blocks 120 then set the respective phase voltages U _U1 , U _V1 , U _W1 for the first winding system 42 such as U _U2 , U _V2 , U _W2 for the second winding system 44 ready. This can be supplemented accordingly for other winding systems, as in 14th is shown in dashed lines.

The indices of the duty cycle signals U ', V ' , W ' relate to the respective winding system 42 , 44 . The index 1 refers to the first winding system 42 whereas the index 2 on the second winding system 44 relates. These duty cycle signals U ' , V ' , W ' will then be in the block 122 - how to 15th explained - treated, whereby the corresponding duty cycle signals U ' _U1 , U ' _V1 ' , U ' _W1 and U ' _U2 , U ' _V2 , U ' _W2 be determined. These are then the block 112 according to 12 fed.

Out 14th it can also be seen that a block 124 is provided, which is composed of signals for the angles ϑ ₁ , ϑ ₂ , the phase currents strö _ph and the currents I _{C, soll} and I _{c, is} modulation phase angles γ ₁ , γ ₂ for the winding systems 42 , 44 determined. For each of the winding systems 42 , 44 is a respective individual block 122 provided, to which the respective individual modulation phase angle γ ₁ , γ _{2 is} fed. The blocks 122 then determine the duty cycle signals U ' _U1 , U ' _V1 ' , U ' _W1 , U ' _U2 , U ' _V2 , U ' _W2 depending on the respective modulation phase angles γ ₁ , γ ₂ and phase voltages.

The effects of this control or process management are based on the following 3 to 10 further explained. Based on 16 to 27 measurement results are then discussed.

3 shows a schematic diagram representation of a voltage-angle diagram in which an abscissa is assigned to the angle in ° and an ordinate is assigned to a voltage normalized with respect to half an intermediate circuit voltage. Using two graphs 68 , 70 are midpoint voltages of a selected one of the phases of the two winding systems 42 , 44 the asynchronous machine 14th according to 1 shown, the half of the DC link voltage on the DC link 46 are normalized. The midpoint voltages are between a respective phase connection of the inverter 12 recorded in relation to a mean potential of the DC link voltage. The mean potential is approximately in the middle with respect to the two electrical potentials of the DC link voltage.

3 shows a corresponding voltage curve with a graph 68 for the first winding system 42 and with a graph 70 for the second winding system 44 . In the type of modulation selected here, space-vector pulse-wide modulation (SVPWM) modulation, i.e. continuous PWM, is provided. In 3 is an angle β ₁ , shown, to the switching signals for the switching elements of the inverter 12 in 4th are shown.

4th shows in a schematic signal representation a clock period T of the switching signals for the angle β ₁ according to 3 . Due to the geometric offset of the winding systems 42 , 44 can be a slight relief on the DC link 46 can be achieved, but the impact is limited, as shown in the in 4th shown switching signals can be seen. A significant relief on the DC link 46 , especially on the intermediate circuit capacitor 50 is therefore not to be expected with this control method.

5 now shows in a schematic diagram how 3 by means of two graphs 72 , 74 the conditions when using a GDPWM. In the diagram in 5 angles β ₂ to β _{5 are} highlighted and can be assigned to the corresponding possible operating situations. 6th shows in a schematic signal representation how 4th the switching signals corresponding to the selected angles β ₂ to β ₅ .

The left clock period T in 6th is assigned to the angle β ₂ , the subsequent clock period T to the angle β ₃ , the subsequent clock period T to the angle β ₄ , and the last clock period T to the angle β ₅ . Due to the geometric offset of the two winding systems 42 , 44 of about 30 ° occur according to 6th cases of switching signals shown.

In the cases β ₂ and β _{4 there} are active pointer and zero pointers of the two winding systems 42 , 44 opposite, which significantly reduces the stress on the DC link 46 and here also the intermediate circuit capacitor 50 can be reached. Nevertheless, it should be noted that in these cases the formation of harmonics can be increased, especially when the winding systems 42 , 44 are magnetically coupled. The modulation phase angles γ ₁ and γ ₂ are here approximately -15 °. With the graph 72 , 74 the corresponding midpoint voltages are shown again.

For the cases β ₃ and β ₅ , there are opposite effects, because namely active space vectors and zero pointers of the two winding systems 42 , 44 occur essentially simultaneously. This is similar to the aforementioned SVPWM.

By mixing the two aforementioned case constellations, their effects can be combined; for example, by changing at a rhythm of about 30 °, the load on the DC voltage intermediate circuit can be moderately reduced 46 , especially the intermediate circuit capacitor 50 can be achieved.

7th now shows in a further schematic representation how 5 the corresponding ratios if for the first winding system 42 a modulator phase angle γ ₁ of about -15 ° and for the second winding system 44 a modulator phase angle γ ₂ of approximately + 15 ° is selected. In the diagram in 7th two angles β ₆ and β ₇ are shown highlighted as an example. Using graphs 76 , 78 are the respective midpoint voltages for a respective one of the phases of the two winding systems 42 , 44 shown.

8th shows in a schematic signal representation how 6th the corresponding switching signals for the two in 7th highlighted angles β ₆ and β ₇ . The left representation of the switching signals of the clock period T relates to the angle β _6, whereas the right representation of the switching signals of the clock period T relates to the angle β ₇ . In this modulation method, the two modulation phase angles γ ₁ and γ ₂ are varied so that mainly the cases β ₂ and β ₄ according to 6th occur. This allows the DC link 46 , especially the intermediate circuit capacitor 50 , moderately to heavily relieved.

In order to also be able to achieve an optimization with regard to harmonics, a control method according to the schematic diagram in FIG 9 intended. 9 shows in a schematic diagram representation how 7th using graphs 80 , 82 the corresponding midpoint voltages, where for the first winding system 42 a modulator phase angle γ ₁ of about + 15 ° and for the second winding system 44 a modulator phase angle γ ₂ of approximately -15 ° is provided. The graph 80 shows the midpoint voltage for the first winding system 42 whereas the graph 82 the midpoint voltage for the second winding system 44 represents. 10 shows in a schematic signal representation how 8th the corresponding switching signals for two highlighted angles β ₈ and β ₉ . The left clock period T is assigned to the angle β ₈ and the right clock period T is assigned to β ₉ .

It can be seen that in this embodiment, predominantly pulse patterns according to the angles β ₃ and β ₅ according to FIG 5 and 6th occur. This configuration makes it possible to have a moderate or even low load in relation to the stress on the DC voltage intermediate circuit 46 or the intermediate circuit capacitor 50 to achieve, but a better overall efficiency can be achieved, since harmonics or harmonics can be reduced with respect to the fundamental.

Of course, the aforementioned variants can also be varied with one another, it being possible to switch between different variants as required. In particular, the modulation phase angles γ ₁ and γ ₂ can be set differently and as required, depending on whether the overall system relies more heavily on relief at the DC voltage intermediate circuit 46 or to be optimized for an improvement in the overall efficiency.

In particular, it should be noted that for the winding systems 42 , 44 so no shift of the respective clock signal needs to be provided. This is a great advantage if, for example, the clock frequency is to be changed.

With the 16 to 27 are metrologically recorded effects of the 3 to 10 discussed control method shown and further explained. The 16 to 27 represent for this purpose respective schematic diagram representations of torque-speed diagrams in which an abscissa is assigned to a speed in 1 / min and a respective ordinate is assigned to a respective torque in the unit Nm. The parameter shown in the diagrams relates to the DC link load factor kd, as described above.

The diagram according to 16 refers to an SVPWM with a clock rate of around 10 kHz. This can be seen from 16 that with a small amount of torque and a speed of more than about 2000 rpm, the load on the intermediate circuit capacitor 50 is comparatively large. It is well above 0.6 here. 17th shows the corresponding relationships for a clock rate of 8 kHz. It can be seen that the load is corresponding to the situation 16 even worse.

18th now shows a schematic diagram of how 16 , but for DPWM1 modulation at a clock rate of 10 kHz. In the case of high torques in terms of magnitude, the intermediate circuit capacitor can be relieved 50 can be achieved, but the load on the intermediate circuit capacitor remains 50 in the case of torques that are small in terms of magnitude and speeds of greater than about 2000 rpm, comparably high, and in fact similar to the situation according to 16 . In this area, therefore, no relief can be achieved with the DPWM1 modulation.

19th shows a schematic diagram of how 18th , but for GDPWM modulation at a clock rate of 10 kHz. It can be seen that the stress essentially corresponds to what it is based on 18th for DPWM1 modulation has already been explained, which is why reference is made to the relevant statements.

A significant reduction in the stress on the intermediate circuit capacitor 50 results, however, for a GDPWM modulation using operating conditions as they are based on 7th and 8th are explained. This is based on the schematic diagram according to FIG 20th shown which how 19th is scaled. It can be seen that the load on the intermediate circuit capacitor 50 in the diagram according to 20th is not only smaller over a large area, but also that the stress can be further reduced with small torques in terms of amount. A considerable reduction in the stress on the intermediate circuit capacitor can therefore be achieved here 50 can be achieved.

21st represents in a schematic diagram like 20th represents the corresponding ratios of the modulation according to the parameters as they are based on the 9 and 10 are described. It can be seen that this is the load on the intermediate circuit capacitor 50 compared to the operating conditions according to 20th is increased. At the same time, however, it should be remembered that the harmonics are significantly reduced with this modulation.

With the 22nd to 27 are now the conditions of the stress on the intermediate circuit capacitor 50 shown, relative to the operating situation according to 19th , namely the GDPWM at a clock rate of 10 kHz. The parameter in the 22nd to 27 thus relates to the relative change in relation to the representation according to 19th . 25th serves as a reference in the present discussion for the other cases, which is why in 25th a uniform graph is shown.

22nd is with regard to the operating situation of 16 assigned. It can be seen that the load on the intermediate circuit capacitor 50 according to the operating situation as determined by 16 was explained, compared to the operational situation as it is based on 19th has been explained is significantly deteriorated. The stress on the intermediate circuit capacitor 50 is considerably increased, especially at low torques and high speeds. Basically the same conditions also show 23 related to the operating situation of the 17th corresponds. Here, too, the stress on the intermediate circuit capacitor is significantly worse 50 to be recorded.

24 according to the operating situation 18th corresponds to, shows slight improvements in relation to the stress on the intermediate circuit capacitor 50 .

However, a particularly significant improvement can be made according to the operating situation 20th how this can be achieved using 26th can be seen. Even with small torques and high speeds, the DC link capacitor can still be relieved of load 50 can be achieved.

On the other hand, there is a deterioration compared to the reference according to FIG 25th for the operating situation according to 21st how to do this using 27 is shown.

If, in the present embodiment, an optimization is carried out with regard to harmonics, this can result in less favorable stress or greater stress on the intermediate circuit capacitor 50 have as a consequence. If, however, the load on the intermediate circuit capacitor 50 alone optimized how this is based on 26th is shown, a considerable reduction in the stress on the intermediate circuit capacitor 50 can be achieved.

By combining the operating situations according to 20th and 21st and 26 and 27, respectively, an optimization can thus be achieved overall, so that on the one hand the stress on the intermediate circuit capacitor 50 can be reduced, on the other hand and at the same time an optimization with regard to harmonics can be achieved. For this purpose, for example, a change between different operating modes, as for example based on the 7th to 10 have been explained.

Overall, the invention allows that only a single carrier signal needs to be used, whereby all currents of the series connections can be detected by means of current sensors that are arranged on low-site MOSFETs, for example on their emitters or the like. In this way, an inexpensive implementation can be achieved. In addition, only a single timer unit is required for the control device 52 , in particular their computer unit to be provided. Furthermore, by simultaneous sampling and conversion of the recorded currents, a stability of the regulation by the control device can be achieved 52 can be improved because both machine-related and coupling-related interference, for example due to harmonics, can be better tolerated. Finally, the invention can also be implemented simply by means of a computer program, so that it can also be implemented in a simple manner with existing inverters or control devices or even retrofitted.

In addition, great flexibility with regard to changes in the clock rate can also be implemented in order to optimize or improve efficiency or NVH, for example. Finally, the invention also makes it possible to achieve a high voltage utilization, in particular greater than with continuous modulation methods such as SVPWM or the like, as a result of which high power and high efficiency can also be achieved in a field weakening range.

Of course, the invention is not limited to use in systems with two winding systems. If necessary, the number of winding systems can also be selected differently from this.

Overall, the example shows how the invention can improve the stress on the DC voltage intermediate circuit, in particular the intermediate circuit capacitor, by manipulating the modulator phase angle γ.

The exemplary embodiments presented above serve exclusively to explain the invention and are not intended to restrict it.

QUOTES INCLUDED IN THE DESCRIPTION

This list of the documents listed by the applicant was generated automatically and is included solely for the better information of the reader. The list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions.

Patent literature cited

• US 2011/0221365 A1 [0008]
• JP 2001197779 A [0011, 0012, 0013]

Non-patent literature cited

• "A High-Performance Generalized Discontinuous PWM Algorithm" by Ahmet M. Hava in IEEE Transactions on Industry Applications, Vol. 34, No. 5, September / October 1998 [0033]
• "Analytical calculation of the RMS current stress on the DC-link capacitor of voltage-PWM converter systems" by J. W. Kolar and S. D. Round in IEE Proc.-Electr. Power Appl., Vol. 153, No. 4, July 2006 [0058]

Claims (14)

Method for operating switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) of an inverter (12) to which a polyphase stator winding of an electrical machine (14) is connected, wherein the inverter (12) for each of the phases (U1, U2, V1, V2, W1, W2) of the stator winding (42, 44) has at least one series connection of the switching elements (18, 20, 22, 24, 26, 28, 30 , 32, 34, 36, 38, 40) in order to electrically couple the stator winding to a DC voltage intermediate circuit (46) connected to the inverter (12) by connecting the switching elements of the series circuits with the respective phase voltages of the respective phases (U1, U2, V1 , V2, W1, W2) dependent switching signals for the respective switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) are applied, the switching signals based on a clock signal using a discontinuous pulse width modulation can be determined, with a clock period of the clock signal small r is an oscillation period of a fundamental oscillation of the phase voltages, the switching signals being determined as a function of at least one operating state indicator (84), which in turn depends on the switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) achievable switching states of the inverter (12) certain switching state vectors (86, 88, 90, 92, 94, 96, 100, 102) is determined in a corresponding space vector system (98), the operating state vector for one of the period the fundamental frequency and the number of phases (U1, U2, V1, V2, W1, W2) dependent number of immediately consecutive clock periods (T) is determined in such a way that for a control angle range (v) in relation to the period duration of the fundamental frequency of these clock periods (T) at least one of the phases (U1, U2, V1, V2, W1, W2) by means of at least one respective one of the switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38 , 40) with one of at least two intermediate circuit potentials of the DC voltage intermediate circuit (46) is electrically coupled, characterized in that the control angle range (v) is set as a function of a modulator phase angle (γ) within an adjustment range (κ), for which purpose a loading of an intermediate circuit capacitance of the DC voltage intermediate circuit (46) ) is determined and the modulator phase angle (γ) is determined depending on the determined stress. Procedure according to Claim 1 , characterized in that the switching signals for all switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) are based on the same clock signal. Method according to one of the preceding claims, characterized in that the modulator phase angle (γ) is determined as a function of a degree of modulation. Method according to one of the preceding claims, characterized in that the modulator phase angle (γ) is determined as a function of the fundamental oscillation. Method according to one of the preceding claims, characterized in that the modulator phase angle (γ) is determined as a function of a ripple current amplitude of an intermediate circuit capacitor (50) of the DC voltage intermediate circuit (46). Method according to one of the preceding claims, characterized in that the modulator phase angle (γ) is determined as a function of an amplitude of at least one harmonic with respect to the fundamental oscillation. Method according to one of the preceding claims, characterized in that the stator winding has at least two electrically separate winding systems (42, 44), each of the winding systems (42, 44) corresponding to the number of phases (U1, U2, V1, V2, W1, W2) of the electrical machine (12), the inverter (12) for each of the phases (U1, U2, V1, V2, W1, W2) of the winding systems (42, 44) having at least one series connection of the switching elements ( 18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40). Procedure according to Claim 7 , characterized in that individual winding system modulator phase angles (γ ₁ , γ ₂ ) are determined for the at least two winding systems (42, 44) depending on the load on the intermediate circuit capacitance and the respective control angle range (v) is determined depending on the respective winding system modulator phase angles (γ ₁ , γ ₂ ) is set individually within the setting range (κ). Procedure according to Claim 8 , characterized in that the individual winding system modulator phase angles (γ ₁ , γ ₂ ) are changed by changing between the at least two winding systems (42, 44). Procedure according to Claim 9 , characterized in that the change takes place after a predetermined number of clock periods (T) of the clock signal. Procedure according to Claim 9 or 10 , characterized in that the change takes place at a predetermined time interval. Procedure according to Claim 11 , characterized in that the time interval is specified as a function of the thermal properties of the switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40). Control device (52) for operating switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) of an inverter (12) to which a polyphase stator winding of an electrical machine (14) is connected, the inverter (12) for each of the phases (U1, U2, V1, V2, W1, W2) of the stator winding (42, 44) at least one series circuit of the switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) to electrically couple the stator winding to a DC voltage intermediate circuit (46) connected to the inverter (12), the control device being designed: the switching elements of the series connections with respective phase voltages to apply switching signals dependent on the respective phases (U1, U2, V1, V2, W1, W2) for the respective switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40), - to determine the switching signals based on a clock signal using a discontinuous pulse width modulation, where if a clock period of the clock signal is less than an oscillation period of a fundamental oscillation of the phase voltages, - to determine the switching signals as a function of at least one operating state indicator (84), which the control device (52) in turn depends on by the switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) achievable switching states of the inverter (12) determined switching state vectors (86, 88, 90, 92, 94, 96, 100, 102) in a corresponding space vector system (98) by determining the operating state pointer for a number of immediately consecutive clock periods (T) dependent on the period duration of the fundamental oscillation and the number of phases (U1, U2, V1, V2, W1, W2) in such a way as to provide a control angle range (v) in relation to the period of these clock periods (T) corresponding to the period of the fundamental oscillation, at least one of the phases (U1, U2, V1, V2, W1, W2) by means of at least one respective one of the switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) to be electrically coupled to one of at least two intermediate circuit potentials of the DC voltage intermediate circuit (46), characterized in that the control device (52) is designed is to - set the control angle range (v) as a function of a modulator phase angle (γ) within an adjustment range (κ), - to determine a load on an intermediate circuit capacitance of the DC voltage intermediate circuit (46), and - to determine the modulator phase angle (γ) as a function of the determined load. Inverter (12) for connection to a polyphase stator winding of an electrical machine (14), the stator winding being designed according to the number of phases (U1, U2, V1, V2, W1, W2) of the electrical machine (14), with - at least one series connection of the switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40) for each of the phases (U1, U2, V1, V2, W1, W2) of the stator winding in order to electrically couple the electrical machine (14) to a DC voltage intermediate circuit (46) that can be connected to the inverter (12), by using the series connections for the respective phase voltages for the respective phases (U1, U2, V1, V2, W1, W2) the stator winding are provided depending on switching signals for the respective switching elements (18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40), and - a control device (52) for providing the switching signals, thereby characterized in that the control device (52) according to Claim 13 is trained.

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