Summary of the invention
The present invention is directed to the deficiency that above-mentioned existing current mode resonance DC converter exists, its purpose is to provide a kind of novel current-type multi-resonance direct current (DC) converter.This converter can provide higher voltage gain, all has less circulation in full-load range, and switching tube and rectifier diode all are operated in the ZCS state, and also has higher efficient during underloading work.
In order to achieve the above object, the present invention is achieved by the following technical solutions.
A kind of current-type multi-resonance direct current (DC) converter, the square wave current source generator, multi resonant vibrating network, the rectifying and wave-filtering output unit that comprise series connection successively, it is characterized in that, described multi resonant vibrating network comprises transformer, parallel resonant inductor, parallel resonance electric capacity and series resonance inductance, and described series resonance inductance is connected the former limit of transformer; Described rectifying and wave-filtering output unit comprises diode rectifier circuit and is connected in parallel on the filter capacitor of diode rectifier circuit output.
Characteristics and effect of the present invention are described as follows:
(1) described square wave current source generator adopts current mode half-bridge structure, current mode full bridge structure or current mode push-pull configuration to constitute.The square wave current source generator can produce square wave current, has less input current ripple, and ripple frequency is the twice of switching frequency, can be by this ripple of polarity free capacitor filtering of low-cost small size.
(2) multi resonant vibrating network comprises transformer, parallel resonant inductor, parallel resonance electric capacity and series resonance inductance, and the series resonance inductance is connected the former limit of transformer.When the conducting simultaneously of the switching tube of square wave current source generator, series resonance inductance and parallel resonance electric capacity generation resonance, make the electric current that flows into multi resonant vibrating network that soft commutation take place, realized the ZCS of switching tube, reduced the switch tube voltage spike that the series resonance inductance causes simultaneously.The resonance that multi resonant vibrating ruton is crossed shunt inductance and shunt capacitance provides high voltage gain for converter.In the multi resonant vibrating network, the high voltage gain that shunt inductance and shunt capacitance provide by resonance, can further reduce parasitic parameters such as transformer voltage ratio and associated leakage inductance and turn-to-turn capacitance, at this moment, the diode junction capacitance of the turn-to-turn capacitance of transformer and rectifying and wave-filtering output unit is used as parallel resonance electric capacity, can be by good restraining by the current spike that they cause.
(3) switching tube of described square wave current source generator open duty ratio greater than 0.5, form one section half duty ratio overlap the time (switching tube open duty ratio, less than 0.7) greater than 0.5 less than the harmonic period of series resonance inductance and parallel resonance electric capacity.In the time that duty ratio overlaps, the input of multi resonant vibrating network is by short circuit, series resonance inductance and parallel resonance electric capacity are realized the soft commutation of multi resonant vibrating network input current by resonance, multi resonant vibrating network input current after the commutation flows through its inverse parallel diode after switching tube turn-offs, the switching tube of square wave current source generator can be realized zero-current switching, has reduced the switch tube voltage spike that the series resonance inductance causes simultaneously.
(4) the rectifying and wave-filtering output unit comprises diode rectifier circuit and the filter capacitor that is connected in parallel on the diode rectifier circuit output.Described diode rectifier circuit adopts full bridge rectifier, full-wave rectifying circuit or voltage doubling rectifing circuit.Adopt voltage doubling rectifing circuit can further improve voltage gain, thereby reduce the no-load voltage ratio and the parasitic parameters such as associated leakage inductance and turn-to-turn capacitance of transformer, raise the efficiency.
Because the rectifying and wave-filtering output unit adopts the rectification circuit that has filter capacitor, thereby the maximum voltage of parallel resonance electric capacity is can filtered electric capacity clamped to output voltage, has reduced the on-state loss that circulation energy and circulation brought in parallel resonant inductor and the parallel resonance electric capacity thus.The electric current of rectifying and wave-filtering output unit rectifier diode of flowing through is multi resonant vibrating network input current and parallel resonant inductor difference between currents, in rectifier diode ON time section, multi resonant vibrating network input current is approximately constant, the parallel resonant inductor electric current is linear to be increased up to equating with multi resonant vibrating network input current, rectifier diode can be realized zero-current switching thus, has solved the rectifier diode reverse-recovery problems.
During underloading, because the rectifier diode ON time shortens, parallel resonance electric capacity was also reduced relatively by the time of clamper, the time of parallel resonant inductor and parallel resonance capacitor resonance can be elongated, the initial energy storage of parallel resonance electric capacity and series resonance inductance resonance is reduced, thereby can limit the overshoot of converter multi resonant vibrating network input current when underloading is worked.
The present invention's " current-type multi-resonance direct current (DC) converter " is the efficient that has improved boost power converter on the prior art basis (the actual measurement converter can obtain about 95% peak efficiencies under 15 times of conditions of boosting), greatly reduce the input current ripple, thereby can more efficient use comprise many new forms of energy of solar energy and fuel cell with low-voltage and high-current direct current output characteristic.
Embodiment
(1) further specifies circuit topological structure of the present invention.
With reference to Fig. 1, for the typical current type multi resonant of the present invention DC converter of shaking, be three grades of series systems, wherein the first order is the square wave current source generator, and the second level is multi resonant vibrating network, and the third level is the rectifying and wave-filtering output unit.
(1) square wave current source generator
Low-voltage dc power supply V
InBe connected between the positive-negative input end of square wave current source generator, between the positive-negative input end of square wave current source generator, be parallel with the low-voltage filter capacitor C respectively
In, first switching branches, second switch branch road.First switching branches is by first inductance L
In1Series connection forms with first switching tube, wherein diode D
Q1, capacitor C
Q1Be switching tube Q
1Self inverse parallel diode and output capacitance.Second switch props up route second inductance L
In2With second switching tube Q
2Series connection forms, wherein diode D
Q2, capacitor C
Q2Be switching tube Q
2Self inverse parallel diode and output capacitance.First inductance L
In1With the first switching tube Q
1The series connection contact, second inductance L
In2With second switch pipe Q
2The series connection contact, draw positive-negative output end respectively as the square wave current source generator.Form square wave current by the break-make of controlling first switching tube and second switch pipe.
The square wave current source generator can adopt current mode half-bridge structure, current mode full bridge structure or current mode push-pull configuration, the square wave current source generator can produce square wave current, has less input current ripple, and ripple frequency is the twice of switching frequency, can be by this ripple of polarity free capacitor filtering of low-cost small size.
(2) multi resonant vibrating network
Multi resonant vibrating network comprises transformer T, parallel resonant inductor L
p, the parallel resonance capacitor C
pWith the series resonance inductance L
r, the series resonance inductance L
rBe connected the former limit of transformer, parallel resonant inductor L
p, the parallel resonance capacitor C
pBe connected the transformer secondary, shown in Fig. 2 (a).In conjunction with Fig. 2 (b) parallel resonant inductor L
p, the parallel resonance capacitor C
pAlso can be connected the former limit of transformer; In conjunction with Fig. 2 (c), 2 (d) parallel resonant inductor L
p, the parallel resonance capacitor C
pAlso can be connected to former limit of transformer and secondary.
As shown in Figure 3, transformer T also can be that secondary has centre tapped structure, parallel resonant inductor L
p, the parallel resonance capacitor C
pCan all be connected the former limit of transformer with reference to Fig. 3 (a), also can shown in Fig. 3 (b), all be connected the transformer secondary, perhaps be connected to former limit of transformer and secondary with reference to Fig. 3 (c), 3 (d).
Transformer T also can adopt former sideband as shown in Figure 4 that centre tapped structure is arranged, at this moment the series resonance inductance L
rBe connected with the former limit of transformer centre cap.Parallel resonant inductor L
p, the parallel resonance capacitor C
pCan shown in Fig. 4 (a), all be connected the transformer secondary.Shown in Fig. 4 (b), parallel resonant inductor L
P1And L
P2, the parallel resonance capacitor C
pAlso can all be connected the former limit of transformer.Parallel resonant inductor L
p, the parallel resonance capacitor C
pCan also be connected to former limit of transformer and secondary with reference to Fig. 4 (c), 4 (d).
The series resonance inductance L
rCan be external independent inductance, also can be the leakage inductance of transformer T; Parallel resonant inductor L
pCan be external independent inductance, also can be the magnetizing inductance of transformer; The parallel resonance capacitor C
pCan be the turn-to-turn capacitance of transformer, also can be external independent capacitance.As parallel resonant inductor L
pWith the parallel resonance capacitor C
pThe two one of or when all being placed on the former limit of transformer, the two ends of parallel resonance element are directly in parallel with the former limit of transformer two-port, the series resonance inductance L
rPort of an end and the former limit of transformer link to each other, the other end links to each other with a port of square wave current source generator.
(3) rectifying and wave-filtering output unit
The rectifying and wave-filtering output unit comprises diode rectifier circuit and is connected in parallel on the filter capacitor of diode rectifier circuit output.The diode full-wave rectifying circuit can adopt the full bridge rectifier shown in Fig. 5 (a), can adopt with reference to the full-wave rectifying circuit shown in Fig. 5 (b).Shown in Fig. 5 (c), the diode full-wave rectifying circuit also can adopt voltage doubling rectifing circuit further to improve voltage gain, thereby reduces the no-load voltage ratio and the parasitic parameter such as associated leakage inductance and turn-to-turn capacitance of transformer, raises the efficiency.D among Fig. 5 (a)
1-D
4Be rectifier diode, C
0Be filter capacitor, R
LIt is the load resistance of converter.D among Fig. 5 (b)
1-D
2Be rectifier diode, C
0Be filter capacitor, R
LIt is the load resistance of converter.D among Fig. 5 (c)
1-D
2Be rectifier diode, C
1, C
2Be the multiplication of voltage filter capacitor, R
LIt is the load resistance of converter.
(2) further specify the shake Parameters design of DC converter of typical current type multi resonant shown in Figure 1.
(a) design of resonant network:
When derivation transducer gain G because the switching tube duty ratio of square wave current source generator overlaps the time less, suppose that the electric current of inflow series resonant network is a square wave current, the simplified electrical circuit diagram of converter and oscillogram as shown in Figure 6 and Figure 7, v wherein
G1And v
G2Be the drive signal of first, second switching tube, V
oBe the output voltage of converter, v
LpBe to flow into L
pElectric current, i
CSBe square wave current, v
RecBe the output voltage of multi resonant vibrating network, i
RecBe the electric current that flows into the rectifying and wave-filtering output unit, v
Rec_1, i
Rec_1And i
CS_1Be respectively v
Rec, i
RecAnd i
CSThrough the fundametal compoment after the Fourier expansion, θ is the conducting phase angle of rectifier diode, and φ, Ψ and α are respectively v
Rec_1, i
Rec_1And i
CS_1Phase angle, δ is v
Rec_1And i
Rec_1Phase difference, λ is v
Rec_1And i
CS_1Phase difference.
Because v
Rec_1Lag behind i
Rec_1, angle of retard is δ, the output of multi resonant vibrating network is capacitive load, thus can draw the simplified model of ac equivalent circuit, as shown in Figure 8, i wherein
CS_1Be square wave current i
CSFirst-harmonic after the Fourier expansion, R
eAnd C
eBe the output capacitive loading of multi resonant vibrating network equivalence, Z
InBe the input impedance of ac equivalent circuit, V
Rec1_pkThe output voltage that is multi resonant vibrating network is through the first-harmonic peak value after the Fourier expansion, I
CS1_pkBe to flow into the square wave current of series resonant network through the first-harmonic peak value after the Fourier expansion.R
e, C
e, Z
InAnd I
CS1_pkExpression formula as (2), (3), (4) are shown in (5).
Wherein, F is that the switching frequency angular frequency is with respect to series resonant network resonance frequency angular frequency
pNormalized value, Z
pBe the impedance of series resonant network, Q
pBe the quality factor of converter output loading, Q
eBe the loaded quality factor of ac equivalent circuit, expression formula is respectively by formula (6), and (7), (8) are shown in (9).Formula (10) is the expression formula of the efficiency eta of converter.Wherein, N is the no-load voltage ratio of transformer, V
In, I
InBe the input voltage and the input current of converter, V
oBe the output voltage of converter, R
LIt is the load resistance of converter.
tan(|δ|)=ωC
eR
e (3)
By (5), (1), (4), (9) and (10) can draw the gain G of converter, as the formula (11).The input phase angle λ of the resonant network of ac equivalent circuit as the formula (12).
When the design resonant network, at first, according to formula (1), (2), (3) determine the scope of operating frequency, resonance angular frequency ω
pAnd the quality factor q of load
L,, promptly realize multi resonant vibrating network input current i so that when satisfying the pressure regulation requirement, realize the ZCS of square wave current source generator switching tube
PriSoft commutation.Realize multi resonant vibrating network input current i
PriThe condition of soft commutation be i
PriBe ahead of the voltage v at multi resonant vibrating network two ends
Rec, promptly λ is less than zero, and
Moment C
pThe energy of storage
Be greater than resonant inductance L
rThe required energy of electric current commutation, as the formula (14).
λ<0 (13)
Secondly, calculate minimum voltage gain G
Min, select transformer voltage ratio N.According to fixing ω
pAnd Q
L, through type (7) calculates L
pAnd C
PValue.Calculate L according to formula (14) then
rValue.Utilize resonant network to absorb the parasitic parameter of transformer.
(b) calculating of duty ratio:
According to L
r, C
PAnd the value of N, by Fig. 9 and formula (15), (16), (17), (18), (19), (20), (21), (22), (23) can obtain the restrictive condition of duty ratio.Wherein, D
MinAnd D
MaxBe maximum duty cycle and minimum duty cycle, Δ T
1,2With Δ T
1,4Be respectively t among Figure 11
1To t
2Time period and t
1To t
4Time period, C '
pBe C
pEquivalence is to the capacitance on the former limit of transformer, ω
rBe L
rWith C '
pThe series resonance angular frequency, Z
rBe L
rWith C '
pThe impedance of the series resonance network of forming.
ΔT
1,2=t
2-t
1 (17)
ΔT
1,4=t
4-t
1?(18)
C′
p=N
2C
p (19)
(c) calculating of input inductance:
When switching tube conducting simultaneously, input current i
InLinear increase, i in all the other times
InLinearity reduces.Therefore the frequency of input current ripple is the twice of switching frequency, and the amplitude of input current ripple can be calculated by formula (24), wherein, and Δ i
InBe the peak-to-peak value of input current ripple, Δ T
OvIt is the time of former limit switching tube conducting simultaneously.
Input current ripple value according to selected can calculate input inductance L by formula (24)
In1And L
In2Value L.
(3), the shake operation principle of DC converter of typical current type multi resonant shown in Figure 1 is described further combined with Figure 10, Figure 11.
Wherein, Figure 10 (a) to (f) is six working mode figures of current-type multi-resonance direct current (DC) converter in preceding half period, six working mode figures and the preceding half period symmetry of this converter in the half period of back, and Figure 11 is corresponding circuit working oscillogram.
At t
0Constantly, Q
1Conducting, Q
2Turn-off.Input power supply V
InGive input inductance L
In1Charging, input inductance L
In2Energy stored is by rectifier diode D
2And D
3Pass to load.The parallel resonance capacitor C
pVoltage v
CpBe clamped at negative output voltage V
o, parallel resonant inductor L
pLinear increase of current reversal.Flow through rectifier diode D
2And D
3Electric current equal transformer secondary current i
SecDeduct the L that flows through
pCurrent i
Lp
Pattern 1 (t
0-t
1): at t
0Constantly, i
LpEqual i
Sec, flow through D
2And D
3Electric current drop to zero.After this Q
1Continue conducting, Q
2Keep turn-offing, all rectifier diodes turn-off, V
InContinue to give L
In1Charging, the series resonance inductance L
rBe transfused to inductance L
In2Absorb, flow through L
In2Electric current flow into by L
pAnd C
pThe resonant network that constitutes, during pattern 1, C
pDischarge, its energy storage reduces.At t
1Constantly, Q
2Open-minded, pattern 1 finishes.
Pattern 2 (t
1-t
2): from t
1Constantly begin Q
1, Q
2Conducting simultaneously, 2 of A, B are by short circuit, and all rectifier diodes keep turn-offing the series resonance inductance L
rParticipate in L
pAnd C
pResonance, capacitor C
pEnergy stored makes primary current i
PriBegin commutation, flow through Q this moment
2Current i
Q2Increase, flow through Q
1Electric current t
Q1Reduce.At t
2Constantly, i
Q1Be reduced to zero, pattern 2 finishes.Design suitable L
rValue is vital, and this can guarantee overlapping in the switching tube conducting of former limit i in the time
Q1Be reduced to zero and have only very little overshoot, make Q
1Under the prerequisite of minimum circulation, realize zero-current switching.During pattern 2, V
InGive L simultaneously
In1And L
In2Charging, C
pVoltage continue to reduce.
Mode 3 (t
2-t
3): mode 3 and pattern 2 are similar, just i
Q1Direction is opposite, L
rContinue to participate in L
pAnd C
pResonance, work as i
PriResonance is during to peak value, C
pVoltage be zero, flow through L
pElectric current also reach peak value.At t
3Moment Q
1Turn-off, mode 3 finishes.During mode 3, C
pVoltage v
CpBegin after the zero passage to increase, but v
CpValue still less than v
o
Pattern 4 (t
3-t
4): pattern 4 is similar with mode 3, just t
3Moment Q
1Have no progeny in the pass, i
Q1Flow through Q
1The inverse parallel diode.At t
4Constantly, t
Q1Arrive zero through reversed peak resonance, Q
1Inverse parallel diode D
Q1Turn-off, pattern 4 finishes.By rational control Q
1Turn-off time can reduce the inverse parallel diode current flow time, thereby reduce inverse parallel diode on-state loss.
Pattern 5 (t
4-t
5): from t
4Constantly begin D
Q1Turn-off Q
2Continue conducting, L
rBy big inductance L
In1Absorb, flow through L
In1Electric current flow into by L
pAnd C
pThe resonant network that constitutes, V
InGive L
In2Charging.During this period, C
pVoltage continues to increase, up to t
5Constantly, v
CpEqual v
o, pattern 5 finishes.
Pattern 6 (t
5-t
6): t
5Constantly, v
CpEqual v
o, rectifier diode D
1And D
4Open-minded, L
In1Energy stored is by diode D
1And D
4Pass to load, V
InGive L
In2Charging, V
oGive L
pCharging.Flow through D
1And D
4Electric current equal the current i that the transformer secondary flows out
SecWith flow through L
pCurrent i
LpDifference, at t
6Constantly, i
LpEqual i
Sec, flow through D
1And D
4Electric current drop to zero, pattern 6 finishes.
As shown in figure 12, switching tube is operated in the ZCS state, and switching tube has less due to voltage spikes and current over pulse.v
ABThe voltage spine mainly be by switching tube Q
1And Q
2Inverse parallel diode D
Q1And D
Q2Reverse recovery cause.
As shown in figure 13, under the underloading condition, the rectifier diode ON time shortens, parallel resonance electric capacity was also reduced relatively by the time of clamper, the time of parallel resonant inductor and parallel resonance capacitor resonance can be elongated, the initial energy storage of parallel resonance electric capacity and series resonance inductance resonance is reduced, thus when underloading is worked i
PriLess overshoot is arranged, and the light-load efficiency of converter does not have too big influence.
Shown in Figure 14 and 15, under full load and underloading condition, rectifier diode all is operated under the ZCS condition.Under the underloading condition, the ON time of rectifier diode and v
CpThe clamped time can obviously reduce, the initial energy storage of parallel resonance electric capacity and series resonance inductance resonance is reduced, thereby can limit converter i when underloading is worked
PriOvershoot.
The waveform that shown in Figure 16 is when not having input filter capacitor, as seen from the figure, i
InLess ripple is arranged, and its frequency is the twice of switching frequency.
As seen from Figure 17, the light-load efficiency of converter is not serious reduces.
In sum, converter of the present invention can provide higher voltage gain, all has less circulation in full-load range, and switching tube and rectifier diode all are operated in the ZCS state, and also has higher efficient during underloading work.