CA2983328C - Constant current fast charging of electric vehicles via dc grid using dual inverter drive - Google Patents

Abstract

A DC charging circuit includes a first inverter module corresponding to a first battery; a second inverter module corresponding to second battery; and DC terminals tapping off a high-side of the first inverter module and a low-side of the second inverter module.

Description

CONSTANT CURRENT FAST CHARGING OF ELECTRIC VEHICLES
VIA DC GRID USING DUAL INVERTER DRIVE
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] This application claims all benefit including priority to U.S.
Provisional Patent Application No. 62/519,946, filed June 15, 2017, and entitled "CONSTANT CURRENT FAST CHARGING OF ELECTRIC VEHICLES VIA DC
GRID USING DUAL INVERTER DRIVE".
FIELD

[0002] Embodiments of the present disclosure relate generally to the field of electronic charging, and some embodiments particularly relate to the field of electronic charging of vehicles.
BACKGROUND

[0003] Electric vehicles have the potential to reduce energy consumption in the transportation sector which covers 27% of the total global consumption [1]. With their rapid deployment in the near future, consumers will expect greater drive range and fast charging rates. AC level 1 & 2, and DC charging are the presently available charging methods. DC charging is an attractive option over AC level 1 or 2 charging due to its potential to fully charge the electric vehicle in less than an hour [2]. The International Electrotechnical Commission (IEC) has established standardized connector protocols (CHAdeMO, Combined Charging System, etc.) that can be interfaced with charging systems fed by AC or DC mains [3].

[0004] Existing fast chargers require the electric vehicle supply equipment (EVSE) to be installed off-board due to physical size and mass limitations of the vehicle. The EVSE typically consists of a rectifier, LC filter, and high-power dc/dc converter. Unlike AC level charging units ($200-$300/ kW), DC
fast ($400/kW) are more costly in comparison due to increasing power level and system complexity [4]. Components rated for higher amperage CAN_DMS \109164075 \ 1 - 1 -contribute to the cost increase. Thus, lower component count and charger complexity are preferred.
SUMMARY

[0005] Existing integrated chargers are configured to charge from single or three-phase AC networks. With the rapid emergence of DC grids, there is growing interest in the development of high-efficiency, low-cost integrated chargers interfaced with DC power outlets. This application describes a new integrated charger which in some embodiments may offer electric vehicle fast charging from emerging DC distribution networks. In absence of a DC grid, the charger can alternatively be fed from a simple uncontrolled rectifier. The proposed charger leverages the dual inverter topology previously developed for high-speed drive applications. By connecting the charger inlet to the differential ends of the traction inverters, charging is enabled for a wide battery voltage range previously unattainable using an integrated charger based on the single traction drive. An 11 kW experimental setup demonstrates rapid charging using constant current control and energy balancing of dual storage media. To minimize the harmonic impact of the charger on the DC distribution network, a combination of complementary and interleaved switching methods is demonstrated.

[0006] In accordance with one aspect, there is provided A DC charging circuit including: a first inverter module corresponding to a first battery; a second inverter module corresponding to second battery; and DC terminals tapping off a high-side of the first inverter module and a low-side of the second inverter module.
DESCRIPTION OF THE FIGURES

[0007] Reference will now be made to the drawings, which show by way of example embodiments of the present disclosure.

[0008] FIG. 1 shows five different examples of charger topologies (a)-(e).

[0009] FIG. 2 shows an example dual inverter charger.

[0010] FIG. 3 shows an example circuit model of an upper module (a) and and an average model of a dual inverter integrating identical DC sources. In some embodiments, switch averaging can model each of the six half-bridges as an ideal voltage source.

[0011] FIG. 4 shows Phase "a" voltage and current waveforms ford = 0.53.

[0012] FIG. 5 shows a chart illustrating a normalized inductor current ripple.

In some embodiments, inductor current ripple size varies with conversion ratio, where Vo = V1 = V2. When each battery pack has nominal voltage near the input DC voltage, the operating region near 1:1 voltage ratio may achieve optimal ripple reduction.

[0013] FIG. 6 shows an example complementary and interleaved switching sequence for inner switches operated at d = 0.53. di., and d21 are mapped to inner switches S11 and S21, respectively. The most significant harmonic frequencies are shown.

[0014] FIG. 7 shows a comparison of i1 with and without interleaved switching, at d = 0.53. Phase currents in the top plot overlap when interleaving is not applied. Interleaved switching increases the ripple frequency and reduce peak-to-peak ripple.

[0015] FIG. 8 shows an example control diagram for controlling current.

[0016] FIG. 9 shows example simulation results of constant current control with sref step from 22 A to 44 A. Difference between -out]. and Lutz is due to voltage balancing controller acting on voltage mismatch.

[0017] FIG. 10 shows example simulation results of voltage balancing control.
V1 and V2 have a 7V deviation at t = 0.

[0018] FIG. 11 shows example simulation results of switching ripple in i =s,abcf idcf Ii, and i2, showing cancellation of most significant harmonic(s).

[0019] FIGS. 12A and 12B show an example laboratory prototype of 11 kW
dual inverter charger with a salient-pole rotor mimicking a permanent-magnet rotor. FIG. 12A shows a circuit diagram and FIG. 23B shows an experimental setup.

[0020] FIGS. 13A and 13B show example experimental results of constant current control at operating points (a) V1 = V2 = 175V , VC = 230V and (b) V1 = V2 = 245V , VC = 230V . The input current is initially stepped up to its rated value (45 A), and then stepped down by 50% at t = is.

[0021] FIGS. 14A and 14B show example experimental results of switching ripple for 'dc, is,abcf ilf and i2 using the described example switching method.
FIG 14A is a current waveform, and FIG. 14B is a Fourier spectrum of current ripple.

[0022] FIG. 15 shows example experimental result of voltage balancing control. Supercapacitor banks are pre-charged with 7V deviation, and controller regulate Ad to achieve voltage balance.
DESCRIPTION OF EXAMPLE EMBODIMENTS

[0023] To address charger complexity, combined traction and charging systems have been studied extensively in the past decade. The concept is to configure on-board traction components for charging, thus eliminating or greatly reducing the complexity of battery chargers. Subotic et al. proposed an integrated charger based on a 9-phase traction system [5]. As shown in Fig. 1(a), the machine's neutral points can be directly connected to a three-phase AC input, thus requiring no additional hardware between the AC grid and traction system. This topology also produces no net torque for vehicle propulsion in the charging process. Other multiphase machines for integrated charging are summarized in [6]. In terms of integrated charging via single-phase AC systems, Fig. 1(b) shows the topology proposed by Pellegrino et at.
It employs the traction system as a PFC boost converter, which is interfaced to a single-phase AC source via rectifier [7]. In Fig. 1(c), Tang et at. used a set of parallel-connected traction inverters and two motors to charge from a single-phase AC source and thereby eliminates the need for the rectifier [8].
In either topology, the charger requires no additional dc/dc converters, thus addressing weight, volume, and cost considerations of the EVSE. However, in both cases the minimum allowable battery voltage must always exceed the peak voltage of the AC mains.

[0024] The integrated chargers previously discussed are specifically for single-phase or three-phase AC systems. Due to the rapid penetration of renewables, grid-connected storage and DC-supplied loads, there is already significant effort in integrating DC micro grids within existing AC networks [9]. Ideally future EV chargers would accommodate charging from both existing DC fast chargers as well as from DC microgrid networks.

[0025] In some embodiments described herein, an integrated charger can offer, in some situations, electric vehicle fast charging from emerging DC
distribution networks. It leverages the existing dual inverter drive to operate as aforementioned integrated chargers, with the added benefits of improved voltage range and harmonic performance. The dual inverter traction system may, in some situations, provide increased speed range and battery integration without use of dc/dc power converters or additional magnetic materials, thus may offer an efficient and light-weight solution attractive for electric vehicles. Although two inverters are required, there is marginal increase in cost because each inverter stage is rated for half the total processing power. The dual inverter can, in some situations, facilitate power transfer between two isolated DC sources and the open-ended windings of the motor via differential connection of two voltage source converters. From previously proposed applications of the dual inverter for all-electric vehicles, the energy source is either a split-battery pack or a battery and floating capacitor bridge [11], [12]. The dual inverter configuration may, in some situations, offer voltage boost from the secondary inverter to enable high speed operation, improved efficiency at high speed, modular battery installation, and hybrid energy storage integration [10]-[15].

[0026] A challenge associated with the dual inverter drive is the need to charge two independent batteries. Hong et. al demonstrated that a single charger could be utilized for charging both batteries [16]. Shown in Fig.
1(d), the primary battery is charged using a standalone charger, while the secondary battery is charged from the first via the traction system.

[0027] In some embodiments, the present application describes a means which may, in some instances, eliminate the standalone charger in cases where DC power network access is available. The topology can be backwards compatible to conventional DC fast charging infrastructure. The proposed charger in this work is shown in Fig. 1(e). Contrary to other integrated chargers discussed earlier, placing the DC input at the differential connection of the traction system may enable rapid charging of dual storage media without a standalone charger. The topology may address the limited voltage range in the single inverter charging systems by using the series connection of two traction inverters, thus providing charging functionality even when the battery is at low state-of-charge. While the embodiments described below focus on vehicle charging, in some embodiments, the topology can be capable of bi-directional energy exchange with an external DC power network.

[0028] In some situations, embodiments of the present application may provide: an integrated charger suited for emerging DC networks, where fast charging is enabled by direct connection to a DC source; improved input voltage range using differential connection of dual inverter topology, requiring no external hardware; and/or a switching method utilizing complementary and interleaved phase shift to improve harmonic performance compared to single inverter systems.

[0029] The new architecture may offer rapid EV charging from the emerging DC grid with the potential to reduce charger cost, weight, and complexity by integrating charging functionality into the traction system.
TOPOLOGY

[0030] An example DC charging configuration is shown in Fig. 2. For the purpose of this paper, switches, voltage and current quantities for the upper and lower modules are labeled "1" and "2", respectively. The EV battery pack, consisting of n-strings, is split evenly between a pair of 2-level voltage source inverters. Each battery string has the same number of cells per string, thus maintaining the same nominal voltage as the combined battery pack.
The AC side is connected to the open-ended windings of the electric motor such that the machine leakage inductance is shared between the two switch networks.

[0031] A feature of the example dual inverter drive not previously exploited is its ability to leverage differential connections for EV charging. The DC
terminals tap off the high-side of module 1 and low-side of module 2. Power can be fed directly from a DC microgrid without a dc/dc intermediate stage.
Each set of 3 half-bridge switch networks is connected in a cascaded manner with the DC input and batteries to account for any voltage mismatch. In addition, the dual battery pack enables doubling of the motor voltage. Unlike the single traction-based integrated charger in Fig. 1(b), this permits charging even when the voltage in each battery pack is less than the DC
input voltage. This may be crucial for future trends in bulk power transfer, where fast charging stations are expected to support up to 1000 V at the vehicle inlet [3], [17].

[0032] Another potential benefit of utilizing two traction inverters is current ripple reduction. Since the motor leakage inductance, Ls, is limited by the magnetics of the EV motor, it is beneficial to minimize potentially high ripple component via controls. Thus, two types of switching methods are deployed.
The combination of 180 . phase shift between upper/lower cells, and 120 .
interleaving between parallel phases both reduce switching ripple in , and Complementary switching is not feasible for the integrated charger in Fig.
1(b).

[0033] Power transfer between the DC input and each battery unit is achieved by regulating the inductor currents. Its principle of operation is akin to the single string multi-port dc/dc converter developed in [18], however, the developed converter is reconfigured for 3-phase motor drives in this work.
OPERATION

[0034] In some embodiments, the dual inverter is configured to operate as a set of dc/dc converters in charging mode, as opposed to performing dc/ac conversion in traction mode. Its principle of operation is analyzed via the average model depicted in Fig. 3. This section also highlights the impact of complementary and interleaved switching on harmonic performance.
A. Average Model

[0035] The average model of the dual inverter is developed for identical energy storage integration, as in the case of the split-battery pack. Battery pack balancing will be addressed in Section IV. A dynamic model of the half-bridge network for a multilevel converter was developed in [19], but can also be used to represent the average switch model. Each of the six half-bridge converters is modeled as an ideal, controlled voltage source. The voltage depends on the duration in which the storage unit is inserted. The battery currents, il and i2, are derived from power balance. Although power flow can be bidirectional, this work identifies Vdc as the input and V1 & V2 as outputs.

[0036] In Fig. 3(a), each half-bridge is modeled as:
= ( 1) V2i = (2) where i = {a, b, c} for 3 interleaved dc/dc stages.

[0037] Only the switch network in the upper module is shown because the two inverters are identical, except V21 is the average voltage measured across the bottom set of switches instead of the top. As shown in (1) and (2), the duty cycle regulates the duration in which each battery voltage, V1 and V2, is inserted. Thus, the average voltage across each set of switches is a fraction of the associated battery voltage. Switch averaging for a single half-bridge was also discussed in [20].

[0038] Note that the following relation = d1 (3) (12 = (19i (4) is valid for this analysis assuming identical half-bridge switch networks top and bottom.

[0039] Applying KVL to any arbitrary phase (neglecting losses), the voltage conversion ratio is VdC = V. 11i (5) Assuming d11 = c12; = d for an idealized symmetric system yields:
Vde= (Vl +1/2)d (6a) TABLE I. Switching States 59i Upper module Lower module on on insert insert on off insert bypass off on bypass insert off off bypass bypass + V9 (6b) Vde

[0040] Notice the conversion ratio is similar to that of the boost converter, >
suggesting l+V2 'I( to enable boost operation. This is not a limiting factor for EV charging because the charging station's DC output voltage is 60 V to 500 V [3], and each string of EV battery cells spans from 300V to 500V
[21]. By assigning one battery string to each module, the minimum output voltage always exceeds the input voltage. Furthermore, the battery management system shall not permit the battery to discharge below the minimum voltage specified by the manufacturer.

[0041] Figure 3 also shows that the DC input current is the sum of the inductor currents:
= isa isb se (7) Output currents i and i2 can be derived from power balance:
i1/ui = di Usti isb isc) (8a) = (8b) i2 = idc(12 (8c) where i1 and i2 are fractions of the DC input current set by the duty cycle in each module.

[0042] Using (8), the average power supplied to each battery pack is = V d (9a) P.) =V2ith.d9 (9b) The average current into the battery is thus a function of the combined stator currents and duty cycle. Through proper switching action of the half-bridge switch networks, the proposed charger can effectively control the individual battery pack currents.
B. Switching Sequence , ,

[0043] For the remainder of this paper, d11 and d21 are mapped to inner switches SI., and S2i, respectively. For instance, 1.
5,1 a , ( t. ) = 01 1 r (10) ,t, < d1 õ Tsõ, {
u , alai sw < t < Tsui

[0044] 1) Complementary switching: A complementary strategy is applied to switches between the upper and lower modules. Thus, the following analysis examines the impact of complementary switching on phase "a". Gating signals for the inner switches, Vsa' isa' ila' and i2a are shown in Fig. 4.
Under balanced load conditions, each pair of "inner" and "outer" switches have the same percentage on-time in one switching period. However, the gating .
pulses between the two modules can be phase-shifted by 180 as demonstrated in [18]. This strategic overlap of gating pulse reduces the energy variation in the inductor, resulting in half the ripple current at twice the switching frequency.

[0045] The peak-to-peak inductor current ripple for V1 = V2 = Vo (idealized symmetric system) is (Vic ¨ -17,) _____________________________ di T., ,, , (11a) L, V i T . , 1 17 A : = ( ( ,s a ( 1 Vo c) ( 1 A v dc (lib) where the second expression is derived by combining (6b) and (11a).
Plotting (11b) in Fig. 5, this expression highlights one of the key features of this topology: the inductor energy variation, or current ripple, depends on the voltage difference Vdc ¨Vo. Notice for the case where the battery packs are balanced, and V1 = V2 = Vdc, this yields zero inductor current ripple. The ideal operating range is centered around Vde to minimize distortion in the supply lines.

[0046] The branch current of i1 and i2 from any arbitrary phase, denoted by pulsates due to the discontinuous conduction of the switch network:
= (12) 12i = isiS2i (13) Notice that the inductor ripple also propagates into the battery. Since the inductor ripple is negligible relative to the pulsating current generated by summing the branch currents, complementary switching has minimal effect on the battery currents. Thus, to minimize current harmonics in the batteries, interleaved switching between parallel phases is used. The proposed switching method also reduces the switching ripple at the DC input.

[0047] 2) Interleaved switching: This switching strategy has not been previously studied in an integrated charger based on the dual inverter. As shown in Fig. 6, the gating pulses between phase a, b, and c are phase shifted by 120 . This further reduces the peak ripple observed in idc. Due to the phase-shift of stator currents, the peak-to-peak 'dc is approximately 1/3 of the ripple generated using in-phase switching, and the most significant switching component is shifted to the 6th harmonic.

[0048] Figure 7 shows the impact of phase interleaving on output currents and i2. As discussed previously, the currents in all switches are "chopped"
regardless of the switching pattern. The unfiltered battery currents are the sum of the pulsating currents in the inner switches:
== ila -F ilb ;lc (14) i2 " i2a -11r- i9b -F 12c (15)

[0049] To minimize the switching ripple due to discontinuous conduction, interleaved switching enables continuous conduction of ii and i2 for < d < 1- . The battery currents conduct through at least one of the 3 phases. The third plot in Fig. 7 shows that at d = 0.53, interleaving results in approximately of the ripple component, and the most significant harmonic is shifted to 3fsw. The contribution of the inductor current ripple to the total harmonic distortion in i1 and i2 is negligible at this operating point.

[0050] In summary, the proposed switching sequence produces Ais,abc, /id, and ii,2 at 2f8w,6f8w, and 3fsw, respectively. This effectively leads to reduced THD and semiconductor losses. Reduction in peak-to-peak output current ripple also helps to prevent battery capacity fade and impedance degradation [22].

[0051] Recall that an ideal, symmetrical system having balanced energy sources was studied in previous sections. This allows the controller to set equal duty cycles to both the upper and lower modules. To address the scenario where the isolated battery packs have a different state-of-charge during the charging process, the duty cycles are decomposed into sum and difference terms, defined as:
di 1 ¨ Ed -= d9 T ( 1 6) 75 Ad

[0052] In some instances, the objective of the DC charger may be to 1) regulate the DC inductor current using the sum component 2) equalize the stored energy in the split energy source using the difference component.
Note that coupling between the two terms may be present.
A. Inductor current control

[0053] In Fig. 8, three PI controllers are implemented for constant current control of parallel phases. Since the EVSE typically regulates the DC current at the vehicle inlet, each inductor current will track one-third of the DC bus current reference.

[0054] An expression for the dynamics of the system is developed by applying KVL to the average model:
al Vde d /7.42i + isiRs + Ls __ = 0 (17a) (It Vdc (vt __ +9142 )Edi (1/1 V2),Adi 'Si 17b) Rs +
where d1, and d21 have been replaced by Id and Ad as per (16). Ideally, if the battery voltages are balanced, then only the sum term drives the DC current.
However, the difference term is coupled to the current controller. To avoid stability issues, voltage balancing controller can be designed to have significantly slower response to voltage dynamics. Thus, (V1¨ V2) Ad, can be regarded as a DC offset in the time scale of the current controller.

[0055] The example controller discussed in this work is developed for constant current charging. The control scheme for constant voltage charging may be investigated in future works.
B. Energy balancing

[0056] In Fig. 8, the voltage balancing controller takes the voltage difference and outputs Ad, which is then subtracted from d11 and added to d2i=
Therefore, if the DC source in the upper module is overcharged relative to the lower, then the lower one will be inserted more frequently. Both sources are charged simultaneously but with an offset to shift the power distribution.

To ensure this offset does not exceed the operating limits of the converter, a limiter is implemented at the output of the voltage balancing controller. Note that the balancing controller uses voltage to extrapolate the total stored energy in the DC source. Other parameters may be used for energy management, such as comparing state-of charge (Coulomb count) of a split-battery pack.

SIMULATION RESULTS

[0057] A full-switch model of the proposed integrated charger is implemented in MATLAB/SIMULINK with a PLECS toolbox. The circuit diagram is shown in Fig. 12(a), and simulation parameters are listed in Table II.
TABLE II. Simulation Parameters Parameter Symbol Value Input power 50k W
Power/module P1, P9 25kW
DC bus voltage Vdc 380V
Initial SC voltage VI, V2 360V-365V
DC bus current idc 132A
Stator current is,abe 44A
Capacitance/SC bank c,õõ1, C9 16.6 F
Output capacitors Cl C2 9.6MF
Stator inductance , 0.8 rn I-1 Stator resistance R
Switching frequency f 7.5k Hz

[0058] In place of EV batteries, two supercapacitor banks are used in this simulation study to mirror the experimental system. The faster charge/discharge rates of the supercapacitor vs. a battery facilitates a less time consuming study of storage energy balancing algorithms. All current quantities are positive in the direction indicated by the arrow, which shows power transfer from the DC input to supercapacitors. This simulation study demonstrates = Current control and voltage balancing functionality = DC charging at operating point V1 < Vdc, V2 < Vdc, which is one limitation of previously proposed integrated chargers = Current ripple reduction using proposed switching method TABLE III. Experimental Parameters General Parameters Symbol Value Input power 'dc 10.35kW
Power/module P.1., P2 5.17k1V
DC bus voltage 230V
Case #1: 171 < Vdc, V2 < Vdc Initial SC voltage V1. V2 175V
Case #2: V1 > V. V2 > Vdc Initial SC voltage Vit ,11;, 245V
DC bus current idc .45A
Stator current is,abc 15A
Capacitance/SC bank Csci Csa 16.6F
Output capacitors C2 9.6m F
Switching frequency fs 7.5kHz.
Machine Parameters Symbol Value Power Prated Line-to-line voltage Vrated 2201' Line current irated 39.4.1 Stator inductance U.5m 1.1 Stator resistance Ps Rotor excitation current if 5A

[0059] 1) Constant current control: Fig. 9 shows the system response when a current step is applied at t = 0.1s. The inductor reference current, i -sref is stepped from 22 A to 44 A. This allows the total input power, DC bus current, and current into the supercapacitors to double accordingly. Id initially drops, as derived in (17b), to act on the increase in current demand and settles to its new value in 10 ms. After the transient, the charger operates at rated conditions (50 kW), which is the typical system rating for the CHAdeM0 EVSE
[23].

[0060] 2) Voltage balancing: Fig. 10 demonstrates the effect of voltage balancing control on energy distribution. The super-capacitor banks have a 7 V difference at t = 0, and achieves energy balance when V1 = V2. The delta term, Ad, regulates the rate of convergence. The voltage balancing response can also be observed in Fig. 9, where iout1 and iout2 are regulated such that P1 = 18kW and P2 = 32kW. If supercapacitors are balanced, then Ad = 0 to deliver 25 kW to each module.

[0061] 3) Harmonic analysis: Fig. 11 verifies the harmonic decomposition of is,abc, LC/ ti 1 and 12 for the balanced voltage operating scenario. The most significant harmonic frequencies in the inductors, DC bus, and supercapacitor prior to filtering are 2fsw, and 3fsw, respectively. Observe that for i1 and i2, the 6th harmonic from idc propagates to the output. However, it has negligible impact on output peak-to-peak ripple because the DC current is significantly larger than the inductor ripple.
EXPERIMENTAL RESULTS

[0062] This section discusses experimental testing of an 11 kW laboratory prototype based on the proposed charger topology. One of the most commonly adopted DC fast chargers (CHAdeM0) is rated at 50 kW. In this work, the system rating is scaled-down to verify basic charging functionality using a dual inverter powertrain. Experimental results show constant current control, voltage balancing, and switching ripple reduction in a wide operating region. Charging at two operating points will be validated: 1) V1 < Vdc, V2 <
Vdc, and 2) V1 > Vdc, V2 > Vdc. In either case, the system is operating at 94% of the rated power of the motor.

[0063] The laboratory setup is shown in Fig. 12, and system parameters in Table III. A Regatron power supply provides 230 V at the DC input, where the terminals represent the charging inlet of the vehicle. A 0.5 kWh supercapacitor bank is connected to each 2-level VSC. Each supercapacitor bank consists of 180 series-connected cells with 3000 F per cell. Thus, each string has total capacitance of 16.6 F. Permanent magnet synchronous motors (PMSM) and induction motors are the most commonly used electric motors in EVs. Thus, the wound rotor SM in the prototype is operated with constant field, similar to a PMSM. This is achieved by exciting the rotor windings to ensure rotor flux is present. The impact of rotor saliency on phase current ripple discussed below.

[0064] The control strategy in Fig. 8 can be implemented on a real-time linux PC controller with integrated FPGA.

[0065] A. Case #1: Charging at V1 < Vdc, V2 < Vdc

[0066] Figure 13(a) shows experimental results of constant current control when each supercapacitor voltage is less than the input voltage. This is analogous to charging a high-energy, low-voltage EV battery pack, or batteries at low state-of charge. The results demonstrate functionality of the controller when isref is stepped up from 0 to 15 A, and then stepped down to 50% of its rated current. The input current is shown to be the sum of the phase currents. The combined energy storage system, with 175 V per supercapacitor bank, charges from a 230 V DC supply at 10.35 kW rated power, hence charging batteries with power comparable to rated machine power. Similar to the case presented in simulation, idc and is,abc tracks the new current reference.

[0067] B. Case #2: Charging at V1 > Vdc, V2 > Vdc

[0068] Figure 13(b) shows experimental results of constant current control when each supercapacitor voltage exceeds the input voltage. This operating scenario applies to charging EV batteries designed for high-voltage, high-speed operation. The input voltage is fixed at 230 V and each supercapacitor bank charges at 245 V, and the total charging power is also 10.35 kW. The same current steps are applied to this operating point. As shown in Fig.
14(a), the peak-to-peak ripple between phase currents are not identical. Use of a salient-pole rotor leads to asymmetry in flux linkage between stator and rotor, which marginally affects the total inductance per phase.

[0069] C. Voltage Balancing

[0070] Fig. 15 demonstrates the functionality of voltage balancing control.
The supercapacitor voltages prior to charging are 154 V and 147 V. When the controller is enabled, the DC bus current steps from 0 to 10A, drawing 2.3 kW from the DC supply. Due to the applied offset between dl and d2, the "undercharged" supercapacitor bank has a faster rate of charge compared to the "overcharged" supercapacitor bank. The supercapacitor voltages converge at approximately 178 V. The results verify operation of the balancing controller in response to the initial voltage deviation.

[0071] D. Discussion of Switching Ripple and Rotor Saliency

[0072] Fig. 14(a) shows the switching ripple of idcr is,abcf i1,and i2 for case #1, but at lower current reference. This is to show that the magnitude of the peak-to-peak ripple is independent of the average charging current.
Neglecting switching noise in the current reference step from Fig. 13(a), the switching ripple between charging at 'dc = 15A and 'dc = 45A is identical.
Comparing the Fourier spectrum of the simulation and experimental study, the switching ripple at the switching frequency (7.5 kHz) is eliminated in both systems. Any discrepancy between simulation and experimental results is due to differences in operating point, and rotor saliency. For example, output currents i1 and i2 from laboratory results have higher 6th harmonic than 3rd in comparison with simulation results, where the 3rd harmonic is dominant.
This is due to the fact that the simulation model is operated at rated conditions. In the experimental work, charging at low currents introduces higher 6th harmonic ripple.

[0073] Also note that isb ripple components in Fig. 14 are noticeably smaller than the other two phases. This results from using a salient-pole rotor, where the phase inductance depends on the rotor's electrical position [7]. In the experimental results, the rotor was arbitrarily oriented to produce the asymmetric phase current ripple in Fig. 14(a). In Fig. 14, difference in phase current ripple increases the 2nd harmonic component in 'dc. However, the 6th harmonic is shown to be the dominant switching component in the input current.

[0074] Some embodiments of the present application present a new integrated charger topology that may offer direct charging from the DC grid without any off-board hardware. The concept is to connect the vehicle charging input to the differential ends of the dual traction system. Although a =
second converter is required, higher motor voltages and lower currents may be utilized, and the net switch VA rating remains unchanged.

[0075] In some instances, the proposed integrated charger based on the dual inverter has been demonstrated to enable charging over a wide voltage range. An 11 kW laboratory prototype verifies DC charging for supercapacitor voltages V1 and V2 above and below the DC input voltage. Furthermore, results show effective current control and energy balancing amongst the two supercapacitor banks, which are used in place of batteries to reduce experimental run-time. The proposed switching method may, in some instances, attenuate significant switching harmonics, which is essential for addressing the use of limited motor inductance as interface inductors. The control method for constant voltage charging will be studied in future works.
In practice, the proposed topology's charging rate is limited by thermal constraints of the motor and traction power electronics, thus highlighting its ability to charge at the rated power of the traction system ideal for electric vehicle fast charging.

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Claims (29)

What is claimed is:

1. A device adapted to provide both drive and charging functionality, the device comprising:
an electric motor in open stator winding configuration;
a first inverter circuit including a first traction inverter and a first energy storage device coupled to the electric motor and to a power source;
a second inverter circuit including a second traction inverter and a second energy storage device coupled to the electric motor and the power source; a connection between the positive terminal of the power source and the positive terminal of the first energy storage device and a connection between the negative terminal of the power source and the negative terminal of the second energy storage device such that the power source is coupled at a differential connection of the first traction inverter and the second traction inverter; and a controller circuit configured for providing constant current control of motor inductor currents of the first traction inverter and the second inverter such that each motor phase current of the motor inductor currents tracks one third of a current reference of the power source.

2. The device of claim 1 wherein the first inverter circuit and the second inverter circuit each comprise a set of three half-bridge switch networks connected in a cascaded manner with the positive and negative terminals and the first and second energy storage devices.

3. The device of claim 2, wherein the controller circuit is configured to control the set of three half-bridge switch networks with interleaved switching between the parallel phases with the 120 degree phase shifting established between three phases, phase a, phase b, and phase c, shifting a most significant switching sinusoidal component to a second, a third or a sixth harmonic frequency.

4. The device of claim 2, wherein the controller circuit is configured to control inverter switch networks corresponding to the first inverter circuit and the second inverter circuit such that complementary switching of switches between the first inverter circuit and the second inverter circuit include a 180 degree phase shift between the gating pulses, causing an overlap of the gating pulses that reduces an energy variation in an inductor of the device by halving the ripple current of the reduced ripple current waveform at a doubled switching frequency.

5. The device of claim 1, wherein the controller circuit is configured to control power distribution between the first inverter circuit and second inverter circuit to balance energy between the first energy storage device and the second energy storage device.

Date Recue/Date Received 2021-02-22

6. The device of claim 1, wherein the electric motor is mounted in a vehicle and the electric motor is configured for dual-mode operation comprising a first mode wherein the electric motor provides the drive functionality to impart forces to move the vehicle, and a second mode wherein the electric motor provides the charging functionality when electrically coupled to the power source.

7. The device of claim 1, further comprising a gating signal controller configured for providing fault blocking capability at the power source, protecting the first and the second energy storage devices.

8. The device of claim 1, wherein the differential connection is coupled to a DC microgrid free of a DC/DC intermediate stage.

9. The device of claim 1, wherein the device is configured for rapid charging of the first energy storage device and the second energy storage device free of a standalone charger.

10. The device of claim 1, wherein the device is configured for charging of the first energy storage device and the second energy storage device when at least one of the first energy storage device and the second energy storage device are at a low state of charge.

11. The device of claim 1, wherein the first energy storage device and the second energy storage device are EV energy storage device packs consisting of n-strings.

12. The device of claim 11, wherein the first energy storage device and the second energy storage device include evenly split pairs of 2-level voltage source inverters.

13. The device of claim 12, wherein the first energy storage device and the second energy storage device include energy storage device strings having a same number of cells per string.

14. The device of claim 1, wherein AC terminals of each of the first inverter circuit and the second inverter circuit are coupled to open-ended windings of an electric motor such that machine leakage inductance appears between the first inverter circuit and the second inverter circuit.

15. The device of claim 1, wherein each of the first inverter circuit and the second inverter circuit includes at least a set of half-bridge switch networks.

16. The device of claim 1, wherein each of the first inverter circuit and the second inverter circuit includes a set of 3 half-bridge switch networks.
Date Recue/Date Received 2021-02-22

17. The device of claim 16, wherein each set of 3 half-bridge switch networks is coupled in a cascaded topology with a DC input and the first energy storage device and the second energy storage device to account for any voltage mismatch.

18. The device of claim 1, wherein the first inverter circuit and the second inverter circuit include a corresponding upper set of half-bridge switch networks and a corresponding lower set of half-bridge switch networks.

19. The device of claim 18, wherein the upper set of half-bridge switch networks and the lower set of half-bridge switch networks have a phase shift of 180 degrees.

20. The device of claim 18, wherein parallel phases of signals of the upper set of half-bridge switch networks and the lower set of half-bridge switch networks have a phase shift of 120 degrees.

21. The device of claim 18, wherein the upper set of half-bridge switch networks and the lower set of half-bridge switch networks have a phase shift of 180 degrees; and wherein parallel phases of signals of the upper set of half-bridge switch networks and the lower set of half-bridge switch networks have a phase shift of 120 degrees.

22. A method for controlling charging input from a power source for a device adapted to provide both drive and charging functionality, the method comprising:
providing constant current control of motor inductor currents of a first traction inverter and a second inverter such that each motor phase current of the motor inductor currents tracks one third of a current reference of a power source;
wherein a first traction inverter circuit includes the first traction inverter and a first energy storage device coupled to an electric motor in open stator winding configuration and to a power source;
wherein a second traction inverter circuit includes second first traction inverter and a second energy storage device coupled to the electric motor in open stator winding configuration and to the power source;
and wherein the power source is coupled at a differential connection of the first traction inverter and the second traction inverter.

23. The method of claim 22, comprising controlling switch networks of the first inverter circuit and the second inverter circuit with interleaved switching between the parallel phases with the 120 degree phase shifting established between three phases, phase a, phase b, and phase c, shifting a most significant switching sinusoidal component to a second, a third or a sixth harmonic frequency.

Date Recue/Date Received 2021-02-22

24. The method of claim 22, comprising controlling inverter switch networks corresponding to the first inverter circuit and the second inverter circuit such that complementary switching of switches between the first inverter module and the second inverter circuit include the 180 degree phase shift between gating pulses, causing an overlap of the gating pulses that reduces an energy variation in an inductor of the device by halving the ripple current of the reduced ripple current waveform at a doubled switching frequency.

25. The method of claim 22, comprising coupling the first inverter circuit and the second inverter circuit to the electric motor.

26. The method of claim 25, wherein the electric motor is coupled to an electric vehicle.

27. The device of claim 1, wherein the first energy storage device is a battery and the second energy storage device is a battery.

28. The device of claim 1, wherein the first energy storage device is a battery and the second energy storage device is of a different type.

29. The device of claim 28, wherein the second energy storage device is a supercapacitor.

Date Recue/Date Received 2021-02-22