WO2018051719A1 - Inverter apparatus and vehicle electric compressor provided with same - Google Patents

Inverter apparatus and vehicle electric compressor provided with same Download PDF

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Publication number
WO2018051719A1
WO2018051719A1 PCT/JP2017/029582 JP2017029582W WO2018051719A1 WO 2018051719 A1 WO2018051719 A1 WO 2018051719A1 JP 2017029582 W JP2017029582 W JP 2017029582W WO 2018051719 A1 WO2018051719 A1 WO 2018051719A1
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WO
WIPO (PCT)
Prior art keywords
power semiconductor
loss
semiconductor element
junction temperature
calculation unit
Prior art date
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PCT/JP2017/029582
Other languages
French (fr)
Japanese (ja)
Inventor
順貴 川田
大輔 廣野
Original Assignee
サンデン・オートモーティブコンポーネント株式会社
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Application filed by サンデン・オートモーティブコンポーネント株式会社 filed Critical サンデン・オートモーティブコンポーネント株式会社
Priority to DE112017004631.9T priority Critical patent/DE112017004631T5/en
Priority to CN201780055588.2A priority patent/CN109831931A/en
Publication of WO2018051719A1 publication Critical patent/WO2018051719A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/327—Means for protecting converters other than automatic disconnection against abnormal temperatures
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/60—Controlling or determining the temperature of the motor or of the drive
    • H02P29/68—Controlling or determining the temperature of the motor or of the drive based on the temperature of a drive component or a semiconductor component

Definitions

  • the present invention relates to an inverter device that operates a motor of an electric compressor, for example, and a vehicle electric compressor including the same.
  • an electric compressor is used as a refrigerant compressor.
  • This electric compressor drives a compression element by a motor fed from a vehicle battery, and this motor is operated by an inverter device.
  • This type of inverter device controls the energization of each phase of the motor by switching a power semiconductor element (IGBT, MOSFET, etc.) having a bridge configuration.
  • IGBT power semiconductor element
  • MOSFET MOSFET
  • This junction temperature is the temperature of the chip inside the power semiconductor element (surface temperature of the IGBT chip, MOSFET chip, and FWD chip), and the temperature of the substrate on which the power semiconductor element is mounted (near the power semiconductor element).
  • the temperature rise value corresponding to the amount of heat generated by the loss consisting of the switching loss and steady loss (conduction loss or conduction loss) of the power semiconductor element is detected by the temperature sensor (temperature detector). It is obtained by adding (for example, refer patent document 1).
  • FIG. 5 is a diagram for explaining a conventional protection method based on the junction temperature.
  • the vertical axis indicates the phase current value of the inverter circuit composed of power semiconductor elements in a bridge configuration
  • the horizontal axis detects the temperature of the substrate on which the power semiconductor elements are mounted (the temperature in the vicinity of the power semiconductor elements).
  • the detected value of the temperature sensor The broken line shown in FIG. 5 indicates that the current is cut off when the phase current reaches the predetermined value Asstop until the detection value of the temperature sensor rises to T1, and the detection value of the temperature sensor rises from T1 to T2.
  • the protection threshold value for performing the blocking with a value smaller than the predetermined value Asstop (the oblique line in the broken line in FIG.
  • this protection threshold value is as follows. That is, assuming that the voltage (HV voltage) applied from the vehicle battery (HV power supply for the vehicle) is a maximum value (for example, 300 V), the applied voltage (HV voltage) is based on the characteristics of the power semiconductor element. ) And the phase current value, the loss (heat generation amount) of the power semiconductor element is calculated, and the temperature rise value is calculated in advance from this loss (the relationship between the phase current and the temperature rise value is determined by the maximum applied voltage). It is obtained in advance as a value).
  • the protection threshold value (broken line) in FIG. 5 is shown. Conventionally, based on the protection threshold value shown in FIG. 5, for example, until the detected value of the temperature sensor at that time reaches T1, the power semiconductor element that cuts off the current when the phase current value rises to Asstop. Had done protection.
  • this protection threshold is the maximum value of the applied voltage (HV voltage), that is, the worst-case protection threshold, when the applied voltage (HV voltage) is low, the current is applied at a stage where it is not necessary to perform protection. There was a drawback of blocking.
  • the phase current is assumed to be a sine wave, and the loss is calculated from an average value of a relatively long period.
  • the applied voltage is modulated, the waveform of the phase current is not an ideal sine wave, but includes ripples and harmonic components. For this reason, even if the average value is not known, there is an instantaneous situation where the temperature limit of the power semiconductor element is exceeded. In the past, this could not be accurately determined and protected.
  • the present invention has been made to solve the conventional technical problems, and an inverter device capable of protecting a power semiconductor element with high accuracy from an instantaneous temperature rise and a vehicle using the same An object is to provide an electric compressor.
  • An inverter device of the present invention includes an inverter circuit having a power semiconductor element having a bridge configuration and an inverter control unit having a PWM control unit for driving the power semiconductor element, and is provided in the vicinity of the power semiconductor element.
  • a temperature detector for detecting the temperature and a phase current detector for detecting the phase current of the inverter circuit are provided.
  • the inverter control unit is a power semiconductor element based on at least one phase current detected by the phase current detector and the applied voltage. Add the temperature rise value obtained from the loss of the power semiconductor element calculated by the loss calculation section to the temperature detected by the loss detector and the temperature detector that calculates the loss of the power, and calculate the junction temperature of the power semiconductor element.
  • a junction temperature estimation calculation unit for estimation, and for each PWM carrier period in the PWM control unit, the loss calculation unit of the power semiconductor element The junction temperature of the power semiconductor element is estimated by the junction temperature estimation calculation unit, and the junction temperature of the power semiconductor element for each PWM carrier period estimated by the junction temperature estimation calculation unit exceeds a predetermined value. In this case, a predetermined protection operation is performed.
  • the inverter device according to a second aspect of the invention is characterized in that, in the above invention, the PWM carrier cycle is sufficiently smaller than the frequency of the phase current of the inverter circuit and sufficiently shorter than the thermal time constant of the power semiconductor element.
  • the loss calculation unit calculates the loss of the power semiconductor element from the switching loss and the steady loss of the power semiconductor element, and the junction temperature estimation calculation unit calculates the loss.
  • the temperature rise value is calculated by multiplying the loss of the power semiconductor element calculated by the unit by the thermal resistance value of the power semiconductor element.
  • the loss calculation unit calculates the loss of the power semiconductor element of each phase of the bridge configuration from the phase current and applied voltage of each phase of the inverter circuit, and performs inverter control. The unit performs a protection operation based on a junction temperature of the highest power semiconductor element.
  • an inverter device according to any of the above-mentioned inventions, wherein the power semiconductor element is a composite of a semiconductor switching element and a free wheel diode, and the loss calculation unit calculates the loss of the semiconductor switching element and the free wheel diode
  • the junction temperature estimation calculation unit estimates the junction temperature of the semiconductor switching element and the junction temperature of the free-wheeling diode.
  • the inverter control unit limits the current flowing through the inverter circuit when the junction temperature of the power semiconductor element exceeds the first predetermined value, and the junction temperature is When the second predetermined value higher than the first predetermined value is exceeded, the current flowing through the inverter circuit is cut off.
  • a vehicle electric compressor including a motor that is operated by the inverter device according to each of the above inventions, and is mounted on the vehicle.
  • an inverter device including an inverter circuit having a power semiconductor element having a bridge configuration and an inverter control unit having a PWM control unit for driving the power semiconductor element
  • the temperature in the vicinity of the power semiconductor element is increased.
  • a temperature detector to detect and a phase current detector to detect the phase current of the inverter circuit, and the inverter control unit detects the power semiconductor element from at least one phase current detected by the phase current detector and the applied voltage. Add the temperature rise value obtained from the loss of the power semiconductor element calculated by the loss calculation section to the temperature detected by the loss detector and the temperature detector that calculates the loss of the power, and calculate the junction temperature of the power semiconductor element.
  • the power semiconductor element can be protected with high accuracy even from a temperature rise.
  • the loss calculation section calculates the loss of the power semiconductor element from the switching loss and the steady loss of the power semiconductor element as in the invention of claim 3, and the junction temperature estimation calculation section calculates the loss calculation section. If the temperature rise value is calculated by multiplying the loss of the power semiconductor element by the thermal resistance value of the power semiconductor element, the instantaneous value of the junction temperature of the power semiconductor element is accurately calculated and estimated. Will be able to.
  • the loss calculation unit calculates the loss of the power semiconductor element of each phase of the bridge configuration from the phase current and applied voltage of each phase of the inverter circuit, and the inverter control unit calculates the highest power.
  • the protection operation is executed based on the junction temperature of the semiconductor element for power, the power semiconductor element in the phase where the temperature rises most rapidly can be safely and accurately protected.
  • the loss calculation unit calculates the loss of the semiconductor switching element and the free-wheeling diode, If the temperature estimation calculation unit estimates the junction temperature of the semiconductor switching element and the junction temperature of the free-wheeling diode, the temperature estimation calculation unit can protect the power semiconductor element having the free-wheeling diode without any trouble.
  • the inverter control unit limits the current flowing through the inverter circuit when the junction temperature of the power semiconductor element exceeds the first predetermined value, and the junction temperature is less than the first predetermined value.
  • the high second predetermined value is exceeded, by interrupting the current flowing through the inverter circuit, it is possible to avoid unnecessary current interruption while reliably protecting the power semiconductor element. It becomes.
  • a very effective overheat protection can be implement
  • FIG. 2 It is a schematic sectional drawing of the electric compressor for vehicles of one Example to which the inverter apparatus of this invention is applied. It is an electric circuit diagram of the inverter apparatus of one Example of this invention. It is a figure which shows the switching control of the inverter apparatus of FIG. 2, the loss of a semiconductor switching element and a free-wheeling diode, and the estimated value of junction temperature. It is a figure which shows the flow of estimation calculation of the junction temperature which the inverter control part of FIG. 2 performs. It is a figure explaining the conventional protection system by junction temperature.
  • FIG. 1 shows a schematic sectional view of a vehicular electric compressor 1 to which the present invention is applied.
  • the electric compressor 1 according to the embodiment constitutes a part of a refrigerant circuit of an air conditioner that air-conditions the interior of a vehicle (not shown), and is mounted in an engine room of the vehicle.
  • the electric compressor 1 includes a motor 3 in a housing 2 and a scroll-type compression element 6 driven by a rotating shaft 4 of the motor 3.
  • An inverter device 7 of the present invention is further attached to the housing 2, and the motor 3 is operated by the inverter device 7 to drive the compression element 6.
  • FIG. 2 shows an electric circuit diagram of the inverter device 7.
  • the inverter device 7 includes a control board 11 on which an inverter circuit 8 and a smoothing capacitor 9 are mounted, and an inverter control unit 12 configured by a microcomputer (processor).
  • the positive DC bus 13 of the inverter circuit 8 is connected to the + terminal of a vehicle battery (HV power supply for vehicle) B (not shown), and the negative DC bus 14 is connected to the negative terminal of the battery B.
  • the smoothing capacitor 9 is connected between the two DC buses 13 and 14 of the inverter circuit 8.
  • the inverter circuit 8 changes the switching state of the plurality of power semiconductor elements constituting the bridge, converts the direct current applied from the battery B into alternating current, and supplies the alternating current to the motor 3. Specifically, three power semiconductor elements 16U, 16V, and 16W constituting the upper phase of the bridge and three power semiconductor elements 17U, 17V, and 17W constituting the lower phase of the bridge are provided. Each of the power semiconductor elements 16U, 16V, 16W and 17U, 17V, 17W is a composite of a semiconductor switching element 18 and a freewheeling diode 19 connected in reverse parallel thereto. DC power is supplied from the battery B to the buses 13 and 14.
  • the upper-phase power semiconductor elements 16 U, 16 V, and 16 W and the lower-phase power semiconductor elements 17 U, 17 V, and 17 W are one-to-one. Correspondingly, they are connected in series.
  • the pair of semiconductor switching elements 18 of the power semiconductor elements 16U to 17W connected in series is referred to as a switching leg.
  • the switching leg 21U configured by a pair of the semiconductor switching element 18 of the power semiconductor element 16U and the semiconductor switching element 18 of the power semiconductor element 17U, the semiconductor switching element 18 of the power semiconductor element 16V, and the power A switching leg 21V configured by a pair of semiconductor switching elements 18 of the semiconductor element 17V, a switching leg 21W configured by a pair of the semiconductor switching elements 18 of the power semiconductor element 16W and the semiconductor switching elements 18 of the power semiconductor element 17W, There is.
  • the switching legs 21U, 21V, and 21W are connected between the positive DC bus 13 and the negative DC bus 14, respectively.
  • the intermediate points MU, MV, MW of the respective switching legs 21U, 21V, 21W are nodes for outputting the phase voltages Vu, Vv, Vw of each phase (U phase, V phase, W phase) of the output AC.
  • Each intermediate point MU, MV, MW is connected to each phase of the motor 3.
  • the semiconductor switching element 18 uses an IGBT (Insulated Gate Bipolar Transistor).
  • the semiconductor switching element 18 is not limited to the IGBT and may be a MOSFET or the like.
  • a temperature sensor 22 as a temperature detector is mounted on the control board 11 in the vicinity of the power semiconductor elements 16U to 17W. In this embodiment, the temperature sensor 22 is a thermistor.
  • a shunt resistor 23 as a phase current detector is connected to the negative DC bus 14 at a position where current from the motor 3 flows.
  • the phase current detector is not limited to the shunt resistor, and may be configured with a current transformer or the like.
  • the inverter control unit 12 includes a motor control unit 26, a PWM control unit 27, a current detection unit 28, a gate driver 29, a loss calculation unit 31, a junction temperature estimation calculation unit 32, and a temperature protection unit 33. I have.
  • the HV voltage (applied voltage) of the DC bus 13 on the positive side is input to the PWM control unit 27 and the loss calculation unit 31.
  • the motor control unit 26 outputs a target waveform (modulated wave) of the three-phase sine wave applied to the motor 3 to the PWM control unit 27.
  • the PWM control unit 27 generates a duty (Duty: upper phase ON time) that is a drive signal by comparing the level of the modulated wave output from the motor control unit 26 with the level of the carrier (triangular wave). This duty is generated for each of the U-phase, V-phase, and W-phase, and is sent to the gate driver 29 that drives (ON-OFF) the gate of each semiconductor switching element 18.
  • the frequencies of the phase currents Iu, Iv, Iw, which are the rotation speeds of the motor 3 of the embodiment, are 400 Hz to 500 Hz, and the carrier cycle (hereinafter referred to as PWM carrier cycle) in the PWM control unit 27 is higher than that. It is 20 kHz which is sufficiently small (or short enough).
  • the thermal time constant of the power semiconductor elements 16U to 17W (the time taken to transmit the loss temperature rise value to the temperature sensor 22) is about 50 msec, and the PWM carrier cycle is sufficiently shorter than this thermal time constant. (Or fast enough).
  • the current detector 28 receives the voltage across the shunt resistor 23 and calculates the phase currents Iu, Iv, Iw from the resistance value of the shunt resistor 23.
  • the calculated phase currents Iu, Iv, Iw are input to the loss calculator 31.
  • the loss calculation unit 31 includes the phase currents Iu, Iv, Iw of the U-phase, V-phase, and W-phase input from the current detection unit 28 and the HV voltage (applied voltage) of the DC bus 13 on the positive side. Based on the duty input from the PWM control unit 27, the loss of each of the power semiconductor elements 16U to 17W is calculated.
  • the loss calculation unit 31 includes the switching loss of the semiconductor switching element 18 constituting each of the power semiconductor elements 16U to 17W, the steady loss (conduction loss or conduction loss), the switching loss of the freewheeling diode 19, and The steady loss (conduction loss or conduction loss) is calculated separately.
  • the switching loss and steady loss (conduction loss or conduction loss) of the semiconductor switching element 18 are the loss of the semiconductor switching element 18 and the amount of heat generated by the semiconductor switching element 18. Further, the switching loss and steady loss (conduction loss or conduction loss) of the freewheeling diode 19 are the losses of the freewheeling diode 19 and the amount of heat generated by the freewheeling diode 19. These are losses of the power semiconductor elements 16U to 17W.
  • the loss of each power semiconductor element 16U to 17W calculated by the loss calculation unit 31 is input to the junction temperature estimation calculation unit 32.
  • the junction temperature estimation calculation unit 32 increases the temperature obtained from the loss of each power semiconductor element 16U to 17W calculated by the loss calculation unit 31 to the temperature Tth in the vicinity of the power semiconductor elements 16U to 17W detected by the temperature sensor 22. By adding the value ⁇ T, the estimated values of the junction temperature Tji of the semiconductor switching element 18 of each power semiconductor element 16U to 17W and the junction temperature Tjd of the freewheeling diode 19 are calculated. In this case, the junction temperature estimation calculation unit 32 multiplies the loss of each power semiconductor element 16U to 17W calculated by the loss calculation unit 31 by the thermal resistance value (heat transfer coefficient) Tr of the power semiconductor elements 16U to 17W (multiplication). ) To calculate the temperature rise value ⁇ T.
  • the estimation calculation in the junction temperature estimation calculation unit 32 is expressed by the following equation (1).
  • Tji, Tjd Tth + ⁇ T (1)
  • ⁇ T loss ⁇ Tr o The calculated junction temperatures Tji and Tjd are input to the temperature protection unit 33.
  • the temperature protection unit 33 performs a predetermined protection operation based on the junction temperature Tji of the semiconductor switching element 18 of each power semiconductor element 16U to 17W and the junction temperature Tjd of the free wheel diode 19 estimated by the junction temperature estimation calculation unit 32. Execute. This protection operation is divided into two stages in the embodiment. First, the highest one of the junction temperatures Tji, Tjd of any one of the power semiconductor elements 16U to 17W exceeds the first predetermined value TS1.
  • the temperature protection unit 33 outputs a current limit signal to the motor control unit 26.
  • the motor control unit 26 adjusts the modulation wave so as to limit the current flowing through the inverter circuit 8 to a predetermined value.
  • the temperature protection unit 33 sends a current interruption signal to the motor control unit 26 when the highest one of the junction temperatures Tji, Tjd exceeds a second predetermined value TS2 higher than the first predetermined value TS1. Output.
  • the motor control unit 26 When the motor control unit 26 receives the current cut-off signal from the temperature protection unit 33, the motor control unit 26 stops the output of the modulated wave, thereby turning off the semiconductor switching elements 18 of all the power semiconductor elements 16 U to 17 W and causing the inverter circuit 8 to Cut off the flowing current.
  • the first predetermined value TS1 and the second predetermined value TS2 are values set from the temperature limits of the semiconductor switching element 18 and the free wheeling diode 19 constituting the power semiconductor elements 16U to 17W.
  • the U-phase, V-phase, and W-phase duties output from the PWM control unit 27 are shown.
  • the phase current waveform shown below the U-phase phase current Iu The fine broken line is the waveform of the V-phase phase current Iv, and the rough broken line is the waveform of the W-phase phase current Iw.
  • the IGBT loss shown below is the loss of the semiconductor switching element 18 calculated by the loss calculation unit 31.
  • the solid line is the loss of the U-phase power semiconductor element 16U, the semiconductor switching element 18 of the 17U, and the fine broken line is the V-phase.
  • the loss of the semiconductor switching element 18 of the power semiconductor elements 16V and 17V and the rough broken line are the loss of the semiconductor switching element 18 of the W-phase power semiconductor elements 16W and 17W.
  • the diode loss shown below is the loss of the freewheeling diode 19 calculated by the loss calculating unit 31.
  • the solid line is the loss of the U-phase power semiconductor element 16U, the freewheeling diode 19 of 17U, and the fine broken line is the V-phase power.
  • the loss of the free-wheeling diode 19 of the semiconductor elements 16V and 17V and the rough broken line are the loss of the free-wheeling diode 19 of the W-phase power semiconductor elements 16W and 17W.
  • the temperature sensor temperature below it is a temperature Tth in the vicinity of the power semiconductor elements 16U to 17W detected by the temperature sensor 22.
  • the estimated IGBT junction temperature shown below is the junction temperature Tji of the semiconductor switching element 18 calculated by the junction temperature estimation calculation unit 32.
  • the solid line shows the semiconductor switching elements of the U-phase power semiconductor elements 16U and 17U. 18 is a junction temperature Tji, a fine broken line is the junction temperature Tji of the V-phase power semiconductor element 16V and 17V, and a rough broken line is the junction temperature of the W-phase power semiconductor element 16W and 17W of the semiconductor switching element 18. Tji.
  • a dashed-dotted line is an average value of junction temperature Tji.
  • the estimated diode junction temperature shown below is the junction temperature Tjd of the freewheeling diode 19 calculated by the junction temperature estimation calculating unit 32.
  • the solid line is the junction temperature of the freewheeling diode 19 of the U-phase power semiconductor elements 16U and 17U.
  • Tjd the fine broken line is the junction temperature Tjd of the V-phase power semiconductor elements 16V and 17V
  • the rough broken line is the junction temperature Tjd of the W-phase power semiconductor elements 16W and 17W.
  • a dashed-dotted line is the average value of junction temperature Tjd.
  • FIG. 4 shows the flow of estimation calculation of the junction temperature executed by the inverter control unit 12.
  • the inverter control unit 12 calculates the amount of heat generated by the semiconductor switching element 18 (IGBT) and the freewheeling diode 19 (Diode) for each PWM carrier period. That is, the loss calculation unit 31 calculates each phase (for each PWM carrier cycle obtained from the duty output from the PWM control unit 27 from the HV voltage (applied voltage) during driving and the instantaneous values of the phase currents Iu, Iv, and Iw. The losses of the power semiconductor elements 16U to 17W of the U-phase, V-phase, and W-phase upper and lower phases are calculated (step S1).
  • the loss calculating unit 31 calculates the loss of the semiconductor switching element 18 from the phase current and HV voltage (applied voltage) flowing when the upper phase semiconductor switching element 18 is ON based on the duty of each phase.
  • Switching loss and steady loss are calculated, and for the lower-phase semiconductor switching element 18, the semiconductor switching is performed from the phase current and HV voltage (applied voltage) that flow when the upper-phase semiconductor switching element 18 is OFF.
  • the loss (switching loss and steady loss) of the element 18 is calculated.
  • the loss (switching loss and steady loss) of the upper and lower phase freewheeling diodes 19 the phase current and the HV voltage (applied to the freewheeling diode 19 when the semiconductor switching element 18 which is a composite with the loss is switched off.
  • the loss (switching loss and steady loss) of the freewheeling diode 19 is calculated from the voltage). For example, taking the U phase in FIG.
  • Steady loss the loss of the freewheeling diode 19 that is a composite with it is zero
  • the loss of the freewheeling diode 19 of the lower-phase power semiconductor element 17U is calculated (and The loss of the semiconductor switching element 18 as a composite is zero).
  • the loss calculation unit 31 takes in a phase current (instantaneous value) in a half cycle of the PWM carrier cycle, performs loss calculation in the remaining half cycle, and outputs the loss calculation to the junction temperature estimation calculation unit 32.
  • each loss in FIG. 3 and the junction temperature are delayed by one cycle from the phase current.
  • the junction temperature estimation calculation unit 32 is calculated by the loss calculation unit 31 for each PWM carrier cycle, and the output power loss of the semiconductor switching elements 18 of the power semiconductor elements 16U to 17W and the loss of the freewheeling diode 19 are heated.
  • a temperature rise value ⁇ T caused by each loss is calculated (step S2).
  • the junction temperature estimation calculation unit 32 takes in the temperature Tth in the vicinity of the power semiconductor elements 16U to 17W detected by the temperature sensor 22 (step S3), and at this temperature Tth, the semiconductor switching element 18 and the free wheel diode 19 By adding the temperature increase value ⁇ T calculated from the loss (Equation 1), the estimated values of the junction temperature Tji of the semiconductor switching element 18 and the junction temperature Tjd of the free-wheeling diode 19 of each of the power semiconductor elements 16U to 17W are calculated ( Step S4). Since each junction temperature Tji, Tjd is calculated for every PWM carrier period, those estimated values are instantaneous values.
  • the calculated junction temperatures Tji and Tjd of the power semiconductor elements 16U to 17W are output to the temperature protection unit 33, and the temperature protection unit 33, as described above, outputs the semiconductor switching elements of the power semiconductor elements 16U to 17W.
  • the highest one of the junction temperature Tji of 18 and the junction temperature Tjd of the reflux diode 19 is extracted.
  • the temperature protection unit 33 outputs a current limiting signal to the motor control unit 26, and further, the second predetermined value TS1.
  • a current interruption signal is output to the motor control unit 26.
  • the motor control unit 26 limits or cuts off the current flowing through the inverter circuit 8 to a predetermined value based on the current limit signal or the current cut-off signal from the temperature protection unit 33.
  • the inverter control unit 12 calculates the loss of the power semiconductor elements 16U to 17W from the phase currents Iu, Iv, Iw detected by the shunt resistor 23 and the HV voltage (applied voltage).
  • the temperature increase value ⁇ T obtained from the loss of the power semiconductor elements 16U to 17W calculated by the loss calculation unit 31 is added to the temperature Tth detected by the calculation unit 31 and the temperature sensor 22, and the power semiconductor elements 16U to 17W
  • the calculation unit 32 estimates the junction temperature of the power semiconductor elements 16U to 17W. Since the carrier period is sufficiently smaller than the frequency of the phase current of the inverter circuit 8 and sufficiently shorter than the thermal time constant of the power semiconductor elements 16U to 17W, the power semiconductor element 16U has a high-speed period such as every PWM carrier period.
  • the instantaneous value of the junction temperature of ⁇ 17W can be estimated.
  • the junction temperature (instantaneous value) of the power semiconductor elements 16U to 17W for each PWM carrier period estimated by the junction temperature estimation calculation unit 32 exceeds a predetermined value (first predetermined value, second predetermined value).
  • a predetermined value first predetermined value, second predetermined value.
  • the loss calculating unit 31 calculates the loss of the power semiconductor elements 16U to 17W from the switching loss and the steady loss of the power semiconductor elements 16U to 17W, and the junction temperature estimation calculating unit 32 is calculated by the loss calculating unit 31.
  • the temperature rise value ⁇ T is calculated by multiplying the loss of the power semiconductor elements 16U to 17W by the thermal resistance value Tr of the power semiconductor elements, the instantaneous value of the junction temperature of the power semiconductor elements 16U to 17W is accurately calculated. And can be estimated.
  • the loss calculation unit 31 uses the phase currents Iu, Iv, Iw of each phase (U phase, V phase, W phase) and the HV voltage (applied voltage) of the inverter circuit 8 to power semiconductor elements 16U for each phase in a bridge configuration.
  • the temperature protection unit 33 performs a protection operation based on the junction temperature of the highest power semiconductor element 16U to 17W, so that the power semiconductor element 16U to 17W of the phase where the temperature rises most rapidly is calculated.
  • the loss calculation unit 31 calculates the loss of the semiconductor switching element 18 and the loss of the free wheel diode 19. Since the junction temperature estimation calculation unit 32 estimates the junction temperature Tji of the semiconductor switching element 18 and the junction temperature Tjd of the freewheeling diode 19, in the case of the power semiconductor elements 16U to 17W having the freewheeling diode 18, Can be protected without any problem. Further, when the junction temperature of the power semiconductor elements 16U to 17W exceeds the first predetermined value, the temperature protection unit 33 limits the current flowing through the inverter circuit 8, and the second temperature is higher than the first predetermined value.
  • the current flowing through the inverter circuit 8 is cut off when the predetermined value is exceeded, it is possible to avoid unnecessary current interruption while reliably protecting the power semiconductor elements 16U to 17W. It becomes.
  • the electric compressor 1 for vehicles like the Example used in a high temperature environment by operating the motor 3 by the inverter apparatus 7 of this invention, it can implement
  • the junction temperatures of the power semiconductor elements 16U to 17W of the U phase, V phase, and W phase are estimated and protected.
  • the invention other than claim 4 is not limited thereto. You may make it protect by estimating the junction temperature of the power semiconductor element only for any one phase or only two phases.
  • the power semiconductor elements 16U to 17W made of the composite of the semiconductor switching element 18 (IGBT, MOSFET) and the freewheeling diode 19 have been described as examples, but the invention other than claim 5 is not limited thereto, The present invention is also effective for an inverter circuit having only a semiconductor switching element (IGBT, MOSFET) having no freewheeling diode. Furthermore, in the embodiments, the present invention has been described with an inverter device that drives a motor of an electric compressor mounted on a vehicle. However, the invention is not limited to the invention other than claim 7, and an inverter circuit having a power semiconductor element in a bridge configuration. The present invention is effective for all inverter devices using the.

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Abstract

Provided is an inverter apparatus that is capable of protecting power semiconductor elements with high precision even from instantaneous temperature increase. An inverter control unit 12 calculates losses of power semiconductor elements 16U to 17W by a loss calculation unit 31 for each PWM carrier period in a PWM control unit 27, and estimates a junction temperature of the power semiconductor elements by a junction temperature estimation calculation unit 32. A temperature protection unit 33 executes a prescribed protection operation when the junction temperature of the power semiconductor elements estimated by the junction temperature estimation calculation unit for each PWM carrier period exceeds a prescribed value.

Description

インバータ装置及びそれを備えた車両用電動圧縮機INVERTER DEVICE AND VEHICLE ELECTRIC COMPRESSOR HAVING THE SAME
　本発明は、例えば電動圧縮機のモータを運転するインバータ装置、及び、それを備えた車両用電動圧縮機に関するものである。 The present invention relates to an inverter device that operates a motor of an electric compressor, for example, and a vehicle electric compressor including the same.
　近年の環境問題からハイブリッド自動車や電気自動車が注目されてきているが、この種ハイブリッド自動車等の空調装置では冷媒圧縮機として電動圧縮機が使用される。この電動圧縮機は、車両のバッテリから給電されるモータにより圧縮要素を駆動するものであるが、このモータはインバータ装置により運転される。
　この種のインバータ装置は、ブリッジ構成の電力用半導体素子（ＩＧＢＴやＭＯＳＦＥＴ等）をスイッチングすることにより、モータの各相の通電を制御するものであるが、これら電力用半導体素子は、損失分の発熱を生じるため、特に電動圧縮機がエンジンルームなどの過酷な温度（高温）環境下で使用される車両用電動圧縮機である場合には、インバータ装置を構成する電力用半導体素子の過熱保護が極めて重要となる。
　係る電力用半導体素子の発熱を考慮した保護方式として、電力用半導体素子のジャンクション温度を推定し、当該ジャンクション温度が所定値に上昇したことに基づいて運転を停止するものがある。このジャンクション温度とは、電力用半導体素子内部のチップの温度（ＩＧＢＴチップやＭＯＳＦＥＴチップと、ＦＷＤチップの表面温度）であり、電力用半導体素子が実装された基板の温度（電力用半導体素子の近傍の温度）を温度センサ（温度検出器）で検出し、この検出値に電力用半導体素子のスイッチング損失と定常損失（導通損失又は通電損失）から成る損失で生じる発熱量に相当する温度上昇値を加えることで得られるものである（例えば、特許文献１参照）。 In recent years, hybrid vehicles and electric vehicles have been attracting attention due to environmental problems. In such air conditioners such as hybrid vehicles, an electric compressor is used as a refrigerant compressor. This electric compressor drives a compression element by a motor fed from a vehicle battery, and this motor is operated by an inverter device.
This type of inverter device controls the energization of each phase of the motor by switching a power semiconductor element (IGBT, MOSFET, etc.) having a bridge configuration. In order to generate heat, especially when the electric compressor is an electric compressor for a vehicle used in a severe temperature (high temperature) environment such as an engine room, the overheat protection of the power semiconductor element constituting the inverter device is prevented. It becomes extremely important.
As a protection method in consideration of heat generation of the power semiconductor element, there is a method of estimating the junction temperature of the power semiconductor element and stopping the operation based on the junction temperature rising to a predetermined value. This junction temperature is the temperature of the chip inside the power semiconductor element (surface temperature of the IGBT chip, MOSFET chip, and FWD chip), and the temperature of the substrate on which the power semiconductor element is mounted (near the power semiconductor element). The temperature rise value corresponding to the amount of heat generated by the loss consisting of the switching loss and steady loss (conduction loss or conduction loss) of the power semiconductor element is detected by the temperature sensor (temperature detector). It is obtained by adding (for example, refer patent document 1).
特許第３０７５３０３号公報Japanese Patent No. 3075303
　図５は係るジャンクション温度による従来の保護方式を説明する図である。この図において縦軸はブリッジ構成の電力用半導体素子から成るインバータ回路の相電流値の値、横軸は電力用半導体素子が実装された基板の温度（電力用半導体素子の近傍の温度）を検出する温度センサの検出値である。
　図５中に示された折れ線は、温度センサの検出値がＴ１に上昇するまでは、相電流が所定値Ａｓｔｏｐに達したときに電流を遮断し、温度センサの検出値がＴ１からＴ２に上昇するまでは、所定値Ａｓｔｏｐより小さい値（図５の折れ線中の斜め線）で遮断を行う保護閾値を示しており、この保護閾値の計算方法は以下の通りである。
　即ち、車両のバッテリ（車両用ＨＶ電源）から印加される電圧（ＨＶ電圧）が最大値（例えば、３００Ｖなど）であると想定して、電力用半導体素子の特性に基づき、印加電圧（ＨＶ電圧）と相電流の値から当該電力用半導体素子の損失（発熱量）を計算し、この損失から温度上昇値を予め計算しておく（相電流と温度上昇値との関係を、印加電圧が最大値として予め求めておく）。そして、この温度上昇値に温度センサの検出値を加えたジャンクション温度が、電力用半導体素子の温度限界である所定値（例えば、＋１７５℃等）となる相電流と温度センサの検出値との関係を示したのが図５中の保護閾値（折れ線）である。
　そして、従来では図５に示した保護閾値に基づき、例えばそのときの温度センサの検出値がＴ１になるまでは、相電流の値がＡｓｔｏｐまで上昇した時点で電流を遮断すると云う電力用半導体素子の保護を行っていた。しかしながら、この保護閾値は印加電圧（ＨＶ電圧）が最大値のとき、即ち、最悪条件での保護閾値であるため、印加電圧（ＨＶ電圧）が低いときには、保護を行う必要が無い段階で電流を遮断してしまうという欠点があった。
　また、最悪条件での損失計算についても、従来では相電流を正弦波と仮定して比較的長い周期の平均値から損失を計算していた。しかしながら、印加電圧には変調を掛けるため、相電流の波形も理想的な正弦波とはならず、リプルや高調波成分を含んだものとなる。そのため、平均値では分からなくても、瞬時的には電力用半導体素子の温度限界を超える状況も生じており、従来ではこれを的確に判定して保護することができていなかった。
　本発明は、係る従来の技術的課題を解決するために成されたものであり、瞬時的な温度上昇からも電力用半導体素子を高精度で保護することができるインバータ装置及びそれを用いた車両用電動圧縮機を提供することを目的とする。 FIG. 5 is a diagram for explaining a conventional protection method based on the junction temperature. In this figure, the vertical axis indicates the phase current value of the inverter circuit composed of power semiconductor elements in a bridge configuration, and the horizontal axis detects the temperature of the substrate on which the power semiconductor elements are mounted (the temperature in the vicinity of the power semiconductor elements). The detected value of the temperature sensor.
The broken line shown in FIG. 5 indicates that the current is cut off when the phase current reaches the predetermined value Asstop until the detection value of the temperature sensor rises to T1, and the detection value of the temperature sensor rises from T1 to T2. Until this time, the protection threshold value for performing the blocking with a value smaller than the predetermined value Asstop (the oblique line in the broken line in FIG. 5) is shown, and the calculation method of this protection threshold value is as follows.
That is, assuming that the voltage (HV voltage) applied from the vehicle battery (HV power supply for the vehicle) is a maximum value (for example, 300 V), the applied voltage (HV voltage) is based on the characteristics of the power semiconductor element. ) And the phase current value, the loss (heat generation amount) of the power semiconductor element is calculated, and the temperature rise value is calculated in advance from this loss (the relationship between the phase current and the temperature rise value is determined by the maximum applied voltage). It is obtained in advance as a value). The relationship between the phase current at which the junction temperature obtained by adding the detected value of the temperature sensor to the temperature rise value becomes a predetermined value (for example, + 175 ° C.) which is the temperature limit of the power semiconductor element, and the detected value of the temperature sensor The protection threshold value (broken line) in FIG. 5 is shown.
Conventionally, based on the protection threshold value shown in FIG. 5, for example, until the detected value of the temperature sensor at that time reaches T1, the power semiconductor element that cuts off the current when the phase current value rises to Asstop. Had done protection. However, since this protection threshold is the maximum value of the applied voltage (HV voltage), that is, the worst-case protection threshold, when the applied voltage (HV voltage) is low, the current is applied at a stage where it is not necessary to perform protection. There was a drawback of blocking.
Further, regarding the loss calculation under the worst condition, conventionally, the phase current is assumed to be a sine wave, and the loss is calculated from an average value of a relatively long period. However, since the applied voltage is modulated, the waveform of the phase current is not an ideal sine wave, but includes ripples and harmonic components. For this reason, even if the average value is not known, there is an instantaneous situation where the temperature limit of the power semiconductor element is exceeded. In the past, this could not be accurately determined and protected.
The present invention has been made to solve the conventional technical problems, and an inverter device capable of protecting a power semiconductor element with high accuracy from an instantaneous temperature rise and a vehicle using the same An object is to provide an electric compressor.
　本発明のインバータ装置は、ブリッジ構成を成す電力用半導体素子を有するインバータ回路と、電力用半導体素子を駆動するＰＷＭ制御部を有するインバータ制御部を備えたものであって、電力用半導体素子近傍の温度を検出する温度検出器と、インバータ回路の相電流を検出する相電流検出器を備え、インバータ制御部は、相電流検出器が検出する少なくとも一相の相電流と印加電圧から電力用半導体素子の損失を計算する損失計算部と、温度検出器が検出する温度に、損失計算部が計算した電力用半導体素子の損失から得られる温度上昇値を加えて、当該電力用半導体素子のジャンクション温度を推定するジャンクション温度推定計算部と、を有し、ＰＷＭ制御部におけるＰＷＭキャリア周期毎に、損失計算部により電力用半導体素子の損失を計算し、ジャンクション温度推定計算部により電力用半導体素子のジャンクション温度を推定すると共に、このジャンクション温度推定計算部が推定したＰＷＭキャリヤ周期毎の電力用半導体素子のジャンクション温度が所定値を超えた場合、所定の保護動作を実行することを特徴とする。
　請求項２の発明のインバータ装置は、上記発明においてＰＷＭキャリア周期はインバータ回路の相電流の周波数よりも十分に小さく、電力用半導体素子の熱時定数よりも十分短いことを特徴とする。
　請求項３の発明のインバータ装置は、上記各発明において損失計算部は、電力用半導体素子のスイッチング損失と定常損失から当該電力用半導体素子の損失を計算し、ジャンクション温度推定計算部は、損失計算部が計算した電力用半導体素子の損失に当該電力用半導体素子の熱抵抗値を掛けることで温度上昇値を算出することを特徴とする。
　請求項４の発明のインバータ装置は、上記各発明において損失計算部は、インバータ回路の各相の相電流と印加電圧からブリッジ構成の各相の電力用半導体素子の損失を計算すると共に、インバータ制御部は、最も高い電力用半導体素子のジャンクション温度に基づいて保護動作を実行することを特徴とする。
　請求項５の発明のインバータ装置は、上記各発明において電力用半導体素子は、半導体スイッチング素子と還流ダイオードとの複合体であり、損失計算部は、半導体スイッチング素子の損失と還流ダイオードの損失を計算し、ジャンクション温度推定計算部は、半導体スイッチング素子のジャンクション温度と還流ダイオードのジャンクション温度を推定することを特徴とする。
　請求項６の発明のインバータ装置は、上記各発明においてインバータ制御部は、電力用半導体素子のジャンクション温度が第１の所定値を超えた場合、インバータ回路に流れる電流を制限すると共に、ジャンクション温度が第１の所定値より高い第２の所定値を超えた場合、インバータ回路に流す電流を遮断することを特徴とする。
　請求項７の発明の車両用電動圧縮機は、上記各発明のインバータ装置により運転されるモータを備えて車両に搭載されることを特徴とする。 An inverter device of the present invention includes an inverter circuit having a power semiconductor element having a bridge configuration and an inverter control unit having a PWM control unit for driving the power semiconductor element, and is provided in the vicinity of the power semiconductor element. A temperature detector for detecting the temperature and a phase current detector for detecting the phase current of the inverter circuit are provided. The inverter control unit is a power semiconductor element based on at least one phase current detected by the phase current detector and the applied voltage. Add the temperature rise value obtained from the loss of the power semiconductor element calculated by the loss calculation section to the temperature detected by the loss detector and the temperature detector that calculates the loss of the power, and calculate the junction temperature of the power semiconductor element. A junction temperature estimation calculation unit for estimation, and for each PWM carrier period in the PWM control unit, the loss calculation unit of the power semiconductor element The junction temperature of the power semiconductor element is estimated by the junction temperature estimation calculation unit, and the junction temperature of the power semiconductor element for each PWM carrier period estimated by the junction temperature estimation calculation unit exceeds a predetermined value. In this case, a predetermined protection operation is performed.
The inverter device according to a second aspect of the invention is characterized in that, in the above invention, the PWM carrier cycle is sufficiently smaller than the frequency of the phase current of the inverter circuit and sufficiently shorter than the thermal time constant of the power semiconductor element.
In the inverter device according to a third aspect of the invention, in each of the above inventions, the loss calculation unit calculates the loss of the power semiconductor element from the switching loss and the steady loss of the power semiconductor element, and the junction temperature estimation calculation unit calculates the loss. The temperature rise value is calculated by multiplying the loss of the power semiconductor element calculated by the unit by the thermal resistance value of the power semiconductor element.
In the inverter device according to a fourth aspect of the present invention, in each of the above inventions, the loss calculation unit calculates the loss of the power semiconductor element of each phase of the bridge configuration from the phase current and applied voltage of each phase of the inverter circuit, and performs inverter control. The unit performs a protection operation based on a junction temperature of the highest power semiconductor element.
According to a fifth aspect of the present invention, there is provided an inverter device according to any of the above-mentioned inventions, wherein the power semiconductor element is a composite of a semiconductor switching element and a free wheel diode, and the loss calculation unit calculates the loss of the semiconductor switching element and the free wheel diode The junction temperature estimation calculation unit estimates the junction temperature of the semiconductor switching element and the junction temperature of the free-wheeling diode.
In the inverter device of the invention of claim 6, in each of the above inventions, the inverter control unit limits the current flowing through the inverter circuit when the junction temperature of the power semiconductor element exceeds the first predetermined value, and the junction temperature is When the second predetermined value higher than the first predetermined value is exceeded, the current flowing through the inverter circuit is cut off.
According to a seventh aspect of the present invention, there is provided a vehicle electric compressor including a motor that is operated by the inverter device according to each of the above inventions, and is mounted on the vehicle.
　本発明によれば、ブリッジ構成を成す電力用半導体素子を有するインバータ回路と、電力用半導体素子を駆動するＰＷＭ制御部を有するインバータ制御部を備えたインバータ装置において、電力用半導体素子近傍の温度を検出する温度検出器と、インバータ回路の相電流を検出する相電流検出器を備えており、インバータ制御部が、相電流検出器が検出する少なくとも一相の相電流と印加電圧から電力用半導体素子の損失を計算する損失計算部と、温度検出器が検出する温度に、損失計算部が計算した電力用半導体素子の損失から得られる温度上昇値を加えて、当該電力用半導体素子のジャンクション温度を推定するジャンクション温度推定計算部と、を有して、ＰＷＭ制御部におけるＰＷＭキャリア周期毎に、損失計算部により電力用半導体素子の損失を計算し、ジャンクション温度推定計算部により電力用半導体素子のジャンクション温度を推定するようにした。このＰＷＭキャリア周期は、請求項２の発明の如くインバータ回路の相電流の周波数よりも十分に小さく、電力用半導体素子の熱時定数よりも十分短いので、係るＰＷＭキャリア周期毎といった高速の周期で電力用半導体素子のジャンクション温度の瞬時値を推定することができるようになる。
　そして、このジャンクション温度推定計算部が推定したＰＷＭキャリヤ周期毎の電力用半導体素子のジャンクション温度（瞬時値）が所定値を超えた場合、所定の保護動作を実行するようにしたので、瞬時的な温度上昇からも電力用半導体素子を高精度で保護することができるようになる。
　この場合、請求項３の発明の如く損失計算部が、電力用半導体素子のスイッチング損失と定常損失から当該電力用半導体素子の損失を計算し、ジャンクション温度推定計算部が、損失計算部が計算した電力用半導体素子の損失に当該電力用半導体素子の熱抵抗値を掛けることで温度上昇値を算出するようにすれば、電力用半導体素子のジャンクション温度の瞬時値を的確に計算して推定することができるようになる。
　また、請求項４の発明の如く損失計算部が、インバータ回路の各相の相電流と印加電圧からブリッジ構成の各相の電力用半導体素子の損失を計算し、インバータ制御部が、最も高い電力用半導体素子のジャンクション温度に基づいて保護動作を実行するようにすれば、最も温度上昇が激しい相の電力用半導体素子を安全、且つ、正確に保護することができるようになる。
　特に、請求項５の発明の如く電力用半導体素子が、半導体スイッチング素子と還流ダイオードとの複合体である場合に、損失計算部が、半導体スイッチング素子の損失と還流ダイオードの損失を計算し、ジャンクション温度推定計算部は、半導体スイッチング素子のジャンクション温度と還流ダイオードのジャンクション温度を推定するようにすれば、還流ダイオードを有する電力用半導体素子の場合にも支障無く保護を行うことができるようになる。
　更に、請求項６の発明の如くインバータ制御部が、電力用半導体素子のジャンクション温度が第１の所定値を超えた場合、インバータ回路に流れる電流を制限し、ジャンクション温度が第１の所定値より高い第２の所定値を超えた場合、インバータ回路に流す電流を遮断するようにすることで、電力用半導体素子の保護を確実に行いながら、不必要な電流遮断の発生を回避することが可能となる。
　そして、高温環境下で使用される請求項７の発明の如き車両用電動圧縮機において、上記各発明のインバータ装置によりモータを運転することで、極めて効果的な過熱保護を実現することができるようになるものである。 According to the present invention, in an inverter device including an inverter circuit having a power semiconductor element having a bridge configuration and an inverter control unit having a PWM control unit for driving the power semiconductor element, the temperature in the vicinity of the power semiconductor element is increased. A temperature detector to detect and a phase current detector to detect the phase current of the inverter circuit, and the inverter control unit detects the power semiconductor element from at least one phase current detected by the phase current detector and the applied voltage. Add the temperature rise value obtained from the loss of the power semiconductor element calculated by the loss calculation section to the temperature detected by the loss detector and the temperature detector that calculates the loss of the power, and calculate the junction temperature of the power semiconductor element. A junction temperature estimation calculation unit for estimation, and for each PWM carrier period in the PWM control unit, the loss calculation unit performs power semiconductor The loss of the element was calculated and to estimate the junction temperature of the power semiconductor device by a junction temperature estimation calculation unit. Since the PWM carrier period is sufficiently smaller than the phase current frequency of the inverter circuit and sufficiently shorter than the thermal time constant of the power semiconductor element as in the second aspect of the invention, the PWM carrier period is a high-speed period such as every PWM carrier period. An instantaneous value of the junction temperature of the power semiconductor element can be estimated.
Then, when the junction temperature (instantaneous value) of the power semiconductor element for each PWM carrier period estimated by the junction temperature estimation calculation unit exceeds a predetermined value, a predetermined protection operation is executed. The power semiconductor element can be protected with high accuracy even from a temperature rise.
In this case, the loss calculation section calculates the loss of the power semiconductor element from the switching loss and the steady loss of the power semiconductor element as in the invention of claim 3, and the junction temperature estimation calculation section calculates the loss calculation section. If the temperature rise value is calculated by multiplying the loss of the power semiconductor element by the thermal resistance value of the power semiconductor element, the instantaneous value of the junction temperature of the power semiconductor element is accurately calculated and estimated. Will be able to.
According to a fourth aspect of the present invention, the loss calculation unit calculates the loss of the power semiconductor element of each phase of the bridge configuration from the phase current and applied voltage of each phase of the inverter circuit, and the inverter control unit calculates the highest power. If the protection operation is executed based on the junction temperature of the semiconductor element for power, the power semiconductor element in the phase where the temperature rises most rapidly can be safely and accurately protected.
In particular, when the power semiconductor element is a composite of a semiconductor switching element and a free-wheeling diode as in the invention of claim 5, the loss calculation unit calculates the loss of the semiconductor switching element and the free-wheeling diode, If the temperature estimation calculation unit estimates the junction temperature of the semiconductor switching element and the junction temperature of the free-wheeling diode, the temperature estimation calculation unit can protect the power semiconductor element having the free-wheeling diode without any trouble.
Further, as in the sixth aspect of the invention, the inverter control unit limits the current flowing through the inverter circuit when the junction temperature of the power semiconductor element exceeds the first predetermined value, and the junction temperature is less than the first predetermined value. When the high second predetermined value is exceeded, by interrupting the current flowing through the inverter circuit, it is possible to avoid unnecessary current interruption while reliably protecting the power semiconductor element. It becomes.
And in the electric compressor for vehicles like the invention of Claim 7 used in a high temperature environment, a very effective overheat protection can be implement | achieved by operating a motor with the inverter apparatus of said each invention. It will be.
本発明のインバータ装置を適用した一実施例の車両用電動圧縮機の概略断面図である。It is a schematic sectional drawing of the electric compressor for vehicles of one Example to which the inverter apparatus of this invention is applied. 本発明の一実施例のインバータ装置の電気回路図である。It is an electric circuit diagram of the inverter apparatus of one Example of this invention. 図２のインバータ装置のスイッチング制御と半導体スイッチング素子及び還流ダイオードの損失、ジャンクション温度の推定値を示す図である。It is a figure which shows the switching control of the inverter apparatus of FIG. 2, the loss of a semiconductor switching element and a free-wheeling diode, and the estimated value of junction temperature. 図２のインバータ制御部が実行するジャンクション温度の推定計算の流れを示す図である。It is a figure which shows the flow of estimation calculation of the junction temperature which the inverter control part of FIG. 2 performs. ジャンクション温度による従来の保護方式を説明する図である。It is a figure explaining the conventional protection system by junction temperature.
　以下、本発明の実施の形態について、詳細に説明する。図１は本発明を適用した車両用電動圧縮機１の概略断面図を示している。実施例の電動圧縮機１は、図示しない車両の車室内を空調する空気調和装置の冷媒回路の一部を構成するもので、車両のエンジンルームに搭載される。電動圧縮機１は、ハウジング２内にモータ３と、このモータ３の回転軸４により駆動されるスクロール型等の圧縮要素６を備えている。ハウジング２には更に本発明のインバータ装置７が取り付けられており、このインバータ装置７によりモータ３は運転され、圧縮要素６を駆動する。圧縮要素６はモータ３の回転軸４により駆動されて冷媒回路から冷媒を吸い込み、圧縮して再度冷媒回路に吐出するものである。
　次に、図２は係るインバータ装置７の電気回路図を示している。実施例のインバータ装置７は、インバータ回路８及び平滑コンデンサ９が実装された制御基板１１と、マイクロコンピュータ（プロセッサ）により構成されたインバータ制御部１２を備えている。インバータ回路８の正側の直流母線１３は、図示しない車両のバッテリ（車両用ＨＶ電源）Ｂの＋端子に接続され、負側の直流母線１４は、バッテリＢの−端子に接続されている。そして、平滑コンデンサ９はインバータ回路８の二つの直流母線１３、１４間に接続されている。
　インバータ回路８は、ブリッジを構成する複数の電力用半導体素子のスイッチング状態をそれぞれ変化させて、バッテリＢから印加される直流を交流に変換し、モータ３に供給するものである。具体的には、ブリッジの上相を構成する三つの電力用半導体素子１６Ｕ、１６Ｖ、１６Ｗと、ブリッジの下相を構成する三つの電力用半導体素子１７Ｕ、１７Ｖ、１７Ｗを備えている。各電力用半導体素子１６Ｕ、１６Ｖ、１６Ｗ、及び、１７Ｕ、１７Ｖ、１７Ｗは、何れも半導体スイッチング素子１８とそれに逆並列で接続された還流ダイオード１９との複合体であり、このインバータ回路８の直流母線１３、１４にはバッテリＢより直流電源が供給される。
　このインバータ回路８では、上相の電力用半導体素子１６Ｕ、１６Ｖ、１６Ｗの各半導体スイッチング素子１８と下相の電力用半導体素子１７Ｕ、１７Ｖ、１７Ｗの各半導体スイッチング素子１８とが、１対１に対応して直列接続されている。以下では、直列接続された電力用半導体素子１６Ｕ~１７Ｗの各半導体スイッチング素子１８の対をスイッチングレグと称する。即ち、実施例では電力用半導体素子１６Ｕの半導体スイッチング素子１８と電力用半導体素子１７Ｕの半導体スイッチング素子１８の対で構成されたスイッチングレグ２１Ｕと、電力用半導体素子１６Ｖの半導体スイッチング素子１８と電力用半導体素子１７Ｖの半導体スイッチング素子１８の対で構成されたスイッチングレグ２１Ｖと、電力用半導体素子１６Ｗの半導体スイッチング素子１８と電力用半導体素子１７Ｗの半導体スイッチング素子１８の対で構成されたスイッチングレグ２１Ｗとがある。
　これらスイッチングレグ２１Ｕ、２１Ｖ、２１Ｗは、正側の直流母線１３と負側の直流母線１４との間にそれぞれ接続されている。また、それぞれのスイッチングレグ２１Ｕ、２１Ｖ、２１Ｗの各中間点ＭＵ、ＭＶ、ＭＷが出力交流の各相（Ｕ相、Ｖ相、Ｗ相）の相電圧Ｖｕ、Ｖｖ、Ｖｗを出力するノードであり、各中間点ＭＵ、ＭＶ、ＭＷがモータ３の各相に接続されている。
　実施例のインバータ回路８では、半導体スイッチング素子１８はＩＧＢＴ（Ｉｎｓｕｌａｔｅｄ　Ｇａｔｅ　Ｂｉｐｏｌａｒ　Ｔｒａｎｓｉｓｔｏｒ）を用いている。尚、半導体スイッチング素子１８としては係るＩＧＢＴに限らず、ＭＯＳＦＥＴ等でも良い。また、制御基板１１には電力用半導体素子１６Ｕ~１７Ｗの近傍に位置して温度検出器としての温度センサ２２が実装されている。この温度センサ２２は実施例ではサーミスタから構成されている。
　更に、モータ３からの電流が流れ込む位置の負側の直流母線１４には相電流検出器としてのシャント抵抗２３が接続されている。このシャント抵抗２３にモータ３からの電流が流れると、シャント抵抗２３の両端には電位差が生じ、この両端間の電圧を検出することで、相電流Ｉｕ、Ｉｖ、Ｉｗを算出することができる。尚、相電流検出器としては係るシャント抵抗に限らず、カレントトランス等で構成しても良い。
　一方、インバータ制御部１２は、モータ制御部２６と、ＰＷＭ制御部２７と、電流検出部２８と、ゲートドライバ２９と、損失計算部３１と、ジャンクション温度推定計算部３２と、温度保護部３３を備えている。そして、正側の直流母線１３のＨＶ電圧（印加電圧）はＰＷＭ制御部２７と損失計算部３１に入力される。
　モータ制御部２６はモータ３に印加する三相正弦波の目標とする波形（変調波）をＰＷＭ制御部２７に出力する。ＰＷＭ制御部２７はモータ制御部２６が出力する変調波とキャリア（三角波）の高低を比較することにより、ドライブ信号であるデューティ（Ｄｕｔｙ：上相ＯＮ時間）を生成する。このデューティはＵ相、Ｖ相、Ｗ相の各相について生成され、各半導体スイッチング素子１８のゲートをドライブ（ＯＮ−ＯＦＦ）するゲートドライバ２９に送出される。
　尚、実施例のモータ３の回転速度である相電流Ｉｕ、Ｉｖ、Ｉｗの周波数は４００Ｈｚ~５００Ｈｚであり、ＰＷＭ制御部２７におけるキャリアの周期（以下、ＰＷＭキャリア周期と称する）は、それよりも十分小さい（或いは、十分短い）２０ｋＨｚである。また、電力用半導体素子１６Ｕ~１７Ｗの熱時定数（損失分の温度上昇値として温度センサ２２に伝達するまでにかかる時間）は５０ｍｓｅｃ程であり、ＰＷＭキャリア周期はこの熱時定数よりも十分短い（或いは、十分早い）。
　電流検出部２８は、シャント抵抗２３の両端間の電圧を入力し、当該シャント抵抗２３の抵抗値から相電流Ｉｕ、Ｉｖ、Ｉｗを算出する。算出された相電流Ｉｕ、Ｉｖ、Ｉｗは損失計算部３１に入力される。
　損失計算部３１は、電流検出部２８から入力されたＵ相、Ｖ相、Ｗ相の各相の相電流Ｉｕ、Ｉｖ、Ｉｗ、及び、正側の直流母線１３のＨＶ電圧（印加電圧）と、ＰＷＭ制御部２７から入力されるデューティに基づき、各電力用半導体素子１６Ｕ~１７Ｗの損失を計算する。実施例の場合、損失計算部３１は各電力用半導体素子１６Ｕ~１７Ｗを構成する半導体スイッチング素子１８のスイッチング損失、及び、定常損失（導通損失又は通電損失）と、還流ダイオード１９のスイッチング損失、及び、定常損失（導通損失又は通電損失）をそれぞれ別個に計算する。
　この半導体スイッチング素子１８のスイッチング損失、及び、定常損失（導通損失又は通電損失）が半導体スイッチング素子１８の損失であり、当該半導体スイッチング素子１８の発熱量となる。また、還流ダイオード１９のスイッチング損失、及び、定常損失（導通損失又は通電損失）が還流ダイオード１９の損失であり、当該還流ダイオード１９の発熱量となる。そして、これらが各電力用半導体素子１６Ｕ~１７Ｗの損失となる。この損失計算部３１で算出された各電力用半導体素子１６Ｕ~１７Ｗの損失は、ジャンクション温度推定計算部３２に入力される。
　ジャンクション温度推定計算部３２は、温度センサ２２が検出する電力用半導体素子１６Ｕ~１７Ｗの近傍の温度Ｔｔｈに、損失計算部３１が計算した各電力用半導体素子１６Ｕ~１７Ｗの損失から得られる温度上昇値ΔＴを加えることで、各電力用半導体素子１６Ｕ~１７Ｗの半導体スイッチング素子１８のジャンクション温度Ｔｊｉと還流ダイオード１９のジャンクション温度Ｔｊｄの推定値を計算する。
　この場合、ジャンクション温度推定計算部３２は、電力用半導体素子１６Ｕ~１７Ｗの熱抵抗値（熱伝達係数）Ｔｒを損失計算部３１が計算した各電力用半導体素子１６Ｕ~１７Ｗの損失に掛ける（乗算）ことで上記温度上昇値ΔＴを算出する。上記のジャンクション温度推定計算部３２における推定計算を式で表すと下記式（１）のようになる。
　Ｔｊｉ、Ｔｊｄ＝Ｔｔｈ＋ΔＴ　・・・（１）
　尚、ΔＴは損失×Ｔｒｏ
　そして、算出された各ジャンクション温度Ｔｊｉ、Ｔｊｄは温度保護部３３に入力される。
　温度保護部３３は、ジャンクション温度推定計算部３２が推定した各電力用半導体素子１６Ｕ~１７Ｗの半導体スイッチング素子１８のジャンクション温度Ｔｊｉ、及び、還流ダイオード１９のジャンクション温度Ｔｊｄに基づいて所定の保護動作を実行する。この保護動作は、実施例では二段階に分かれており、先ず、何れかの電力用半導体素子１６Ｕ~１７Ｗの各ジャンクション温度Ｔｊｉ、Ｔｊｄのうちの最も高いものが第１の所定値ＴＳ１を超えた場合、温度保護部３３はモータ制御部２６に電流制限信号を出力する。
　モータ制御部２６は温度保護部３３から電流制限信号を受信した場合、インバータ回路８に流れる電流を所定の値に制限するように変調波を調整する。また、温度保護部３３は、前記ジャンクション温度Ｔｊｉ、Ｔｊｄのうちの最も高いものが前記第１の所定値ＴＳ１より高い第２の所定値ＴＳ２を超えた場合、電流遮断信号をモータ制御部２６に出力する。モータ制御部２６は温度保護部３３から電流遮断信号を受信した場合、変調波の出力を停止することで、全ての電力用半導体素子１６Ｕ~１７Ｗの半導体スイッチング素子１８をＯＦＦし、インバータ回路８に流れる電流を遮断する。これら第１の所定値ＴＳ１及び第２の所定値ＴＳ２は、電力用半導体素子１６Ｕ~１７Ｗを構成する半導体スイッチング素子１８及び還流ダイオード１９の温度限界から設定された値である。
　次に、図３及び図４を参照しながら、インバータ制御部１２による電力用半導体素子１６Ｕ~１７Ｗの発熱量に応じた具体的な過熱保護動作を説明する。図３の最上段にはＰＷＭ制御部２７におけるデューティの生成に使用されるキャリアの波形が示されており、その周期がＰＷＭキャリア周期である。そして、その下にＰＷＭ制御部２７から出力されるＵ相、Ｖ相、Ｗ相のデューティが示されており、その下に示された相電流の波形は、実線がＵ相の相電流Ｉｕ、細かい破線がＶ相の相電流Ｉｖ、荒い破線がＷ相の相電流Ｉｗの波形である。
　その下に示したＩＧＢＴ損失は損失計算部３１が算出する半導体スイッチング素子１８の損失であり、同じく実線がＵ相の電力用半導体素子１６Ｕ、１７Ｕの半導体スイッチング素子１８の損失、細かい破線がＶ相の電力用半導体素子１６Ｖ、１７Ｖの半導体スイッチング素子１８の損失、荒い破線がＷ相の電力用半導体素子１６Ｗ、１７Ｗの半導体スイッチング素子１８の損失である。
　その下に示したＤｉｏｄｅ損失は損失計算部３１が算出する還流ダイオード１９の損失であり、同じく実線がＵ相の電力用半導体素子１６Ｕ、１７Ｕの還流ダイオード１９の損失、細かい破線がＶ相の電力用半導体素子１６Ｖ、１７Ｖの還流ダイオード１９の損失、荒い破線がＷ相の電力用半導体素子１６Ｗ、１７Ｗの還流ダイオード１９の損失である。また、その下の温度センサ温度は、温度センサ２２が検出する電力用半導体素子１６Ｕ~１７Ｗの近傍の温度Ｔｔｈである。
　そして、その下に示したＩＧＢＴジャンクション温度推定値はジャンクション温度推定計算部３２が計算した半導体スイッチング素子１８のジャンクション温度Ｔｊｉであり、同じく実線がＵ相の電力用半導体素子１６Ｕ、１７Ｕの半導体スイッチング素子１８のジャンクション温度Ｔｊｉ、細かい破線がＶ相の電力用半導体素子１６Ｖ、１７Ｖの半導体スイッチング素子１８のジャンクション温度Ｔｊｉ、荒い破線がＷ相の電力用半導体素子１６Ｗ、１７Ｗの半導体スイッチング素子１８のジャンクション温度Ｔｊｉである。尚、一点鎖線はジャンクション温度Ｔｊｉの平均値である。
　その下に示したＤｉｏｄｅジャンクション温度推定値はジャンクション温度推定計算部３２が計算した還流ダイオード１９のジャンクション温度Ｔｊｄであり、同じく実線がＵ相の電力用半導体素子１６Ｕ、１７Ｕの還流ダイオード１９のジャンクション温度Ｔｊｄ、細かい破線がＶ相の電力用半導体素子１６Ｖ、１７Ｖの還流ダイオード１９のジャンクション温度Ｔｊｄ、荒い破線がＷ相の電力用半導体素子１６Ｗ、１７Ｗの還流ダイオード１９のジャンクション温度Ｔｊｄである。尚、一点鎖線はジャンクション温度Ｔｊｄの平均値である。
　図４はインバータ制御部１２が実行するジャンクション温度の推定計算の流れを示している。本発明では、インバータ制御部１２は半導体スイッチング素子１８（ＩＧＢＴ）と還流ダイオード１９（Ｄｉｏｄｅ）の発熱量を、ＰＷＭキャリア周期毎に計算する。即ち、損失計算部３１はＰＷＭ制御部２７から出力されるデューティから得られるＰＷＭキャリア周期毎に、駆動中のＨＶ電圧（印加電圧）と相電流Ｉｕ、Ｉｖ、Ｉｗの瞬時値から、各相（Ｕ相、Ｖ相、Ｗ相の上下相）の電力用半導体素子１６Ｕ~１７Ｗの損失を計算する（ステップＳ１）。
　ここで、各電力用半導体素子１６Ｕ~１７Ｗにおいて、電流は必ず半導体スイッチング素子１８か還流ダイオード１９の何れかにしか流れないため、半導体スイッチング素子１８に電流が流れている期間は還流ダイオード１９の損失は零、逆に還流ダイオード１９に電流が流れている期間は半導体スイッチング素子１８の損失は零である。
　そこで、前述した如く損失計算部３１は各相のデューティに基づいて上相の半導体スイッチング素子１８がＯＮしているときにそれに流れる相電流とＨＶ電圧（印加電圧）から当該半導体スイッチング素子１８の損失（スイッチング損失、及び、定常損失）を計算し、下相の半導体スイッチング素子１８については上相の半導体スイッチング素子１８がＯＦＦしているときに流れる相電流とＨＶ電圧（印加電圧）から当該半導体スイッチング素子１８の損失（スイッチング損失、及び、定常損失）を計算する。
　また、上下相の還流ダイオード１９の損失（スイッチング損失、及び、定常損失）については、それと複合体となる半導体スイッチング素子１８がＯＦＦしているときに還流ダイオード１９に流れる相電流とＨＶ電圧（印加電圧）から当該還流ダイオード１９の損失（スイッチング損失、及び、定常損失）を計算する。
　例えば、図３のＵ相を例に採ると、Ｕ相の相電流Ｉｕが正の値をとる期間ではＵ相の上相の電力用半導体素子１６Ｕの半導体スイッチング素子１８の損失（スイッチング損失、及び、定常損失）を計算し（それと複合体となる還流ダイオード１９の損失は零）、下相の電力用半導体素子１７Ｕの還流ダイオード１９の損失（スイッチング損失、及び、定常損失）を計算する（それと複合体となる半導体スイッチング素子１８の損失は零）。
　また、Ｕ相の相電流Ｉｕが負の値をとる期間ではＵ相の上相の電力用半導体素子１６Ｕの還流ダイオード１９の損失（スイッチング損失、及び、定常損失）を計算し（それと複合体となる半導体スイッチング素子１８の損失は零）、下相の電力用半導体素子１７Ｕの半導体スイッチング素子１８の損失（スイッチング損失、及び、定常損失）を計算する（それと複合体となる還流ダイオード１９の損失は零。他の相も同様）。尚、損失計算部３１は実施例ではＰＷＭキャリア周期の半周期で相電流（瞬時値）を取り込み、残りの半周期で損失計算を行ってジャンクション温度推定計算部３２に出力する。従って、図３中の各損失、及び、ジャンクション温度は相電流より一周期分遅れるかたちをとる。
　前述した如くジャンクション温度推定計算部３２は、損失計算部３１においてＰＷＭキャリア周期毎に算出され、出力された各電力用半導体素子１６Ｕ~１７Ｗの半導体スイッチング素子１８の損失と還流ダイオード１９の損失に熱抵抗値Ｔｒを掛ける（乗算）ことで、各損失によって生じる温度上昇値ΔＴを算出する（ステップＳ２）。
　次に、ジャンクション温度推定計算部３２は、温度センサ２２が検出する電力用半導体素子１６Ｕ~１７Ｗの近傍の温度Ｔｔｈを取り込み（ステップＳ３）、この温度Ｔｔｈに、半導体スイッチング素子１８と還流ダイオード１９の損失から算出された温度上昇値ΔＴを加えることで（式１）、各電力用半導体素子１６Ｕ~１７Ｗの半導体スイッチング素子１８のジャンクション温度Ｔｊｉと還流ダイオード１９のジャンクション温度Ｔｊｄの推定値を計算する（ステップＳ４）。
　各ジャンクション温度Ｔｊｉ、ＴｊｄはＰＷＭキャリア周期毎に算出されるので、それらの推定値は瞬時値となる。そして、算出された各電力用半導体素子１６Ｕ~１７Ｗの各ジャンクション温度Ｔｊｉ、Ｔｊｄは温度保護部３３に出力され、温度保護部３３は前述した如く、各電力用半導体素子１６Ｕ~１７Ｗの半導体スイッチング素子１８のジャンクション温度Ｔｊｉと、還流ダイオード１９のジャンクション温度Ｔｊｄのうち、最も高いものを抽出する。
　そして、当該最も高い値となるジャンクション温度Ｔｊｉ、Ｔｊｄが前述した第１の所定値ＴＳ１を超えた場合、温度保護部３３はモータ制御部２６に電流制限信号を出力し、更に、第２の所定値ＴＳ２を超えた場合は、電流遮断信号をモータ制御部２６に出力する。モータ制御部２６は温度保護部３３からの電流制限信号、又は、電流遮断信号に基づいてインバータ回路８に流れる電流を所定の値に制限し、又は、遮断するものである。
　以上詳述した如く本発明では、インバータ制御部１２が、シャント抵抗２３で検出される相電流Ｉｕ、Ｉｖ、ＩｗとＨＶ電圧（印加電圧）から電力用半導体素子１６Ｕ~１７Ｗの損失を計算する損失計算部３１と、温度センサ２２が検出する温度Ｔｔｈに、損失計算部３１が計算した電力用半導体素子１６Ｕ~１７Ｗの損失から得られる温度上昇値ΔＴを加えて、当該電力用半導体素子１６Ｕ~１７Ｗのジャンクション温度を推定するジャンクション温度推定計算部３２を有しており、ＰＷＭ制御部２７におけるＰＷＭキャリア周期毎に、損失計算部３１により電力用半導体素子１６Ｕ~１７Ｗの損失を計算し、ジャンクション温度推定計算部３２により電力用半導体素子１６Ｕ~１７Ｗのジャンクション温度を推定するようにしており、このＰＷＭキャリア周期は、インバータ回路８の相電流の周波数よりも十分に小さく、電力用半導体素子１６Ｕ~１７Ｗの熱時定数よりも十分短いので、係るＰＷＭキャリア周期毎といった高速の周期で電力用半導体素子１６Ｕ~１７Ｗのジャンクション温度の瞬時値を推定することができるようになる。
　そして、このジャンクション温度推定計算部３２が推定したＰＷＭキャリヤ周期毎の電力用半導体素子１６Ｕ~１７Ｗのジャンクション温度（瞬時値）が所定値（第１の所定値、第２の所定値）を超えた場合、所定の保護動作（電流制限、遮断）を実行するので、瞬時的な温度上昇から電力用半導体素子１６Ｕ~１７Ｗを高精度で保護することができるようになる。
　この場合、損失計算部３１は電力用半導体素子１６Ｕ~１７Ｗのスイッチング損失と定常損失から当該電力用半導体素子１６Ｕ~１７Ｗの損失を計算し、ジャンクション温度推定計算部３２は損失計算部３１が計算した電力用半導体素子１６Ｕ~１７Ｗの損失に当該電力用半導体素子の熱抵抗値Ｔｒを掛けることで温度上昇値ΔＴを算出するので、電力用半導体素子１６Ｕ~１７Ｗのジャンクション温度の瞬時値を的確に計算して推定することができるようになる。
　また、損失計算部３１はインバータ回路８の各相（Ｕ相、Ｖ相、Ｗ相）の相電流Ｉｕ、Ｉｖ、ＩｗとＨＶ電圧（印加電圧）からブリッジ構成の各相の電力用半導体素子１６Ｕ~１７Ｗの損失を計算し、温度保護部３３は最も高い電力用半導体素子１６Ｕ~１７Ｗのジャンクション温度に基づいて保護動作を実行するので、最も温度上昇が激しい相の電力用半導体素子１６Ｕ~１７Ｗを安全、且つ、正確に保護することができるようになる。
　特に、実施例の如く電力用半導体素子１６Ｕ~１７Ｗが、半導体スイッチング素子１８と還流ダイオード１９との複合体である場合には、損失計算部３１は半導体スイッチング素子１８の損失と還流ダイオード１９の損失を計算し、ジャンクション温度推定計算部３２は半導体スイッチング素子１８のジャンクション温度Ｔｊｉと還流ダイオード１９のジャンクション温度Ｔｊｄを推定するようにしたので、還流ダイオード１８を有する電力用半導体素子１６Ｕ~１７Ｗの場合にも支障無く保護を行うことができるようになる。
　更に、温度保護部３３は電力用半導体素子１６Ｕ~１７Ｗのジャンクション温度が第１の所定値を超えた場合、インバータ回路８に流れる電流を制限し、ジャンクション温度が第１の所定値より高い第２の所定値を超えた場合、インバータ回路８に流す電流を遮断するようにするので、電力用半導体素子１６Ｕ~１７Ｗの保護を確実に行いながら、不必要な電流遮断の発生を回避することが可能となる。
　そして、高温環境下で使用される実施例の如き車両用電動圧縮機１において、本発明のインバータ装置７によりモータ３を運転することで、極めて効果的な過熱保護を実現することができるようになる。
　尚、実施例ではＵ相、Ｖ相、Ｗ相の各相の電力用半導体素子１６Ｕ~１７Ｗのジャンクション温度を推定して保護を行うようにしたが、請求項４以外の発明ではそれに限らず、何れか一相のみ、或いは、二相のみの電力用半導体素子のジャンクション温度を推定して保護を行うようにしてもよい。
　また、実施例では半導体スイッチング素子１８（ＩＧＢＴ、ＭＯＳＦＥＴ）と還流ダイオード１９の複合体から成る電力用半導体素子１６Ｕ~１７Ｗを例に採って説明したが、請求項５以外の発明ではそれに限らず、還流ダイオードを有しない半導体スイッチング素子（ＩＧＢＴ、ＭＯＳＦＥＴ）のみのインバータ回路にも本発明は有効である。
　更に、実施例では車両に搭載される電動圧縮機のモータを駆動するインバータ装置で本発明を説明したが、請求項７以外の発明ではそれに限らず、ブリッジ構成の電力用半導体素子を有するインバータ回路を用いたインバータ装置全般に本発明は有効である。 Hereinafter, embodiments of the present invention will be described in detail. FIG. 1 shows a schematic sectional view of a vehicular electric compressor 1 to which the present invention is applied. The electric compressor 1 according to the embodiment constitutes a part of a refrigerant circuit of an air conditioner that air-conditions the interior of a vehicle (not shown), and is mounted in an engine room of the vehicle. The electric compressor 1 includes a motor 3 in a housing 2 and a scroll-type compression element 6 driven by a rotating shaft 4 of the motor 3. An inverter device 7 of the present invention is further attached to the housing 2, and the motor 3 is operated by the inverter device 7 to drive the compression element 6. The compression element 6 is driven by the rotating shaft 4 of the motor 3 to suck in the refrigerant from the refrigerant circuit, compress it, and discharge it again to the refrigerant circuit.
Next, FIG. 2 shows an electric circuit diagram of the inverter device 7. The inverter device 7 according to the embodiment includes a control board 11 on which an inverter circuit 8 and a smoothing capacitor 9 are mounted, and an inverter control unit 12 configured by a microcomputer (processor). The positive DC bus 13 of the inverter circuit 8 is connected to the + terminal of a vehicle battery (HV power supply for vehicle) B (not shown), and the negative DC bus 14 is connected to the negative terminal of the battery B. The smoothing capacitor 9 is connected between the two DC buses 13 and 14 of the inverter circuit 8.
The inverter circuit 8 changes the switching state of the plurality of power semiconductor elements constituting the bridge, converts the direct current applied from the battery B into alternating current, and supplies the alternating current to the motor 3. Specifically, three power semiconductor elements 16U, 16V, and 16W constituting the upper phase of the bridge and three power semiconductor elements 17U, 17V, and 17W constituting the lower phase of the bridge are provided. Each of the power semiconductor elements 16U, 16V, 16W and 17U, 17V, 17W is a composite of a semiconductor switching element 18 and a freewheeling diode 19 connected in reverse parallel thereto. DC power is supplied from the battery B to the buses 13 and 14.
In this inverter circuit 8, the upper-phase power semiconductor elements 16 U, 16 V, and 16 W and the lower-phase power semiconductor elements 17 U, 17 V, and 17 W are one-to-one. Correspondingly, they are connected in series. Hereinafter, the pair of semiconductor switching elements 18 of the power semiconductor elements 16U to 17W connected in series is referred to as a switching leg. That is, in the embodiment, the switching leg 21U configured by a pair of the semiconductor switching element 18 of the power semiconductor element 16U and the semiconductor switching element 18 of the power semiconductor element 17U, the semiconductor switching element 18 of the power semiconductor element 16V, and the power A switching leg 21V configured by a pair of semiconductor switching elements 18 of the semiconductor element 17V, a switching leg 21W configured by a pair of the semiconductor switching elements 18 of the power semiconductor element 16W and the semiconductor switching elements 18 of the power semiconductor element 17W, There is.
The switching legs 21U, 21V, and 21W are connected between the positive DC bus 13 and the negative DC bus 14, respectively. Further, the intermediate points MU, MV, MW of the respective switching legs 21U, 21V, 21W are nodes for outputting the phase voltages Vu, Vv, Vw of each phase (U phase, V phase, W phase) of the output AC. Each intermediate point MU, MV, MW is connected to each phase of the motor 3.
In the inverter circuit 8 of the embodiment, the semiconductor switching element 18 uses an IGBT (Insulated Gate Bipolar Transistor). The semiconductor switching element 18 is not limited to the IGBT and may be a MOSFET or the like. Further, a temperature sensor 22 as a temperature detector is mounted on the control board 11 in the vicinity of the power semiconductor elements 16U to 17W. In this embodiment, the temperature sensor 22 is a thermistor.
Further, a shunt resistor 23 as a phase current detector is connected to the negative DC bus 14 at a position where current from the motor 3 flows. When a current from the motor 3 flows through the shunt resistor 23, a potential difference is generated between both ends of the shunt resistor 23, and the phase currents Iu, Iv, Iw can be calculated by detecting the voltage between the both ends. The phase current detector is not limited to the shunt resistor, and may be configured with a current transformer or the like.
On the other hand, the inverter control unit 12 includes a motor control unit 26, a PWM control unit 27, a current detection unit 28, a gate driver 29, a loss calculation unit 31, a junction temperature estimation calculation unit 32, and a temperature protection unit 33. I have. The HV voltage (applied voltage) of the DC bus 13 on the positive side is input to the PWM control unit 27 and the loss calculation unit 31.
The motor control unit 26 outputs a target waveform (modulated wave) of the three-phase sine wave applied to the motor 3 to the PWM control unit 27. The PWM control unit 27 generates a duty (Duty: upper phase ON time) that is a drive signal by comparing the level of the modulated wave output from the motor control unit 26 with the level of the carrier (triangular wave). This duty is generated for each of the U-phase, V-phase, and W-phase, and is sent to the gate driver 29 that drives (ON-OFF) the gate of each semiconductor switching element 18.
The frequencies of the phase currents Iu, Iv, Iw, which are the rotation speeds of the motor 3 of the embodiment, are 400 Hz to 500 Hz, and the carrier cycle (hereinafter referred to as PWM carrier cycle) in the PWM control unit 27 is higher than that. It is 20 kHz which is sufficiently small (or short enough). In addition, the thermal time constant of the power semiconductor elements 16U to 17W (the time taken to transmit the loss temperature rise value to the temperature sensor 22) is about 50 msec, and the PWM carrier cycle is sufficiently shorter than this thermal time constant. (Or fast enough).
The current detector 28 receives the voltage across the shunt resistor 23 and calculates the phase currents Iu, Iv, Iw from the resistance value of the shunt resistor 23. The calculated phase currents Iu, Iv, Iw are input to the loss calculator 31.
The loss calculation unit 31 includes the phase currents Iu, Iv, Iw of the U-phase, V-phase, and W-phase input from the current detection unit 28 and the HV voltage (applied voltage) of the DC bus 13 on the positive side. Based on the duty input from the PWM control unit 27, the loss of each of the power semiconductor elements 16U to 17W is calculated. In the case of the embodiment, the loss calculation unit 31 includes the switching loss of the semiconductor switching element 18 constituting each of the power semiconductor elements 16U to 17W, the steady loss (conduction loss or conduction loss), the switching loss of the freewheeling diode 19, and The steady loss (conduction loss or conduction loss) is calculated separately.
The switching loss and steady loss (conduction loss or conduction loss) of the semiconductor switching element 18 are the loss of the semiconductor switching element 18 and the amount of heat generated by the semiconductor switching element 18. Further, the switching loss and steady loss (conduction loss or conduction loss) of the freewheeling diode 19 are the losses of the freewheeling diode 19 and the amount of heat generated by the freewheeling diode 19. These are losses of the power semiconductor elements 16U to 17W. The loss of each power semiconductor element 16U to 17W calculated by the loss calculation unit 31 is input to the junction temperature estimation calculation unit 32.
The junction temperature estimation calculation unit 32 increases the temperature obtained from the loss of each power semiconductor element 16U to 17W calculated by the loss calculation unit 31 to the temperature Tth in the vicinity of the power semiconductor elements 16U to 17W detected by the temperature sensor 22. By adding the value ΔT, the estimated values of the junction temperature Tji of the semiconductor switching element 18 of each power semiconductor element 16U to 17W and the junction temperature Tjd of the freewheeling diode 19 are calculated.
In this case, the junction temperature estimation calculation unit 32 multiplies the loss of each power semiconductor element 16U to 17W calculated by the loss calculation unit 31 by the thermal resistance value (heat transfer coefficient) Tr of the power semiconductor elements 16U to 17W (multiplication). ) To calculate the temperature rise value ΔT. The estimation calculation in the junction temperature estimation calculation unit 32 is expressed by the following equation (1).
Tji, Tjd = Tth + ΔT (1)
It should be noted, ΔT loss × Tr o
The calculated junction temperatures Tji and Tjd are input to the temperature protection unit 33.
The temperature protection unit 33 performs a predetermined protection operation based on the junction temperature Tji of the semiconductor switching element 18 of each power semiconductor element 16U to 17W and the junction temperature Tjd of the free wheel diode 19 estimated by the junction temperature estimation calculation unit 32. Execute. This protection operation is divided into two stages in the embodiment. First, the highest one of the junction temperatures Tji, Tjd of any one of the power semiconductor elements 16U to 17W exceeds the first predetermined value TS1. In this case, the temperature protection unit 33 outputs a current limit signal to the motor control unit 26.
When the motor control unit 26 receives the current limit signal from the temperature protection unit 33, the motor control unit 26 adjusts the modulation wave so as to limit the current flowing through the inverter circuit 8 to a predetermined value. Further, the temperature protection unit 33 sends a current interruption signal to the motor control unit 26 when the highest one of the junction temperatures Tji, Tjd exceeds a second predetermined value TS2 higher than the first predetermined value TS1. Output. When the motor control unit 26 receives the current cut-off signal from the temperature protection unit 33, the motor control unit 26 stops the output of the modulated wave, thereby turning off the semiconductor switching elements 18 of all the power semiconductor elements 16 U to 17 W and causing the inverter circuit 8 to Cut off the flowing current. The first predetermined value TS1 and the second predetermined value TS2 are values set from the temperature limits of the semiconductor switching element 18 and the free wheeling diode 19 constituting the power semiconductor elements 16U to 17W.
Next, a specific overheat protection operation according to the amount of heat generated by the power semiconductor elements 16U to 17W by the inverter control unit 12 will be described with reference to FIGS. 3 shows the waveform of the carrier used for generating the duty in the PWM control unit 27, and the period is the PWM carrier period. Below that, the U-phase, V-phase, and W-phase duties output from the PWM control unit 27 are shown. The phase current waveform shown below the U-phase phase current Iu, The fine broken line is the waveform of the V-phase phase current Iv, and the rough broken line is the waveform of the W-phase phase current Iw.
The IGBT loss shown below is the loss of the semiconductor switching element 18 calculated by the loss calculation unit 31. Similarly, the solid line is the loss of the U-phase power semiconductor element 16U, the semiconductor switching element 18 of the 17U, and the fine broken line is the V-phase. The loss of the semiconductor switching element 18 of the power semiconductor elements 16V and 17V and the rough broken line are the loss of the semiconductor switching element 18 of the W-phase power semiconductor elements 16W and 17W.
The diode loss shown below is the loss of the freewheeling diode 19 calculated by the loss calculating unit 31. Similarly, the solid line is the loss of the U-phase power semiconductor element 16U, the freewheeling diode 19 of 17U, and the fine broken line is the V-phase power. The loss of the free-wheeling diode 19 of the semiconductor elements 16V and 17V and the rough broken line are the loss of the free-wheeling diode 19 of the W-phase power semiconductor elements 16W and 17W. The temperature sensor temperature below it is a temperature Tth in the vicinity of the power semiconductor elements 16U to 17W detected by the temperature sensor 22.
The estimated IGBT junction temperature shown below is the junction temperature Tji of the semiconductor switching element 18 calculated by the junction temperature estimation calculation unit 32. Similarly, the solid line shows the semiconductor switching elements of the U-phase power semiconductor elements 16U and 17U. 18 is a junction temperature Tji, a fine broken line is the junction temperature Tji of the V-phase power semiconductor element 16V and 17V, and a rough broken line is the junction temperature of the W-phase power semiconductor element 16W and 17W of the semiconductor switching element 18. Tji. In addition, a dashed-dotted line is an average value of junction temperature Tji.
The estimated diode junction temperature shown below is the junction temperature Tjd of the freewheeling diode 19 calculated by the junction temperature estimation calculating unit 32. Similarly, the solid line is the junction temperature of the freewheeling diode 19 of the U-phase power semiconductor elements 16U and 17U. Tjd, the fine broken line is the junction temperature Tjd of the V-phase power semiconductor elements 16V and 17V, and the rough broken line is the junction temperature Tjd of the W-phase power semiconductor elements 16W and 17W. In addition, a dashed-dotted line is the average value of junction temperature Tjd.
FIG. 4 shows the flow of estimation calculation of the junction temperature executed by the inverter control unit 12. In the present invention, the inverter control unit 12 calculates the amount of heat generated by the semiconductor switching element 18 (IGBT) and the freewheeling diode 19 (Diode) for each PWM carrier period. That is, the loss calculation unit 31 calculates each phase (for each PWM carrier cycle obtained from the duty output from the PWM control unit 27 from the HV voltage (applied voltage) during driving and the instantaneous values of the phase currents Iu, Iv, and Iw. The losses of the power semiconductor elements 16U to 17W of the U-phase, V-phase, and W-phase upper and lower phases are calculated (step S1).
Here, in each of the power semiconductor elements 16U to 17W, the current always flows only to either the semiconductor switching element 18 or the freewheeling diode 19, and therefore the loss of the freewheeling diode 19 during the period in which the current flows through the semiconductor switching element 18 Is zero, and conversely, the loss of the semiconductor switching element 18 is zero during the period in which the current flows through the freewheeling diode 19.
Therefore, as described above, the loss calculating unit 31 calculates the loss of the semiconductor switching element 18 from the phase current and HV voltage (applied voltage) flowing when the upper phase semiconductor switching element 18 is ON based on the duty of each phase. (Switching loss and steady loss) are calculated, and for the lower-phase semiconductor switching element 18, the semiconductor switching is performed from the phase current and HV voltage (applied voltage) that flow when the upper-phase semiconductor switching element 18 is OFF. The loss (switching loss and steady loss) of the element 18 is calculated.
Further, regarding the loss (switching loss and steady loss) of the upper and lower phase freewheeling diodes 19, the phase current and the HV voltage (applied to the freewheeling diode 19 when the semiconductor switching element 18 which is a composite with the loss is switched off. The loss (switching loss and steady loss) of the freewheeling diode 19 is calculated from the voltage).
For example, taking the U phase in FIG. 3 as an example, the loss (switching loss, and loss of the semiconductor switching element 18 of the power semiconductor element 16U in the upper phase of the U phase during the period in which the phase current Iu of the U phase takes a positive value. , Steady loss) (the loss of the freewheeling diode 19 that is a composite with it is zero), and the loss of the freewheeling diode 19 of the lower-phase power semiconductor element 17U (switching loss and steady loss) is calculated (and The loss of the semiconductor switching element 18 as a composite is zero).
Further, during the period in which the phase current Iu of the U phase takes a negative value, the loss (switching loss and steady loss) of the return diode 19 of the power semiconductor element 16U for the upper phase of the U phase is calculated (and the composite and The loss of the semiconductor switching element 18 is zero), and the loss (switching loss and steady loss) of the semiconductor switching element 18 of the lower-phase power semiconductor element 17U is calculated (the loss of the freewheeling diode 19 that is combined with it) Zero, other phases as well). In the embodiment, the loss calculation unit 31 takes in a phase current (instantaneous value) in a half cycle of the PWM carrier cycle, performs loss calculation in the remaining half cycle, and outputs the loss calculation to the junction temperature estimation calculation unit 32. Therefore, each loss in FIG. 3 and the junction temperature are delayed by one cycle from the phase current.
As described above, the junction temperature estimation calculation unit 32 is calculated by the loss calculation unit 31 for each PWM carrier cycle, and the output power loss of the semiconductor switching elements 18 of the power semiconductor elements 16U to 17W and the loss of the freewheeling diode 19 are heated. By multiplying (multiplying) the resistance value Tr, a temperature rise value ΔT caused by each loss is calculated (step S2).
Next, the junction temperature estimation calculation unit 32 takes in the temperature Tth in the vicinity of the power semiconductor elements 16U to 17W detected by the temperature sensor 22 (step S3), and at this temperature Tth, the semiconductor switching element 18 and the free wheel diode 19 By adding the temperature increase value ΔT calculated from the loss (Equation 1), the estimated values of the junction temperature Tji of the semiconductor switching element 18 and the junction temperature Tjd of the free-wheeling diode 19 of each of the power semiconductor elements 16U to 17W are calculated ( Step S4).
Since each junction temperature Tji, Tjd is calculated for every PWM carrier period, those estimated values are instantaneous values. The calculated junction temperatures Tji and Tjd of the power semiconductor elements 16U to 17W are output to the temperature protection unit 33, and the temperature protection unit 33, as described above, outputs the semiconductor switching elements of the power semiconductor elements 16U to 17W. The highest one of the junction temperature Tji of 18 and the junction temperature Tjd of the reflux diode 19 is extracted.
When the junction temperature Tji, Tjd that is the highest value exceeds the first predetermined value TS1, the temperature protection unit 33 outputs a current limiting signal to the motor control unit 26, and further, the second predetermined value TS1. When the value TS2 is exceeded, a current interruption signal is output to the motor control unit 26. The motor control unit 26 limits or cuts off the current flowing through the inverter circuit 8 to a predetermined value based on the current limit signal or the current cut-off signal from the temperature protection unit 33.
As described above in detail, in the present invention, the inverter control unit 12 calculates the loss of the power semiconductor elements 16U to 17W from the phase currents Iu, Iv, Iw detected by the shunt resistor 23 and the HV voltage (applied voltage). The temperature increase value ΔT obtained from the loss of the power semiconductor elements 16U to 17W calculated by the loss calculation unit 31 is added to the temperature Tth detected by the calculation unit 31 and the temperature sensor 22, and the power semiconductor elements 16U to 17W A junction temperature estimation calculation unit 32 for estimating the junction temperature of the power, and for each PWM carrier period in the PWM control unit 27, the loss calculation unit 31 calculates the loss of the power semiconductor elements 16U to 17W to estimate the junction temperature. The calculation unit 32 estimates the junction temperature of the power semiconductor elements 16U to 17W. Since the carrier period is sufficiently smaller than the frequency of the phase current of the inverter circuit 8 and sufficiently shorter than the thermal time constant of the power semiconductor elements 16U to 17W, the power semiconductor element 16U has a high-speed period such as every PWM carrier period. The instantaneous value of the junction temperature of ~ 17W can be estimated.
The junction temperature (instantaneous value) of the power semiconductor elements 16U to 17W for each PWM carrier period estimated by the junction temperature estimation calculation unit 32 exceeds a predetermined value (first predetermined value, second predetermined value). In this case, since the predetermined protection operation (current limitation, cutoff) is executed, the power semiconductor elements 16U to 17W can be protected with high accuracy from an instantaneous temperature rise.
In this case, the loss calculating unit 31 calculates the loss of the power semiconductor elements 16U to 17W from the switching loss and the steady loss of the power semiconductor elements 16U to 17W, and the junction temperature estimation calculating unit 32 is calculated by the loss calculating unit 31. Since the temperature rise value ΔT is calculated by multiplying the loss of the power semiconductor elements 16U to 17W by the thermal resistance value Tr of the power semiconductor elements, the instantaneous value of the junction temperature of the power semiconductor elements 16U to 17W is accurately calculated. And can be estimated.
In addition, the loss calculation unit 31 uses the phase currents Iu, Iv, Iw of each phase (U phase, V phase, W phase) and the HV voltage (applied voltage) of the inverter circuit 8 to power semiconductor elements 16U for each phase in a bridge configuration. The temperature protection unit 33 performs a protection operation based on the junction temperature of the highest power semiconductor element 16U to 17W, so that the power semiconductor element 16U to 17W of the phase where the temperature rises most rapidly is calculated. Safe and accurate protection can be achieved.
In particular, when the power semiconductor elements 16U to 17W are a composite body of the semiconductor switching element 18 and the free wheel diode 19 as in the embodiment, the loss calculation unit 31 calculates the loss of the semiconductor switching element 18 and the loss of the free wheel diode 19. Since the junction temperature estimation calculation unit 32 estimates the junction temperature Tji of the semiconductor switching element 18 and the junction temperature Tjd of the freewheeling diode 19, in the case of the power semiconductor elements 16U to 17W having the freewheeling diode 18, Can be protected without any problem.
Further, when the junction temperature of the power semiconductor elements 16U to 17W exceeds the first predetermined value, the temperature protection unit 33 limits the current flowing through the inverter circuit 8, and the second temperature is higher than the first predetermined value. Since the current flowing through the inverter circuit 8 is cut off when the predetermined value is exceeded, it is possible to avoid unnecessary current interruption while reliably protecting the power semiconductor elements 16U to 17W. It becomes.
And in the electric compressor 1 for vehicles like the Example used in a high temperature environment, by operating the motor 3 by the inverter apparatus 7 of this invention, it can implement | achieve very effective overheat protection. Become.
In the embodiment, the junction temperatures of the power semiconductor elements 16U to 17W of the U phase, V phase, and W phase are estimated and protected. However, the invention other than claim 4 is not limited thereto. You may make it protect by estimating the junction temperature of the power semiconductor element only for any one phase or only two phases.
Further, in the embodiment, the power semiconductor elements 16U to 17W made of the composite of the semiconductor switching element 18 (IGBT, MOSFET) and the freewheeling diode 19 have been described as examples, but the invention other than claim 5 is not limited thereto, The present invention is also effective for an inverter circuit having only a semiconductor switching element (IGBT, MOSFET) having no freewheeling diode.
Furthermore, in the embodiments, the present invention has been described with an inverter device that drives a motor of an electric compressor mounted on a vehicle. However, the invention is not limited to the invention other than claim 7, and an inverter circuit having a power semiconductor element in a bridge configuration. The present invention is effective for all inverter devices using the.
　１　電動圧縮機
　２　ハウジング
　３　モータ
　４　回転軸
　６　圧縮要素
　７　インバータ装置
　８　インバータ回路
　１１　制御基板
　１２　インバータ制御部
　１６Ｕ~１７Ｗ　電力用半導体素子
　１８　半導体スイッチング素子
　１９　還流ダイオード
　２２　温度センサ（温度検出器）
　２３　シャント抵抗（相電流検出器）
　２６　モータ制御部
　２７　ＰＷＭ制御部
　２８　電流検出部
　２９　ゲートドライバ
　３１　損失計算部
　３２　ジャンクション温度推定計算部
　３３　温度保護部 DESCRIPTION OF SYMBOLS 1 Electric compressor 2 Housing 3 Motor 4 Rotating shaft 6 Compression element 7 Inverter device 8 Inverter circuit 11 Control board 12 Inverter control part 16U-17W Power semiconductor element 18 Semiconductor switching element 19 Reflux diode 22 Temperature sensor (temperature detector)
23 Shunt resistor (phase current detector)
26 Motor control unit 27 PWM control unit 28 Current detection unit 29 Gate driver 31 Loss calculation unit 32 Junction temperature estimation calculation unit 33 Temperature protection unit

Claims (7)

  1. 　ブリッジ構成を成す電力用半導体素子を有するインバータ回路と、前記電力用半導体素子を駆動するＰＷＭ制御部を有するインバータ制御部を備えたインバータ装置において、
    　前記電力用半導体素子近傍の温度を検出する温度検出器と、前記インバータ回路の相電流を検出する相電流検出器を備え、
    　前記インバータ制御部は、
    　前記相電流検出器が検出する少なくとも一相の相電流と印加電圧から前記電力用半導体素子の損失を計算する損失計算部と、
    　前記温度検出器が検出する温度に、前記損失計算部が計算した前記電力用半導体素子の損失から得られる温度上昇値を加えて、当該電力用半導体素子のジャンクション温度を推定するジャンクション温度推定計算部と、を有し、
    　前記ＰＷＭ制御部におけるＰＷＭキャリア周期毎に、前記損失計算部により前記電力用半導体素子の損失を計算し、前記ジャンクション温度推定計算部により前記電力用半導体素子のジャンクション温度を推定すると共に、
    　該ジャンクション温度推定計算部が推定した前記ＰＷＭキャリヤ周期毎の前記電力用半導体素子のジャンクション温度が所定値を超えた場合、所定の保護動作を実行することを特徴とするインバータ装置。     In an inverter device including an inverter circuit having a power semiconductor element having a bridge configuration and an inverter control unit having a PWM control unit for driving the power semiconductor element,
    A temperature detector that detects the temperature in the vicinity of the power semiconductor element; and a phase current detector that detects a phase current of the inverter circuit;
    The inverter control unit
    A loss calculation unit for calculating a loss of the power semiconductor element from at least one phase current detected by the phase current detector and an applied voltage;
    Junction temperature estimation calculation unit that estimates a junction temperature of the power semiconductor element by adding a temperature rise value obtained from the loss of the power semiconductor element calculated by the loss calculation unit to the temperature detected by the temperature detector And having
    For each PWM carrier period in the PWM control unit, the loss calculation unit calculates the loss of the power semiconductor element, and the junction temperature estimation calculation unit estimates the junction temperature of the power semiconductor element,
    An inverter device, wherein a predetermined protection operation is performed when a junction temperature of the power semiconductor element for each PWM carrier period estimated by the junction temperature estimation calculation unit exceeds a predetermined value.
  2. 　前記ＰＷＭキャリア周期は前記インバータ回路の相電流の周波数よりも十分に小さく、前記電力用半導体素子の熱時定数よりも十分短いことを特徴とする請求項１に記載のインバータ装置。 The inverter device according to claim 1, wherein the PWM carrier cycle is sufficiently smaller than a frequency of a phase current of the inverter circuit and sufficiently shorter than a thermal time constant of the power semiconductor element.
  3. 　前記損失計算部は、前記電力用半導体素子のスイッチング損失と定常損失から当該電力用半導体素子の損失を計算し、
    　前記ジャンクション温度推定計算部は、前記損失計算部が計算した前記電力用半導体素子の損失に当該電力用半導体素子の熱抵抗値を掛けることで前記温度上昇値を算出することを特徴とする請求項１又は請求項２に記載のインバータ装置。     The loss calculator calculates the loss of the power semiconductor element from the switching loss and steady loss of the power semiconductor element,
    The junction temperature estimation calculation unit calculates the temperature increase value by multiplying a loss of the power semiconductor element calculated by the loss calculation unit by a thermal resistance value of the power semiconductor element. The inverter apparatus of Claim 1 or Claim 2.
  4. 　前記損失計算部は、前記インバータ回路の各相の相電流と印加電圧からブリッジ構成の各相の前記電力用半導体素子の損失を計算すると共に、
    　前記インバータ制御部は、最も高い前記電力用半導体素子のジャンクション温度に基づいて前記保護動作を実行することを特徴とする請求項１乃至請求項３のうちの何れかに記載のインバータ装置。     The loss calculation unit calculates the loss of the power semiconductor element of each phase of the bridge configuration from the phase current and applied voltage of each phase of the inverter circuit,
    4. The inverter device according to claim 1, wherein the inverter control unit performs the protection operation based on a junction temperature of the highest power semiconductor element. 5.
  5. 　前記電力用半導体素子は、半導体スイッチング素子と還流ダイオードとの複合体であり、前記損失計算部は、前記半導体スイッチング素子の損失と前記還流ダイオードの損失を計算し、前記ジャンクション温度推定計算部は、前記半導体スイッチング素子のジャンクション温度と前記還流ダイオードのジャンクション温度を推定することを特徴とする請求項１乃至請求項４のうちの何れかに記載のインバータ装置。 The power semiconductor element is a composite of a semiconductor switching element and a free wheel diode, the loss calculation unit calculates a loss of the semiconductor switching element and a loss of the free wheel diode, and the junction temperature estimation calculation unit includes: The inverter device according to claim 1, wherein a junction temperature of the semiconductor switching element and a junction temperature of the free-wheeling diode are estimated.
  6. 　前記インバータ制御部は、前記電力用半導体素子のジャンクション温度が第１の所定値を超えた場合、前記インバータ回路に流れる電流を制限すると共に、
    　前記ジャンクション温度が前記第１の所定値より高い第２の所定値を超えた場合、前記インバータ回路に流す電流を遮断することを特徴とする請求項１乃至請求項５のうちの何れかに記載のインバータ装置。     When the junction temperature of the power semiconductor element exceeds a first predetermined value, the inverter control unit limits a current flowing through the inverter circuit, and
    6. The current flowing through the inverter circuit is cut off when the junction temperature exceeds a second predetermined value that is higher than the first predetermined value. Inverter device.
  7. 　請求項１乃至請求項６のうちの何れかに記載のインバータ装置により運転されるモータを備えて車両に搭載されることを特徴とする車両用電動圧縮機。 An electric compressor for a vehicle comprising a motor operated by the inverter device according to any one of claims 1 to 6 and mounted on a vehicle.
PCT/JP2017/029582 2016-09-14 2017-08-10 Inverter apparatus and vehicle electric compressor provided with same WO2018051719A1 (en)

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CN113320431A (en) * 2021-07-07 2021-08-31 西安星源博睿新能源技术有限公司 Method, device and system for dynamically adjusting temperature protection point of charging module of electric vehicle

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