WO2013166579A1 - Dc-dc converter circuit using an llc circuit in the region of voltage gain above unity - Google Patents

Abstract

A method of operating a resonant DC-DC converter is provided where the resonant DC-DC converter includes a high voltage boost LLC circuit. The method includes providing variable power flow control to the LLC circuit with externally determined input and output voltages using frequency control. Frequency control is applied such that it emulates different loading conditions. For fixed input and output voltages this corresponds to operating along horizontal curves on the voltage gain compared to the switching frequency operating plane. A DC-DC converter is also provided including (A) a low voltage full-bridge or half-bridge DC- AC converter; (B) an LLC resonant tank; (C) a high voltage AC-DC converter or rectifier; and (D) a high voltage controllable switch; wherein the high voltage controllable switch is controllable to regulate power flow from an input to an output of the DC-DC converter based on an externally determined voltage gain ratio, wherein the LLC resonant lank operates with a minimum boosting having an effective value above unity over the entire operating range. A method of designing a resonant DC-DC converter for high voltage boost ratio is also provided.

Description

DC-DC CONVERTER CIRCUIT USING AN LLC CIRCUIT IN THE REGION OF VOLTAGE GAIN ABOVE UNITY

PRIORITY CLAIM

This application claims priority to U,S- Patent Application No. 13/469,060 filed on May 10^th, 2012, which is hereby incorporated by reference in its entire (the "Base Patent").

FIELD OF THE INVENTION

This present invention relates generally to a power converting appartuus, and more specifically to a DC-DC converter using an LLC circuit in the region of voltage gain above unity.

BACKGROUND TO THE INVENTION

Direct current (DC) architectures are well known, for example for the transmission and distribution of power. DC architectures generally provide efficient (low loss) diiiribution of electrical power relative to alternating c urrent (AC) architectures.

The importance of DC architectures has increased because of factors including; (1) the reliance of computing and telecommunications equipment on DC input power; (2) the reliance of variable speed AC and DC drives on DC input ,>ower; (3) the production of DC power by solar photovoltaic systems, fuel cells, and various wind turbine technologies; (4) propulsion systems in electric and hybrid vehicles, marine applications; (5) aerospace applications; (6) micro-grids and smart grids, including the above, energy storage and electric charging stations; and (7) other systems that require converters with varying input voltage and load.

The widespread use of DC architectures has also expanded the need for I>C-DC power convener circuits. Moreover, there is a further need for DC-DC power converter circuits that are efficient and low cost.

I Traditionally, cost reduction is achieved in part by (1) reducing the components of DC-DC power converters, and (2) increasing the switching frequency of DC-DC power converters. These cost reduction methods can be achieved by implementing transformerless DC-DC converters that switch at high frequency. High frequency operation allows the circuit designer to reduce the size, and therefore the cost, of expensive components such as transformers, inductors and capacitors. Two of the most common transformerless DC-DC converters are the buck converter 10, as shown in FIG. 1 , for stepping down the voltage, and the boost converter 12, as shown in PIG. 2, for stepping up the voltage. While both of these circuits are capable of achieving very high conversion efficiency when the input-to-output voltage ratio is near unity and the switching frequency is relatively low, their efficiency is less t an optimal when the oltage ratio becomes high or the switching frequency is increased lo reduce the tot il size of the converter. In addition, in dieir basic form they do not provide galvanic isolation. Loss of efficiency, along with other operational problems, are caused by circuit parasitics, including such circuit effects as diode forward voltage drop, switch and diode conduction losses, switching losses, switch capacii ances, inductor winding capacitance, and lead and trace inductances- Furthermore, it is known in the prior art that boost converters in particular are susceptible to parasitic effects and high efficiency operation requires low step up ratios, e.g. 1 :2 or 1 :3.

B, Buti, P. Bartal, I. Nagy, "Resonant boost converter operating above its resonant frequency," EPE, Dresden, 2005, is an example of a resonant DC-DC power converter, where a resonant tank is excited at its resonant frequency to achieve high siep-up/step-do n conversion ratios without the use of transformers. An Pi- bridge based resonant DC-DC power converter was proposed by D. Jovi:ic (D. Jovc^jc, "Step-up MW DC-DC converter for MW size applications," Institute of Engineering Technology, paper IET-2009-407) and modified for eniianced modularity by A. Abbas and P.Lehn (A. Abbas, P. Lehn, "Power electronic circuits for high voltage DC to DC converters," University of Toronto, In- ention disclosure R1S#10001913, 2009-03-31). The converter disclosed in B. Bud, P. Bartal, I. Nagy, "Resonant boost con^verter operating above its resonant frequency," EPE, Dresden, 2005, requires two perfectly, or near to perfectly, matched inductors, each only utilized half of the lime, to function properly. Perfect matching is not viable in many applies dons. Moreover, the fact that the inductor is only utilized half of the time ef fee l ively doubles the inductive requirements of the circuit. This is undesirable as the inductor is typically the single most expensive component in the power circuit. Furthermore, the converter in B. Buti, P. Bartal, I. Nagy, "Resonant boost converter o eratin above its resonant frequency," EPE, Dresden, 2005, requires both a positive and negative input supply. This is often not available.

The converters disclosed in D. Jovcic, "Step-up MW DC-DC converter fo MW size applications," Institute of Engineering Technology, paper 1ET-2009- 07, and A. Abbas, P. Lehn, "Power electronic circuits for high voltage DC lo DC converters," University of Toronto, Invention disclosure RIS#10001 13, 20⁽19-03- 31, uses four high voltage reverse blocking switching devices. For medium frequency applications (approx. 20kH¾ - 100kHz) such devices are not leadily available thus they need to be created out of a series combination of an insulated- gate bipolar transistor ("IGBT") and a diode, or a metal oxide semiconductor field effect transistor ("MOSFET") MOSFET and a diode. This not only further increases system cost but it also nearly doubles the device conduction losses of the converter.

Galvanic isolation and larger voltage boost and buck ratios are possibk with resonant and quasi-resonant DC-DC converters. These converters use inductive and capacitive components to shape the currents and/or voltages so that the switching losses are reduced allowing higher switching frequencies without a large efficiency penalty as explained in N. Mohan, T. Undeland, . Robbins, "Power electronics: converters, applications, and design," Wiley, 1995. Rt sonant and quasi-resonant DC-DC conveners can be implemented with or ithout galvanic isolation.

A resonant converter with galvanic isolation is found in Bor-Ren Liu ami Shin- Feng Wu, "ZVS Resonant Converter With Series-Connected Transformers," ndustrial Electronics, IEEE Transactions on, vol, 58, No. 8, pp. 3547-3554, Aug- 2011, I this work, a series resonant converter is implemented with mu ltiple transformers connected in series. The proposed convener is designed to be ui ed as a power factor pre-regulator in consumer electronic applications. The converter operates near the characteristic frequency defined by the resonant capacitor and resonant inductor. ZVS is achieved for all of ihe input switching components.

This conveiter analyzed by Bor-Ren Lin and Shin-Feng Wu uses a convencional resonant converter design approach. The resonant tank is only able to piovide minimal voltage boosting, if necessary, and any voltage boosting or bucking must come entirely from the iransformET turns ratio. The small amount of v illage boosting that can be provided is used when the input voltage is low. Furthei more, due to the resonant tank design, this converter would not be suitable to conti ol the power flow between an input and an output voltage source.

Series resonant conveners and parallel resonant converters are known to bi very efficient for a small range of operating points. They can be implemented v ithout galvanic isolation or with galvanic isolation . For applications that require a large range of input voltages and loads, they are not ideal. As shown in B. Yang, "Topology Investigation for Front End DC/DC Power Conversion for Disb ibuted Power System", Ph.D. Dissertation, Virginia Tech, 2003, both series resonant conveners and parallel resonant converters suffer from large circulating currents, and large switching currents when the input voltage is high.

In B. Yang, "Topology Investigation for Front End DC/DC Power Conversion for Distributed Power System", Ph.D. Dissertation, Virginia Tech, 2003 the author shows that some of the limitations in traditional series resonant or parallel resonant converters can be overcome by using an LLC resonant converter.

R. L- Lin and C. W. Lin, "Design criteria for resonant tank οΓ LLC DC to DC resonant converter", ΓΕΕΕ 2010, presents a conventional design approach to obtain an LLC step down converter. The designed converter has a madmum voltage gain from the resonant tank of only 1.44, which is needed when the input voltage is at a minimum. For high input voltage the circuit is operated at, or just below, unity gain. A 9:1 transformer provides the net voltage step down needed for the application.

H. Hu, X, Fang, Q. Zhang, Z. Shen, and I. Batarseh, "Optimal design considerations for a modified LLC converter with wide input voltage r nge capability suitable for PV applications," ECCE 201 ] , is an example of a conventional LLC design methodology applied to a step up converter where the resonant circuit provides close to unity gain. All of the voltage gain is achieved through the output transformer.

In both of the works of R. L. Lin et. al. and H. Hu et. al., the conventional 1 J_-C design methodology used yields a resonant tank with very low voltage boosting properties. Furthermore, both designs require a resistive load ai the output for proper functionality. These converters, and all LLC converters designed with the conventional method, are not suitable for applications where the power How between iwo voltage sources is regulated. In US Patent 6,344,979 an LLC converter is claimed where the converter is opcraied between the two characteristic frequencies of the converter, ω = _^fz ^"^ ^. and at = + ί._Γ)ς., to maintain output voltage regulation. However, the authors failed to address the high voltage gain region of operation and the advantages of operating there, as well as how, by choosing the right compone nts, the designer can always ensure operation in this region. In addition, the ; ero current switching region of operation, designated as "LHS Operation" in ihis document, was noi ui.iliz.ed nor were the benefits of operating in this region identified. The "LHS Operation" region is also only usable by a careful selec ion of resonant tank components, as identified in the current invention. SUMMARY OF INVENTION

In one aspect of the invenlion, a method of operating a resonant DC-DC converter is provided, the resonant DC-DC converter comprising a high voltage boost L LC circuit, wherein the method comprises providing variable power flow control to the LLC circuit with externally determined input and output voltages using frequency control.

In a further aspect of the invention, the externally determined output voltage is created by cither a single externally determined output voltage, or a series connection of two externally determined output voltages to create a bi- olar output.

In another aspect of the invention, a method is provided wherein frequency control is applied such that it emulates different loading conditions thus operating along horizontal curves on the voltage gain compared Lo the switching frequency operating plane.

In a still other aspect of the invention, the LLC circuit includes an LLC resoaant tank, and wherein the LLC resonant tank operates with a minimum boo; ting having an eiPfeclive value that is above unity over Lhe entire operating range.

In another aspect, of the method of the invention, the minimum boosting results in controllable transfer of power via change of switching frequency.

In another aspect of the invention, the method further comprises maintaining an externally determined voltage gain and using frequency control to enable movement between the load curves, and lo control this movement within a frequency control region where there is horizontal separation amongst the load curves.

In another aspect of the invention, the method further comprises: (A) operating the high voltage boost LLC circuit in a region close to a resonant frequency determined by a resonant inductor, magnetizing inductor and a resonant capat itor, to achieve a high vohage boosL; and (B) utilizing unipolar or bipolar resonant tank excitation to improve converter efficiency in the high voltage boost circuit.

In yet another aspect of the invention, a balanced bipolar DC output is pro ided wherein the output capacitor voltages are automatically balanced. In a still other aspect of the invention, the DC-DC convener further includes a resonant inductor, a magnetizing inductor and a resonant capacitor, and the method comprises the further step of selecting these components such that the yield over the entire range of operation is an effective voltage gain that is greater than unity.

In another aspect of the invention, the LLC converter is implemented wiih a transformer to allow decoupling of the resonant circuit gain from the externally determined voltage gain.

In a still other aspect of the invention, the effective voltage gain value and the components are selected so as to minimize the effective voltage gain of the resonant circuit, while being greater than unity, and provide controllability oi the DC-DC converter via frequency.

In another aspect of the invention, the method further comprises operating .it a range of input stage switching frequencies in an LLC circuit whereby a c nge in input voltage results in a change in load or transferred power, such th t a decoupling between the input voltage and load is not required.

In one aspect of the invention, a resonant DC-DC convener is provided for I dgh voltage step-up ratio, where the resonant DC-DC converter for high voltage step- up ratio comprises: (A) a low voltage full-bridge or half-bridge DC-AC conve ter; (B) an LLC resonant tank; (C) a high voltage AC-DC converter or rectifier; and (D) a high voltage controllable switch; wherein the high voltage controllable switch is controllable to regulate power flow from an input to an output of the DC-DC converter based on an externally determined voltage gain ratio, wherein the LLC resonant tank operates with a minimum boosting having an effec tive value above unity over the entire operating range.

In another aspect of the invention, the DC-DC converter is designed to provide variable power flow control using frequency control.

In another aspect of the invention, a DC-DC converter is provided wherein application of frequency control emulates different loading conditions hus enabling operation along horizontal curves on a voltage gain compared 10 a switching frequency operating plane.

In another aspect of the invention, a DC-DC converter is provided wherein the minimum boosLing results in controllable transfer of power based on ch nge of switching frequency.

In yet another aspect of the invention, a DC-DC converter is provided that maintains an externally detennined voltage gain, and uses frequency control to enable movement between the load curves, and controls this movement with in a frequency control region where there is horizontal separation amongst the load curves.

In another aspect of the invention, a DC-DC converter is provided that is designed for: (A) operation of a high voltage boost LLC circuit in a region close to a resonant frequency determined by a resonant inductor, magnetizing inductor and a resonant capacitor, to achieve a high voltage boost; and (B) use of unipolar or bipolar resonant tank excitation to improve converter efficiency in the high voltage boost circuit.

In another aspect of the present invention, a DC-DC converter is provided that further comprises a balanced bipolar DC output wherein output capacitor volt iges are automatically balanced.

In a still other aspect, a DC-DC converter is provided that further includes a resonant inductor, a magnetizing inductor and a resonant capacitor, these components being selected such that the yield over tie entire range of operaiii >n is an effective voltage gain that is greater than unity.

In yet another aspect of the invention, a DC-DC converter is provided that comprises a transformer to allow decoupling of the resonant circuit gain iron the externally determined voltage gain.

In a still other aspect of the invention, a DC-DC converter if provided wherein the components are selected so as to minimize the effective voltage gain ol the resonant circuit, while being greater than unity, and provide controllability of the DC-DC converter via frequency.

In one aspect of the invention, a method of designing a resonant DC-DC converter for high voltage boost ratio is provided, the DC-DC converter comprising: (Λ) a low voltage full-bridge or half-bridge DC-AC converter; (B) an LLC resonant tank; (C) a high voltage AC-DC convener or rectifier; and (D) optionally, a high voltage controllable switch; wherein the high voltage controllable swilcli is controllable to regulate power flow from an inpui to an output of the DC DC converter based on a externally determined input to output voltage gain : atio maintained by the high voltage controllable switch using frequency control, wherein the DC-DC converter includes (i) a resonant capacitor, (ii) a resonant inductor, and (iii) a magnetizing inductor; wherein the design method comprises: (i) determining a minimum gain sufficient to enable high-resolution coiitrc l of frequency using available control hardware; (ii) selecting an l^JL, ratio th.it is suitable for an application for the DC-DC converter; (iii) generating voltage gain curves for various values of Q, and plotting these values so as to graph a boundary curve that defines LHS and RHS regions, and selecting the Q values wliose voltage gain curve intersects with boundary curve at the maximum voltage boost ratio, thereby defining a set of normalized frequency values; and (iv) using Hie Q values and the normalized frequency values found to calculate values fo> the resonant csipacitor, the resonant inductor, and the magnetizing inductor so is to enable selection of suitable components for the application.

In another aspect of the invention, a method of designing a resonant DC -DC converter lor high voltage boost ratio, the DC-DC converter comprising: (A) a low voltage full-bridge or half-bridge DC- AC converter; (B) an LLC res< mailt tank; (C) a high voltage AC-DC converter or rectifier; and (D) optionally, a high voltage controllable switch; wherein the high voltage controllable switi h is controllable to regulate power How from an input to an oulput of the DC-DC converter based on a externally determined input to output voltage gain ;-atio. Power flow control is maintained using frequency control. The DC-DC com ertcr includes (i) a resonant capacitor, (ii) a resonant inductor, and (iii) a magneiizing inductor; wherein the design method comprises; (I) determining a minimum gain sufficient to enable high-resolution control of frequency using available control hardware; (2) selecting an IVL_r ratio that is suitable for an application fo^■ the DC-DC converter; (3) generating voltage gain curves for various values of Q, and plotting these values so as to graph a boundary curve that defines LHS and RHS regions, and selecting the Q values whose voltage gain curve intersects ith boundary curve at the maximum voltage boost ratio, thereby defining a se t of normalized frequency values; and (4) using the Q values and the normalized frequency values found to calculate values for the resonant capacitor, the resonant inductor, and the magnetizing inductor so as to enable selection of suitable components for the application.

It is understood that the invention is capable of operating with other resonant converter configuration known in previous art and/or used in diffej ent applications. It is also understood that the invention is usable in applications v ith different grounding requirements including floating systems, high impedance grounded systems, and solidly grounded systems and that the use or not of a transformer may be influenced by the grounding requirements, fn this respect, before explaining at least one embodiment of the invention in detail, it is to be understood that the invention is not limited in its application to the details of construction and to the arrangements of the components set forth in the following description or illustrated in the drawings. The invention is capable of other embodiments and of being practiced and carried out in various ways. Also, it is to be understood that ihe phraseology and terminology employed her« in are for the purpose of description and should not be regarded as limiting. HJRIEF DESCRIPTION OF THE DRAWINGS

The invention will be betlcr understood and objects of the invention will become apparent when consideration is given to the following detailed description thercc f. Such description makes reference to the annexed drawings wherein:

FIG. 1 is a circuit diagram illustrating a prior art buck converter. FIG. 2 is a circuit diagram illustrating a prior art boost converter.

FJGS. 3(a), 3(b) and 3(c) illustrate three representative implementationii of the half-bridge resonant DC -DC converter, having a single high voltage switch.

FIGS. 4(a) and 4(b) illustrate an implementation of a full-bridge resonant DC-! C converter.

FIGS. 5(a), 5(b), 5(c) and 5(d) illustrate four representative implementations of the full-bridge resonant DC-DC converter of the present invention, having a sin gle high voltage switch and a common ground on the input and the output.

FIGS. 6(a), 6(b) and 6(c) illustrate the three representative circuits of an alternate implementation of the circuit design of the present invention that include transformer.

FIG. 7 is the implementation of FIG. 6(c), using MOSFET switches, with he addition of a anubber diode.

HG. 8 illustrates the voltage and current waveforms associated in operation w ith the circuit of FIG. 7.

FIG. 9 is a specific implementation of the half-bride resonant DC-DC converter of FIG.3(a) using a combination of MOSFET and IGBT switches.

FIG. 10 illustrates the voltage and current waveforms associated in operation with the circuit of FIG. 9. FIG. 11 is a specific implementation of the full-bridge resonant DC-DC convener of FIG.5(a)using a combination of MOSFET and IGBT switches, with the addition of a snubber diode.

FIG. 12 illustrates the voltage and current waveforms associated in operation with the circuit of FIG. 11. FIGS. 13(a) and 13(b) are circuit diagrams illustrating alternate implementations of the full-bridge resonant DC-DC converter of the present invention, wiLli a common ground for the input and the output, but without a high voltage switch.

FIG. 14(a), 14(b) and 14(c) arc a circuit diagrams illustrating a possible LLC converter circuit designs, that are (A) operated in a novel and innovative vay based on the methods of the present invention, and (B) redesigned also as described in this disclosure.

FIG. 15 illustrates a classic LLC circuit equivalent model used for First Harmonic Approximation (FHA) analysis. FIG. 16 illustrates the votlage gain, computed from FHA, achieved with an LLC circuit topology with various loads over a large range of switching frcquencies-

FIG. 17 illustrates the voltage gain, computed from FHA, achieved with an LLC circuii topology with various loads over a large range of switching frequencies with the conventional region of LLC operation and the Interrupt Switch Coni rol Region denoted.

FIG. 18 illustrates the voltage gain achieved with an LLC circuit topology w ith various loads over a large range of switching frequencies with a number of operating regions denoted: (1) RHS Operation region, (2) LHS Operation region, (3) Conventional Opearation region and (4) the LHS/ HS Boundary curve. FIG- 19 jllustrates a voltage gain graph of the LLC converter operated at constant gain with power flow regulated by adjusting the switching frequency.

FIG. 20 illustrates how conventional LLC resonant converters control their voltage boost, and therefore, their power flow.

FIG. 21 illustrates an LLC tank current waveform using an interrupt switch in accordance with another aspect of the present invention.

FIG. 22 illustrates an LLC tank current operating near lull power in accordarice with another aspect of the present invention. FIG. 23 illustrates an LLC tank current operating at low power in accordance with another aspect of the invention.

FIGS. 24(a), 24(b), 24(c) and 24(d) illustrate four representative implementati ons of the resonant DC-DC converter of the present invention with extern illy determined input and output voltages and no interrupt switch, where FIGS. 2 (b) and 24(c) further illustrate representative bi-polar output configurations and 1A (d) further illustrates a possible implementation with auto-balancing bi-polar out ut voltages.

FIG- 25 illustrates the voltage gain achieved with an LLC circuit topology vith various loads over a large range of switching frequencies with LI-IS/RHS Boundary curve denoted.

FIGS. 26(a) and 26(b) show the general form of the current invention with ;md without an interrupt switch.

FIG. 27 illustrates voltage gain curves for the LLC converter focused around a voltage gain of 4 in accordance with an embodiment.

FIG. 28 illustrates a final converter design in accordance with an embodime nt, with the region of operation identified.

In the drawings, embodiments of the invention are illustrated by way of example. It is to be expressly understood that the description and drawings are only for the purpose of illustration and as an aid to understanding, and are not intended a i a definition of the limits of the invention.

DETAILED DESCRIPTION

The present invention describes a number of innovations related to the subject matter of the Base Application. The present invention includes (A) a novel and innovative resonant DC-DC converter that employs a high boost resonant tank to enable power flow control between externally determined input and output voltages using frequency control, with or without use of an interrupt switch (I he "Improved DC-DC Converter"), (B) a method of operating a resonant DC-DC converter to achieve high boost resonant tank operation, which is suitable for improving the performance of resonant converters based on different topologies ("method of operation"), including but not limited to the Improved DC-PC Converter; and (C) a metliod for designing DC-DC converters (having different topologies) for improved performance using the method of operation ("design method"). The design method includes identification of circuit design parameters that enable use of the method of operation. Performance improvements include improved resolution of power flow control between externally determined input and output voltages a d maximization of range of allowable voltage conversion ratios while meeting a specified power flow. Operation of the converter ove> a reduced range of frequencies may also allow circuit components to be beuer optimized for efficiency. In full-bridge embodiments, as exemplified in FIG "4, the use of uni-polar/bi-polar operation may allow enhancement of circuit efficiency over portions of the operating range. In embodiments such as FIG 24(d) the provision of a bi-polar auto-balancing output voltage eliminates need for advanced sensing and controls to achieve voltage balancing in the bi-polar output.

For example, use of the embodiment of FIG 24(d) for solar photovoltaic applications, enables the design of a high voltage boost DC-DC Converter that offers inherent safety through low- voltage operation of the photovoltaic modulus, distributed control for increased energy yield, high-conversion efficiency ovei a wide range of input voltages and power flows, integrated galvanic isolation to isolate faults, use of long-life film capacitors, and full utilization of the AC gi id interconnection inverter.

There are related patent applications to the Base Patent including PCT/CA2011/000185, filed 02/18/2011 , claiming priority to United States Pate nt Application No. 03/18/2010 (the "Related Patents"). Certain details of the Related Patents are restated here to aid in understanding of the invention. More particularly the disclosure discusses DC-DC converters that include an interru t switch and also that do not include an interrupt switch, because aspects (B) and (C) of the invention arc relevant to both types of circuits. One aspect of the invention is a resonant converter circuit design operabl* to achieve high jnput-io-ouipul voltage conversion. In particular the invention ma include a series of converter circuit topologies that provide high resonant tank boost ratio and achieve high efficiency operation. The converter circuit topologies may include a resonant tank and (in one aspect) a means for interrupting die t mk current to produce a near zero-loss "hold" state wherein zero current and/or icro voltage switching is provided, while providing control over the amount of power transfer. Specifically the converter circuit topologies may control energy tran fer by controlling the duration of the near zero-loss "hold". This may be referred t> > as the "interrupt control mode" (again, shown for example in Figs. 19 and 21) This energy power transfer control may be achieved using a single high vol! ge controllable switch.

The present invention may avoid unnecessary circulating current during low power operation, thereby reducing losses within the tank components and the low voltage DC/ AC converter, and also reducing switching losses based on the iero voltage switching of the low voltage DC/AC converter and zero current switching of the low voltage DC/AC converter. Also, 2£ro current switching of the Mgh voltage controllable switch within the tank may be achieved and thereby keep its own switching losses low. As described herein, the present invention may have several embodiments ihat present con verier circuit topologies that provide high input-to-output vollage conversion and achieve high efficiency operation. Examples of tliese embodiments are disclosed herein; however a skilled reader will recognize Λ\ΆΙ these examples do not limit the scope of the present invention and that oi her embodiments of the present invention may also be possible.

For clarity, the term "low voltage" is used in this disclosure to refer to components with voltage ratings comparable to that of the input, and the t ;rm "high voltage" is used in this disclosure lo refer to components with voltage raiing comparable to, or above, the peak voltage level seen across the resonant unk capacitor. In embodiments of the present invention, appropriate implementation of the nea zero-loss hold state, may cause zero voltage switching or zero current switching to be achieved lor all controllable switches within the circuit.

Embodiments of the present invention may provide a lower loss convener circ uit for high input-to-output voltage conversion ratio converters.

A skilled reader will understand that the circuit design of the present invention may include a variety of elements. Jn one embodiment these elements may include: (1) an input DC/AC converter; (2) a resonant tank; (3) a Lank interruption means (such as a switch as described herein); and (4) an output rectifier. T e output rectifier may, for example, include a filter inductor that limits the rate of rise of current in the output diode. Regarding the input DC/AC, a skilled reader will recognize that a number of different types of inverters may be suitable, for example, such as a half-bridge or full-bridge type inverter. A skilled reader will further recognize that the output rectifier may include any output rectifier staj'.e, for example, such as a half-bridge or full-bridge rectifier. In some embodiments of the present invention, a transformer may be included in the circuit, prior to the output rectification stage.

In one embodiment of the present invention, the circuit design may be a circuit that includes: (1) a full-bridge DC/AC converter; (2) a resonant tank consisting of two L components and one C component; (3) a tank interruption switch; and ( 1) an output rectifier stage (full-bridge or half-bridge), wherein a common ground may be provided for both the input voltage and the output voltage. Possible embodiments of the present invention lhaL include such a circuit design are shovn in FIGS. 5a to 5d. The circuit may, or may not, include a transformer. In an embodiment of the present invention wherein a full-bridge output rectifier is utilized a transformer may also be required. In an embodiment of the present invention that includes a transformer, the resonant L components may be integrated into the transform er design. The choice to include a transformer in an embodiment of the present invention may be based on specifications of the circuit of the embodiment of the present invention, or other preferences or considerations. This document discloses and describes some examples of both: embodiments of the present invention (hat include a transformer clement; and embodiments of the present invention that do not include a transformer element, and therefore are transformerless. FIGS. 6(a), 6(b) and 6(c) show embodiments of the present invention that ire circuits 42, 44 and 46 respectively, that include an alternate implementation, wherein additional windings were added to the main inductor's magnetic core thus decreasing the voltage stress on switch S*. The addition of windings may convert the inductor L into a transformer with isolation, which provides additional circuit implementation options. The embodiment of the present invention shown in FIG. 6(c) may provide bipolar output to allow a differential output voltage of 2 x V₂ to be achieved while maintaining a voltage to ground at level V2.

As shown in FIG. 7, a circuit 48 may be one practical implemeniittion of ihe circuit shown in FIG. 6(c). The transformer magnetizing branch may provide I he main resonant tank inductance "L". Through appropriate transformer design, the filter inductance "Lr" may also be integrated into the transformer. This may be done by designing the transformer to have leakage inductance of value "Lr". Vs shown in FIG- 7, all switches may be implemented using MOSFETs. A snubber circuit may be employed to limit the transient voltage across the high voltage MOSFET at the end of the conduction period. Provided the voltage V₂ is low er than the voltage rating of the high voltage MOSFET, the snubber may consist o!^' a single diode from the drain of the MOSFET 10 the positive output V₂. This m.iy allow energy normally lost in snubber circuitry to be transferred to the output, thereby yielding a near lossless snubber. This may improve overall convener efficiency.

As shown in FIG. 8, embodiments of the present invention may produce particular results 50 that include gating signals for the converter of FIG- 7, together with tlie important voltage and current waveforms. The following is a description ol a possible switching cycle method: 1. At time 0.900 ms the cycle may begins with the turn of on of switches S i, Si_p and S_K. Thereafter energy may be transferred into the resonant tanls as seen by the positive voltage V_in and positive tank input current It.

1. When the tank current Ii reaches zero switches Si and S¾, may turn off, almost immediately after which switches St and Si_P may turn on. This may cause the input voltage polarity to become negative at the same time that the current becomes negative.

3. Switch S_x may turn off at the same time as Sj and S2p, though ihc MOS1FET body diode may allow conduction of the negative current. If losses in the MOSFET conduction channel are calculated to be lower than body diode conduction Josses, then the MOSFET should be kept on for ihe duration of the negative current pulse to reduce conduction losses.

4. When the current reaches zero the switch S* must be off. This may interrupt the tank current and allow the circuit to enter a near zero loss "hold state" where the converter operation is suspended and held in a n^ar lossless state.

5. The duration of the hold state may be varied to control the amount of average power transfer from input to output. Following the hold sl ile another similar cycle of operation may follow.

Transfer of power from the resonant tank to the output may occur twice per period, once to the positive DC output, once to the negative DC output. Pov er transfer to the positive output may take place immediately after the turn on of Switches Si and S¾,. Power transfer to the negative output may lake e immediately after the tum on of switches S₂ and S_lp.

In one embodiment of the present invendon, a circuit may be provided consisting of a DC-AC converter followed by a (parallel) resonant tank with single controllable high voltage switch, followed by an AC-DC converter.

IS Embodiments of the present invention that includes the proposed "half-briilge floating tank" resonant DC-DC converter configuration are shown in FIGS. 3' ), 3(b) and 3(c) in three specific representative implementations. The embodiment of the present invention shown in FIG. 3(a), may be a circuit 14 that does not include an output filter inductor. FIG. 3(a) illustrates the basic circuit design concept of the present invention, and presents a half-bridge floating lank convei ter in accordance with the present invention. The embodiment of the present invention shown in FIG. 3(b) may be a circuit 16 that includes an output filter inductor. For most implementations of the invention, it is a practical requirement to include a filter inductor. Generally speaking, there are two locations where it is convenient to add the filter inductor. The first is illustrated in FIG. 3(b). 'J he second is shown in FIG. 3(c), which shows an embodiment of the present invention that may be a circuit 18 that includes a filter inductor integrated in the tank. As shown in FIG.4(a), in one embodiment of the present invention the circuit 20 may be a "full-bridge floating tank" configuration of the circuit design illustrated in FIGS. 3(a), 3(b), and 3(c). FIG. 4(a) may be extension of the conve er illustrated in FIGS. 3(a), 3(b) and 3(c). A skilled reader will recognize that the circuit 20 shown in FIG. 4(a), relative to the circuits 22, 24, and 26 shown in FIGS. 5(a), .5(b), 5(c), and 5(d) respectively, for example, may lack a common ground on the input and the output and therefore may be undesirable for m;iny transformerless applications. In embodiments of the present invention an isolation transformer may be added between the capacitor and the diode rectifier, to allow grounding of both the input and output voltage sources. Embodiments of the present invention, as shown in FIGS. 5(a), 5(b), 5(c) and 5(d), may represent variants of the full-bridge resonant DC-DC converter of the present invention, and may include a single high voltage switch, and a common ground for the input and the output More specifically: the embodiment of the present invention shown in FIG. 5(a), may be a circuit 22 wherein the indue tor current may be switched by the single high voltage switch (S^); the embodiment of the present invention shown in FIG. 5(b), may be a circuit 24 wherein the capacitor current may be switched by the single high voltage switch (S_x); the embodiment of the present invention shown in FIG. 5(c), may be a circuit 26 t at is similar to the circuit 22 shown in PIG. 5(a), and the circuit 26 shown in FfG. 5(c) may include an inductor current that may be switched by S_x and the fi lter inductor may be integrated into the tank; and the embodiment of the present invention shown in FIG. 5(d), may be similar a circuit 28 that is similar to the circuit 24 shown in FIG.5(b), and the circuit 28 shown in FIG. 5(d) may include a capacitor current that may be switched by S_* and the filter inductor may be integrated into the tank. It should be understood that the DC-DC converter of the present invention as shown in FIGS. 5(a), 5(b), 5(c) and 5(d), relative to prior art full-bridge extensions of half-bridge circuits, may display a significant degree of asymmetry. In particular the asymmetry may be displayed in that the grounding is asymmci tic, the input switch configuration is asymmetric, and the output stage is asymmetric. A skilled reader will recognize that other variants and embodiment of the present invention are possible. For example an embodiment of the present invention may use emerging reverse block IGBT devices, in which case S_x may be eliminated, but Si and S2 may each need to consist of a high voltage reverse blocking IGST. Such an embodiment of the present invention may yield precisely the sii e voltage and current waveforms within the tank and output circuitry. Numei ous other variations are possible.

In an embodiment of the present invention, the circuit design may be such thai the high voltage switch needs not be reverse blocking, and thus MOSFETs or IGliTs may be used instead of, for example, thrysitors (which limit Switching frequencies to excessively low values), or MOSFET-series-diode / IGBT-series-diode combiuations-

Also, in embodiments of the present invention, the circuit designs may usi an electrically floating tank, as further explained below. Certain aspects of the invention are explained in greater detail below, however these details should not be read as limiting the scope of the invention in anyw iy, but as examples of embodiments of the present invention.

The Half-Bridge Floating Tank Converter

The half-bridge floating tank converter may be included in embodiments of ihe present invention. In such an embodiment of the present invention, the switching process may vary slightly based on the type of switches used and ihe location/orientation of the high voltage switch (S_x) within the tank circuit. A description of a possible switching process to be used in an embodiment of ihe present invention is provided herein with reference to a topology 30 wherein $| and S2 are implemented using MOSFETS and S_x is implemented using a high voltage IGBT, as shown in FIG. 9.

In one embodiment of the present invention, as shown in FIG- 10, waveform results 32 of use of the embodiment may show particular voltage and current waveforms associated with a half-bridge floating tank converter. For example, ihe converter may operate in a mode where the inductor current is not continuously oscillating but is interrupted, once each period, by the single high voltage switch,

An example of the operation of the circuit may be as follows:

1. Si and S_x may fire to begin one cycle of LC resonant oscillation. For ihe given orientation of the IGBT (S»), the initial condition on the capaciior voltage may be approximately -V2.

2. Current II may be positive and input voltage V_jn may be positive for haif a cycle, transferring energy into the circuit.

3- Once V_Cg reaches V₂, the output diode conductors and II may be transferred to the output, accomplishing output power transfer (the ra| d rale of rise of the output current may be reduced through introduction of an additional current-rate-of-changc limiting inductor placed either in series with the output diode or the tank capacitor). . At zero crossing of the input current Si may be turned off and S2 may be turned on. The output diode may turn off at this time and the IGBT reverse conducting diode may turn on at this time. This allows the tank oscillation to continue, thereby recharging the capacitor to -V2, in preparation for he next cycle.

5. When the current II attempts again to go positive, the IGBT may be in an "off' iBtate, thus interrupting the tank oscillation at a current zero crossin;;. 6. The circuit may then in a 'hold state' until a new pulse of energy is required.

The Full -bridge Floating Tank Converter with Common Ground

Embodiments of the present invention may include a full-bridge floating tank converter with common ground, as shown in FIG. 1 1. In such embodiments of ihc present invention the switching process may vary slightly based on the type of switches used and the location/orientation of the high voltage switch (S_x) within the tank circuit. One embodiment of the present invention include a fulJ-brid¾e floating tank converter with common ground may include a topology 34 where ine four switches Sj, S]_P, ¾ and S_2p are implemented using MOSFETS and is implemented using a high voltage IGBT, as shown in FIG. 11. In an embodiment of the present invention that includes a full-bridge floating tank converter wjih common ground, a snubber circuit may be employed to limit the transient voltage across the high voltage MOSFET at the end of the conduction period. The snubber may consist of a single diode from the collector of the IGBT to the output. This may allow energy normally lost in snubber circuitry to be transferred to tlie output, thereby yielding a near lossless snubber. Such embodiments of the present invention may improve overall converter efficiency.

In one embodiment of the present invention, as shown in FIG. 12, wavefoi m results 36 of use of the embodiment may show particular voltage and current waveforms associated with this a full-bridge floating tank converter with common ground. The converter may operate in a mode where the inductor current js not continuously oscillating but is interrupted, once each period, by the single high voltage switch, S_x. An example of the operation of the circuit may be as follows:

1. For the given orientation of the IGBT (S_x), Si, Si_p and S_x may fire to be nn one cycle of LC resonant oscillation.

2. Current Ii may be positive and input voltage V_in may be positi ve for hal f a cycle, transferring energy into the circuit. 3. When Ii crosses zero Si, Si_p may turn off and i and Si_p may turn un.

Sometime during negative Ii the switch S* may be turned off lossles ily since the current is flowing in the anti-parallel diode,

4. When V-_K reaches power may begin being transferred to the output.

This may continue until the curreavt Vj decays to zero. 5. Capacitor voltage may then be in a 'hold state' until a new pulse of cnei¾y is required.

The Full-bridge Converter with Common Ground and Silicon Carbide Devices

Embodiments of the present invention may include a full-bridge floating tank converter with common ground that is operable to transfer energy during both positive and negative half cycles of the tank current, without use of a transformer, while maintaining a common ground on input and output, as required for many applications. The purpose of S in this circuit may be to achieve zero current/zero voltage switching while still offering control over the amount of power transfer. Thus near z ro switching loss may be achieved while simultaneously maintaining control over the amount of power transfer.

As silicon caibide switching devices, or other devices with low reverse recovery loss, become more cost effective it may become worthwhile to eliminate Nonetheless, a common ground arrangement capable of transferring energy during both positive and negative half cycles of the tank current may still be desired. The circuit topologies 38 and 40 of FIGS. 13(a) and 13(b) accomplish tins. These topologies may be related to the circuit designs shown in FIGS. 5a and 5c. As silicon carbide devices may offer grcady reduced switching losses (esp. the elimination of diode reverse recovery current), maintaining zero current zsro voltage switching may be sacrificed without negatively impacting efficiency. Power transfer may then be achieved via frequency control, as is common in ot er resonant converters, see: R. Erickson, D. Maksimovic, "Fundamentals of Po^ver Electronics," Kluwer Academic Publishers, 2001.

The full-bridge converter with common ground may offers important benefits compared to the conventional resonant converters as outlined in R. Erickson, D. Maksimovic, "Fundamentals of Power Electronics," Kluwer Academic Publishers, 2001. Specifically the topology of an embodiment of the present invention ihat includes a full-bridge converter with common ground may offer common ground on input and output along with a high step-up ratio and may offer power Iran -fer into the tank during both positive and negative half cycles of the tank current.

As examples of embodiments of the present invention and die benefits that these offer over the prior art, benefits of particular features of two principal c ruit arrangements (a half-bridge floating tank converter, and a full-bridge floating t.ink converter with common ground) over the prior art are described below. A skilled reader will recognize that the features and benefits discussed below are mei ely provided as examples, and other embodiments and benefits are also possible.

Benefits of The Half-Bridpe Floating Tank Converter: Embodiments of che present invention that include a half-bridge floating l ink converter may offer particular benefits over the prior art. Some of these bene fits include the following:

1. In comparison to the circuit of A. Abbas, P. Lehn, "Power electn nic circuits for high voltage DC to DC converters," University of Toronto, Invention disclosure RIS#10001913, 2009-03-31, or that of D. Jovdc, "Step-up MW DC-DC converter for MW size applications," Institute of Engineering Technology, paper IET-2009-407, the half-bridge circuits of the present invention may only use one high voltage device, labelled: S_x. Furthermore S may not need to be a reverse blocking device.

2. A single high voltage switch may be operable in embodiments of the present invention to interrupt the resonant operation of the conveicer, thereby controlling energy transfer.

3. Si and Si may be implemented in embodiments of the present invent ion using, only low voltage components, reducing losses.

4. In comparison to the invention of B. Buti, P. Bartal, I. Nagy, "Resonant boost, converter operating above its resonant frequency," EPE, Dresden, 2005, embodiments of the present invention may only require a single source and sing/e tank inductor.

5. Embodiments of the present invention may provide zero cmrentAtero voltage switching of the input AC-DC converter.

Benefits of The Full-bridge Roatiug Tank Converter with Common Ground

Embodiments of the present invention that include a full-bridge floating tink converter with common ground may offer particular benefits over the prior irt. Some of these benefits include the following:

1. In comparison to the circuit of A. Abbas, P. Lebn, "Power electronic circuits for high voltage DC to DC converters," University of Toronto, Invention disclosure RIS#10001913, 2009-03-31, or that of D. Jov ;ic, "Step-up MW DC-DC converter for MW size applications," Institute of Engineering Technology, paper ΓΕΤ-2009-407, the circuit of embodimf nts of the present invenLipn may operate using only one high voltage device, labeled S*. as shown in FIGS. 3(a), 3(b). 3(c), and 3(d). Furthermore S_K may not need to be a reverse blocking device. . In comparison to the circuit of P. Lehn, "A low switch-count resonant dc/d converter circuit for high inpuHo-oulput voltage conversion ratios," University of Toronto, Invention disclosure RTS#10001968, 2009-08- 13, or the half-bridge circuit of the present invention, the full-bridge DC-DC converter of embodiments of the present invention may provide rou ldy double power transfer since energy may be transferred from the souice into the tank during both positive and negative half cycles of the tiink current. . Embodiments of the present invention may provide zero currcnt ζι ο voltage switching of the input AC-DC converter.

4. In embodiments of the present invention common ground may be provided between the input voltage source and output voltage source.

5. In embodiments of the present invention a single high voltage switch may be operable to interrupt the resonant operation of the converter, then by controlling energy transfer.

A skilled reader will recognize that numerous implementations of the technology of the present invention are possible. The circuit designs of embodiments of ihe present invention may present a modular structure and therefore components m y be added or removed, while providing the Functionality of the design, as described above For example, particular embodiments of the DC-DC converter of >he present invention may be transformerless. In other embodiments of the present invention it may be desirable to include a transformer in the circuit, such as ^he circuit shown in FIG- 4(b). For example, a transformer could be included betwe en either the resonant tank inductor or resonant tank capacitor and the diode rectif ier in the circuit shown in FIG. 4(b). Also, while use of S_x is described for some embodiments of the present invention, this component may be eliminated by, Or example, usiag emerging reverse block IG¾T devices, where Si and S_?r would each need to consist of a high voltage reverse blocking IGBT. Variants

A skilled reader will recognize that in embodiments of the present invention specific aspects of the topologies described and shown herein may be modified, without departing from the essence, essential elements and essential functions of the topologies. For example, in the circuit design 42 shown in FIG. 6(h), if L_f iind C are in series with no midpoint, it may be possible to swap Lr and C. Similarly, when used with a transformer, any number of known output winding and rectijier configurations may be applied to achieve the same objective.

In one embodiment of the present invention the switching elements, for example as shown in the FIG. 13(b) may employ silicon carbide devices. Switching may be carried out to provide a square wave voltage switching between +V1 and -VI lo the tank circuit. The switching carried out to provide a square wave voltage may be switching between +V1 and 0 (or between 0 and -VI) to the lank circuit. Tank input voltage switching may occur between +V1 and -VI when operating near rated power and between +V1 and 0 (or between 0 and -VI) under low power. Alternatively, the elements recited in this paragraph may be used in a topology where the inductor Lf is moved to the output path (such as is shown in FIG. 13a).

Voltape Boost Resonant Tank Converter

The inventors have realized DC-DC converters may be provided that include improved performance characteristics of the DC-DC converters disclosed above, however, without the interrupt switch disclosed in the Base Patent.

More particularly, in another aspect of the present invention, it has been realized by the inventors that it is also possible to build a desired resonant DC-EC converter for providing a high voltage step-up ratio without employing a tank interruption switch S_x as exemplified by the circuit topologies 38 and 40 of FlGS- 13(a) and 13(b). More generally, this is achieved by employing a number of concepts including: (i) achieving a high boost through the systematic design oi a resonant tank; (ii) enhancing converter efficiency using a unipolar/bipolar resonant tank excitation; and (iii) employing an output configuration with automatic voltage balancing on output capacitors in conjunction with the high boosting resonant tank circuit to yield a high step-up ratio and a balanced bipolar DC output voltage.

More specifically, it is possible to achieve a high boost ratio from the reson.mt tank through the careful selection of resonant components. An illustrative example is shown in FIG. 24(a) where an LLC converter circuit design in accordance with an embodiment does not require a high voltage switch. Here, C, represents a resonant capacitor, Lr represents a resonant inductor, and Lm represents a magnetizing inductor. The "Classic" LLC Circuit DC-DC converter topologies shown in FIG. 14 have been studied in literature (R.L. Lin et. al. and H. Hu et. al., above), however most of the prior ait is related to step down (buck) realizations of the technology. Tlus type of converter design is commonly used in conventional applications where t tie output voltage is independent of the load, such as a power supply. In thi se applications, the classic LLC circuit topology offers advantages compared to otl icr circuit topologies. To illustrate the functioning of the circuit, the voltage g;iin characteristics of the LLC converter can be approximated using first harmonic approximation (FHA) techniques. Assuming the circuit is stimulated by a perfect sinusoid, one can use conventional circuit analysis to determine the voltage gain of the circuit.

An LLC converter as illustrated in FIG. 14(a) for example may be simplified to provide the circuit shown in FIG. 15 here Re is the equivalent resistance for tlie resonant tank. This equivalent resistance depends on the type of rectifier use d. For the full bridge this may be Re= 8R/7T [discussed for example in H. Huar g, above.] where R is the DC load resistance across the full bridge rectifier output.

It can be shown that the voltage gain of the circuit is defined by;

Where

The voltage gain can be then calculated for different loadings and frequencief to produce the plots shown in FIG. 16- This figure shows the voltage gain achieved by an "LLC Resonant Tank", as a function of normalized switching frequency of the input stage DC-AC converler.

The different lines are plots at different loading conditions (constant R), or staled alternatively, at different Q values as determined by Equation (4) below. As sf en by .he equation, as die load decreases (R decreases), the Q value is reduced in an inverse proportional relationship. ).u FIG. 16 the darker lines represent low Q values, and trie lighter lines represent high Q values.

(4)

The resonant frequencies of the circuit are defined by f_ri and fa, defined below:

(6)

1

In conventional applications, such as power supplies, a Classic LLC Circuit is generally operated near f_rI as indicated in FIG. 17 by the box titled, "Conventional Region of Operation", because a constant output voltage is desired throughout tlie entire load range. The desired ratio between the input voltage and output voltage is predominandy achieved using a transformer in the output stage, and not the LLC Resonant Tank itself. When the input voltage changes the output voltage is maintained at a constant level by adjusting the switching frequency of the in nut stage above or below f,i- The value of Q may not critical to the operation of ihe circuit and it may only be verified that the circuit can provide the required out ut voltage for the maximum load- Values of Q close or even higher than 1 -ire common in conventional circuits.

It has not been obvious to a person skilled in the art that the Classic LLC Circuit topology can be operated over a frequency range well below f_r] (f_rj is not within the operating range) by selecting the components such that the value of Q is well below 1 for the full load range specified. Furthermore, the circuit has not been used in applications that require control of the power transfer between two regulated or unregulated DC sources.

In one embodiment of the present invention, the LLC topology is designed to operate with switching frequencies well below f_rj, close to the second resonant frequency of the circuit, fr_f). Operation in the area near f,o can be divided into two distinct operating regions as shown in FIG. IS- As shown in the figure, the two regions are named the "LHS Operation" and "RHS Operation" regions. The line which intersects both of these regions is called the "LHS RHS Boundary", which is also shown in Fl.G. 18. Operation in the "LHS Operation" region yields zero current switching (ZCS), suitable for switching devices such as IGBTs. Operati m in the "RHS Operation" region yields zero voltage switching (ZVS), suitable ior switching devices such as MOSFETs. Operating in any one of these regions yields a voltage gain above I for loads with Q lower than 1. The values of tne resonant tank components can be selected such that the Q value lower than 1 cm be achieved for all load values (power transfers) required to be handled by the converter. This Q value would be lower for higher voltage boosting requiremenis. The system would then operate at a switching frequency below t^'-i for all steai ly state operating conditions. A resonant tank circuit designed in accordance wirh this embodiment will be called a "High Voltage Boost Circgir (HVBC). All >f the embodiments of the invention shown above use a HVBC resonant tank circuit design. The introduction of a transformer to the circuit does not alter the high boost nature of the tank design.

For a specific application, che range of input voltage and the range of load (po-,ver transfer) is known. The output voltage is also known based on components to be powered by the converter or the externally regulated voltage bus that is to receive power. In one aspect of the invention what follows is a possible method for designing circuits based on said LLC topology, but providing relatively high boost ratios-.

1) Choose an Ln L_t ratio that is suitable for the application. Typical values range from, but are not limited to, 3-10. Large values will result in higher pi ak currents in the tank, while small values will result in larger switching losses at low loads.

2) Generate voltage gain curves for various values of Q. On that plot, ihe boundary curve separating LHS and RHS regions may be graphed in a similar manner to that shown in FIG. 1 S-

3) From the plot, select the Q value whose voltage gain curve intersects ith the boundar curve at the desired voltage boost ratio. Note the Q value and normalised frequency (f«) of this intersection point

4) Using the Q and normalised frequency values found in step 3, calculate ihe I-_T and C_r values.

5) Using the Lr value calculated above and the desired L,„/Lr ratio, calculate l

Power Flow Control and Strategics for the LLC Boosting Converter

In an aspect, the first method discovered to achieve controllability of the above design was the introduction of an interrupt switch in the LLC Resonant Tank (tUe "Interrupt Switch LLC Circuit"). The interrupt switch allows the Q value to i>e solely dependent on the input voltage and not the load. As the input volta ;e increases, the Q value decreases. The Input Stage switching frequency of t ie circuit is used to compensate for changes in the input voltage and the off time of the interrupt switch is used to adjust to the changes in load. The decoupling of the load (using the interrupt switch in the LLC Resonant Tank) from the input volfcige (using the Input Stage switching frequency) allows for a simple implementatioti of a controller and stable control.

As disclosed in earlier described embodiments, the introduction of an interrupt switch into the LLC Resonant Tank also enables the use of the Interrupt Swkch LLC Circuit in new applications where the LLC Resonant Tank is operated in ihe conventional region of operation close to f The use of the Classic LLC Resonant Circuit in this operating region is not easily realizable with the classic frequency control method. In other words, the Interrupt Switch LLC Circuii is suited to new applications where the objective of the LLC circuit is not to regul ite the output voltage but instead to regulate the power delivered to an output voltage regulated externally.

Those skilled in the art will understand that prior to the present invention, DC-DC converters of the type described in this disclosure would be operated in ihe "Conventional Region of Operation" shown in FIG. 18. However, when an LI .C resonant tank is operated in the conventional region of operation, depicted in FIG. 18, the various load curves begin to converge within this region. As ihe separation of the load curves becomes smaller, large load power trans ' er variations begin to occur, even for minute variations in the frequency. This makes power flow control impractical as the gain drops to near unity, since excessive ly fine frequency resolution is needed to achieve acceptable resolution in load pow er flow control. Even using a high performance frequency controller, load power control still becomes theoretically unachievable at unity gain. This is the reason why an interrupt switch is proposed in the Related Patents. The inventors discovered that when operating the LLC resonant tank with a minimum boosting having an effective value above unity over the entire operating range, it was unnecessary to decouple the load from the input voltage using the interrupt switch referred to above.

The inventors discovered that if Ihe LLC resonant tank is operated so as to l ie given sufficient boosting gain, a change in either the input voltage or the switching frequency results in a corresponding change in load (power transfer). This is illustrated in FIG. 1 for one possible circuit design that is adapted to deliver minimum boosting as described.

In particular, FIG. 19 illustrates maintenance of a fixed voltage gain of 2.0 wl iiie using frequency control to enable movement between the load curves. As shown in FIG. 18 for example in the "frequency control regions" there is horizontal separation amongst the load curves. Operation of the LLC Resonant Tank on a maintained basis in the frequency control regions suitably above unity g iin enables better control of power transfer based on switching frequency, while maintaining the boosting ratio shown in FIG. 19.

This provides the reduced frequency range of operation required to control :he load, and chopping of much smaller currents than conventional non-boosting D circuits.

A skilled reader will appreciate that components of a DC-DC converter desigr ed to embody the mode of operation described may be selected so as to improve performance within the frequency range described.

Therefore, the objective of the design method of the present invention is to provide a DC-DC converter that is designed so that the boosting gain is above unity. Theoretically, the boosting gain can be designed as close to unity as desiied provided a frequency controller with an infinite frequency resolution. Practic al implementations of the converter which use frequency controllers with a Finite resolution will require a minimum boosting gain above unity which achieves tlie desired controllability, i.e., the desired power flow resolution. For example, usi ig currently available microcontroller hardware and a resonant frequency in tlie range of 50kHz to 100kHz, a boosting gain of 1.25 may be practical to mainta in power flow controllability with practical power flow resolution over tlie entire operating range. A skilled reader will appreciate that this "minimum boosting gain above unity" will vary depending for example on the particular components selected, or that arc available on an economic basis. Also, this will vary with further technical or manufacturing developments in regards to such component. Through use of au appropriate design methodology, as will be described later, it is possible to transfer any desired amount load power via frequency control by appropriate selection of circuit parameters.

Detailed High Voltage Boost Circuit (HVBC) Operation The operation of the HVBC will now be described in more detail. As discussed, the HVBC is operated in a unique mode of operation. FIG. 18 shows the typical voltage ga that can be achieved with an LLC circuit. The different lines in FtG. 18 represent the same tank circuit with different loads. The lines then trace nit the voltage gain from the converter when operated from about 0.4 times ihc resonant frequency f_r] to 1.2 times the resonant frequency f,i .

Conventionally, LLC power supplies are designed to operate near the reson tnt frequency defined by the resonant inductor and resonant capacitor, f_rl. This region of operation can be seen in FIG. 17 with the resonant frequency f_ri denot d. When operated in this region near f,j, the circuit will exhibit constant voltage gain throughout the entire load range. FIG. 17 also shows the Interrupt Switch Control region, which covers parts of the conventional region of operation.

In the present HVBC embodiment, the LLC is designed such that it is operating very close to the resonant frequency determined by the resonant inductor, magnetizing inductor and the resonant capacitor, which will be referred to as In FIG. 18, this operating region is outlined and labelled "LHS Operation" a id " HS Operation". In these regions of operation, the circuit is able to achieve hi_. ;h boost ratios yet also achieve a reduced switching loss throughout a wide lo id range. Output power is controlled by varying the switching frequency, whi h need only be varied by about 20% of the resonant frequency. It will also be appreciated that the regions of operation as defined by FIGS. 17, I S and 19 are for demonstration purposes only, and as such, they are not fixed co those values depicted in the figures. One of the defining characteristics of the present invention is that the resonant tank is designed and optimized such that it can provide a voltage boost when stimulated with an AC voltage whose frequent y is less than f_ri , or less than a normalised frequency of f„ = 1, as shown in FIG . 1

17, 18 and 19. In FIG. 18, the borders of the "Conventional Region of

Operation", "RHS Operation" and "LHS Operation" regions are not fixed, ext ept

for the border between the "RHS Operation" and "LHS Operation" region. 1 his

line is defined by the points where the resonant tank appears as a resistor to the

AC stimulator, as described in the illustrative design example. The "Conventional

-Region of Operation" is focused around f_r), or a normalized frequency of f„ = ^'} in

FIG. 18. The "LHS Operation" and "RHS Operation" regions are focused around

¾o, or a normalized frequency of about f_n = 0.45 in PIG, 18. This normalized va lue

will be dift^'ereut for every unique resonant tank design. In the same way the

borders of FIG. 18 are not fixed, the regional borders of FIG. 17 are also not fixed,

and are only drawn this way for demonstration purposes.

PIG- 20 shows the current waveform flowing out of the switching network in a

conventional LLC circuit. Due to operation at f_r the switching network must

switch a significant magnetizing current as compared to the peak current. FIG. 21

shows the waveforms associated with an embodiment of the circuit using an

interrupt switch. The interrupt switch waits until negligible current is flowing in

the switch, and opens the switch at near zero current. This effectively means the

circuit is operating approximately on the ZCS ZVS boundary shown in FIG. 18. at

the boundary between the "LHS Operation" and "RHS Operation" regions. Pov er

is controlled by incroducing a "hold" state as shown in FIG. 21. For full pow er

operation, the hold state would be reduced to zero.

Now referring to FIG. 22, shown is a proposed mode of control over the prcsenily

described HVBC embodiment. At full power, the waveform will resemble the full power waveform of the circuit with the interrupt switch, with switching happening

at or very near the zero crossing of the current. Power reduction, however, is

achieved not by introducing a hold state, but rather by slightly increasing the

switching frequency of the converter as shown in FIG. 23. Switching action ne w

occurs somewhat prior to the zero crossing; however the currents at the time of

switching are very small, which can be seen in FIG. 23. Due to the low Q

operation over the entire load range only small variations in the switching

frequency are necessary to tegulale power from full load to zero load. In FIGS. 22 and 23 for example, the switching frequency is increased from 55-5 kffc to 59.7 kHz and the power transfer is reduced by about 25%.

The following is a description of a possible switching cycle method for an embodiment of the present invention utilizing a full-bridge DC-AC inverter and a split output circuit, operating in the "RHS Operation" region. The circuit is shown in FIG 24(b) without transformer, 24(c) wiih transformer and 24(d) w ith transformer and one possible implementation with auto-balancing output voltage. The waveforms are shown in FIG. 22 for full load and FIG. 23 for partial load:

1. At the beginning of a cycle, a positive charge exists on the capacitor. The switching cycle begins with the turn of on of switches Si and §2_Ρ. T he current in the resonant tank may be less than or equal to zero at t'lis moment. Thereafter energy may be transferred into the resonant tank *>nd because there is enough voltage to forward bias the output rectifier dio le, current is injected to the load.

2. The resonance will reduce the voltage in C_r and will increase the current in the inductor Lr. Lm has a constant voltage equal to V_o] across it. T he voltage across C_r will turn negative and the current across Lr will st irt decreasing.

3. When the current across Lr equals the current across Lm, the output rectifier slops conducting and no current is transferred to the load. At this point, Lm is included in the resonance and the same current flows through Lr and Lm.

4. Switches Si and≤ _Ρ are then turned off; almost immediately thereafier switches S2 and Si_p are turned on. This commences the second half eye le which is symmetrical to the first.

5. The length of the switching period may be varied to control the power flow through the converter. Although the above control descriptions are based on the circuit using a rull bridge DC-AC converter and the split output circuit, a person skilled in the art could be able to identify that the general operation- is similar in other embodiments. Differences in the number of pulses transferred per period, ihe type of load receiving the power pulses, or the location of the components used to produce the resonance amongst others do not change the operation principles for the circuit.

The benefits of the circuit over the classical LLC converter control are: (i i a significantly longer switching period (approximately 2 times) for a given set of components; (ii) a reduction in switching losses; (iii) a reduction in losses within the resonant tank (comprised of C_r, L- and L_ra); and (iv) the ability to regul ite power transfer between two externally determined DC sources.

Unipolar/Bipolar Resonani Tank Excitation Control

As described earlier, switching of the DC/AC converter may be carried out such that the DC/AC converter output is either an AC waveform of +V1 and -VI, or an AC waveform of either VI and 0 or -VI and 0. The ability to switch betwe en these modes of operation will be called "Unipolar Bipolar Resonant Tank Excitation Control". Unipolar/Bipolar Resonani Tank Excitation Control changes how the resonant rank is excited in order to operate the converter in its must efficient control mode for a given input power.

Bi-Polar Output

As shown in FIG. 24(b) and FlG. 24(c), an embodiment of the invention includes a bi-polar output voltage. This configuration is advantageous since the maximi m voltage to neutral is reduced by a factor of two. As a consequence, cabling with a lower insolation class can be used, reducing the cost of wiring the converter. Tlie use of two voltage sources to create the bi-polar output ensures that tlie output of the converter i always balanced to the neutral point. Auto Balancing Output

As shown in FIG. 6(c) and FlG. 7, an embodiment of the invention includes a voltage doubling rectifier, which creates a bi-polar output. This bi-polar output must be balanced in order to properly maintain the output DC link. Dae to the boosting namre of the converter, the output capacitors, Co in FIG. 24(d), ire automatically balanced. When one of the capacitors has a lower voltage than i.he other, the operating point of the converter moves vertically down the curves shown in FIG. 18. Moving down these curves corresponds to a higher Q value, or larger load. A larger load means more power will be transferred, which will in turn charge the capacitor back to its nominal operating voltage. No other control circuitry is needed.

In summary, the focus of the present embodiment is on a unique mode of operation that yields a large voltage boost in the resonant tank. This voltage bo )st allows the present HVBC embodiment to achieve very high efficiencies at high conversion ratios. With the present HVBC design, the resonant tank of an LI ,C converter can be designed lo yield high voltage gain, useful for step up convert? rs. As well, the converter can be operated with a low Q over the entire load ran;;e. This is achieved by knowing the load, and designing the resonant components around it. Furthermore, the resonant tank can be stimulated near the reson.mt frequency and operation of the converter in this region yields to ZVS, and lo current switching (LCS), to yield a highly efficient, step up converter. This mode of operation makes is viable for the converter to transfer power between two externally determined voltage sources.

Comparative Analysis of Interrupt Switch Control vs. Frequency Control lor Boosting LLC Tank Circuits

As noted above, both interrupt switch control and frequency control may be us ;d for boosting LLC Tank Circuits. This analysis focuses on the application of tlie interrupt switch concept to LLC converter applications and compares it to frequency control of the LLC converter. Resonant converters are designed lo transfer power from an input source to an output load. The output voltage divide by the input voltage is referred to as the gain of the converter. The theoretical gain of the LLC converter can be approximated using first harmonic approximation (FHA) techniques. It is tiien analyzed using the simplified approximate circuit shown in FIG. 15. where R<; = 8R 7C² [See H. Huang, "LLC Resonant Half Bridge Converter", Texas Instrume nts Presentation from Asia Tech-day, August 27, 2009.] and R is the DC lnad resistance across a conventional full bridge output rectifier.

In many applications we wish 10 supply a constant output voltage, V_D, from a given input voltage source, approximated by V_g. Based on the simplified model, the amount of current, I_m, flowing in Lm will be constant for a given V₀. In contrast, the amount of current flowing in the load, I_c, will depend on the l ad resistance R_e.

The current, I_r, seen by the input ac source, the capacitor C_r and the inductor L- therefore has two components:

(i) the component I_m, set by the desired V₀; and

(ii) the component I_», set by die loading.

The current I_M itself transfers no power to the load, it is merely required to enable the process of energy transfer. At higher load I_c comprises a large percentage of I_r, leading to highly efficii ut operation.

Using frequency control, lighter loading conditions result in I_m comprising a larger percentage of L. Since numerous losses are related the amplitude of L, efficiency will suffer at light load conditions. Particularly at power levels beU 15% of rated power, the efficiency typically becomes very poor.

The interrupt switch enables a high Ic lo I_r ratio to be employed under all loading conditions. At full load the I_e to l_r ratio is high by its very nature, posing no challenge. To operate at reduced load the interrupt switch introduces a near zero loss hold stale. This yields an efficiency that is roughly independent of loadmg conditions- It should also be noted that each time the convert leaves the hold stile one pulse of energy is transferred to the output. For a given input and output voltage the size of this energy pulse is constant. Power transfer is controlled by merely regulating the number of energy pulses that are released by the intemipt switch.

FIG. 22 shows a comparison of where the interrupt circuit operates versus wtn:re the frequency contiol circuit operates for a Fixed V₆ to V₀ ratio of 1:2. Note Llial only one point is shown for the interrupt circuit operation. The interrupt switch pulses the power to the output always at one point on this plane. By conuolling the pulse density the amount of power transfer is linearly controlled.

Under frequency control wc operate along a horizontal line, moving to higher frequencies to decrease power. The amount of power transfer varies nonlineaily with the operating frequency. A clear negative impact of employing the interrupt switch is that this device adds additional conduction losses to the resonant tank circuit.

This leads to a trade-off between low power and high power efficiency as follow s:

• A converter that operates predominantly at a small percentage of its r i id power will benefit from the interrupt switch, since efficiency is held hi¾h even at low power transfer through the interrupt process.

• A converter that operates predominantly at a large percentage of its rat>:d power will benefit from elimination of the interrupt switch, since efficiency of the converter is already high due to the large power transfer. Elimination of the interrupt switch conduction loss can be beneficial. Benefits of Interrupt Switch Control

The following is a list of benefits of the interrupt switch:

• High efficiency at low power as noted above. • The power transfer between two fixed voltage sources is proportional to the time interval between interrupt switch turn-on events. This enables simpJe control of the circuit.

• The power transfer between to the output is easily controllable even under lower boost ratios.

• Use of the interrupt switch reduces switching losses in the input DC/AC converter that is supplied by V_E by ensuring soft-switching.

Drawbacks of Interrupt Switch Control

The following is a list of drawbacks of the interrupt switch; · Addition of switch conduction loss to the tank circuit, reducing high power efficiency.

• Component cost.

Benefits of Frequency Control

The following are benefits of the using frequency control in place of interrupt control in an LLC converter:

• Efficiency at high power can be enhanced through elimination of

conduction losses associated with interrupt switch.

• Reduc tion in component cost, due to elimination of interrupt switch.

• Reduction in both input nd output DC filter size. Drawbacks of Frequency Control

The following are drawbacks of the using frequency control in place of interrupt control in an LLC converter;

• Low efficiency at li ght 1 oads .

• Highly nonlinear power transfer equation leading to more challenging controller design.

• Control challenges in regulating power flow between two fixed voltage sources when the boost ratio is low. Application Examples of the Classic LLC Circuit operating in the Novel Region of Operation:

• Using an operating range on the right hand side of the peak may be implemented with MOSPETs, because these switches have favorable performance when operated with zero voltage switching ("RHS Operation" as illustrated in FIG. 18).

• Using an operating range on the left hand side of the peak may be implemented with IGBTs, because these switches have favorable performance when operated with zero current switching ("LHS Operation" as illustrated in FIG. 18)-

• RHS Operation for use in low voltage applications.

• LHS Operation for use in high voltage applications.

• Such applications include, but are not limited to, solar photovoltaic systems, fuel cells, permanent magnet wind turbines, electric and hybrid vehicles, electric charging stations, aerospace applications, marine applications, mici o- grids, energy storage and other systems that require converters with varying input voltage and load.

Application Examples of the Interrupt Switch LLC Circuit operating in the o^ el Region of Operation: The interrupt switch topology is used in two main applications: 1. In applications where a high efficiency is desired and the converter operates at low power for long periods of times, such as standby pcwer applications.

2. In low boosting applications where the power flow between two voliage sources needs to be controlled, including but not limited to, i) residential application of solar photovoltaic systems (including module level optimizers and micro-inverter), fuel cells, permanent magnet wind turbines, micro-grids and energy storage; ii) small power marine and aerospace applications (low voltage); and iii) and other systems that require converters with varying input voltage and load at low input and output voltages.

Illustrative Design Example

This design example illustrates how the selection of appropriate components i an LLC converter can yield the desired low Q operation, A brief overview of the theory will be presented followed by a step-by-step design example. The document concludes with a discussion section about the component selection.

The theoretical gain of the LLC convener can be approximated using l irst harmonic approximation (FHA) techniques. Assuming the circuit is stimulated by a perfect sinusoid, one can use conventional circuit analysis to determine the voltage gain of the circuit. The LLC converter under study can be simplified to the circuit shown in FIG. 15 where Re = 8R/JI² [H. Huang, above.] and R is the I3C load resistance across a conventional full bridge output rectifier.

It can be shown that the voltage gain of the circuit is defined by:

where

(9)

1

Furthermore, one can find a transfer function between the input voltage and the resonant current, I_r. The phase of the resonant current determines the region of operation of the converter. For example, if the resonant current is leading the input voltage, the LLC converter is in the "LHS Operation" region. Conversely, wlien the resonant current is lagging the input voltage, the converter is in the "KHS Operation" region. The border between the two regions is where the resonant t-\nk behaves like a perfect resistor. The dashed line in FIG. 18 shows this border.

The valvies that make up the dashed line can be detennined by setting the imaginary part of the input voltage to resonant current transfer function to z« ro. The result is to solve for the roots of the following quadratic equation in ω² (ω=2πβ;

R„* - VH CTR? + LrnCr*.* ' in*) - «*i«" - O (12)

For voltage boosting applications, the circuit must be designed such diat it ( an operate with voltage gains greater than 1, In FIG- 18, this is achieved by designing the converter around a low Q value. As shown, lower Q values provide a larger voltage boost at the output. In addition to a low Q value, the convener will be operated at switching frequencies closer to the dashed line. Turse observations are in contrast to traditional LLC designs, where the converter is designed with larger Q values and operated near the resonant frequency, f₀. Designs that follow these traditional constraints exhibit unity voltage gain for .ill loads.

Converter Design Procedure

This section will present an iterative design procedure to design the components for an LLC circuit based on a low Q operation. Consider the following design constraints:

Hz Therefore, we can determine:

Using these design constraints, the C_r, Lj, and need to be determined.

As calculated above, this particular example of a converter requires a maximum gain of 4 based on the voltage that converter will be exposed to. Therefore, the method enables the determination of the resonant components that will yield che required maximum voltage gain, while operating in the LHS region. A skilled reader will appreciate that maximum gain drives the circuit design.

Design Steps 1) Ensure minimum gain is sufficient to offer high-resolution control of power with available control hardware. With existing hardware miniinum greater than 1.25 typically achieves this objective. If this minimum gair is too high for the application, introduce transformer with appropriate im ns ratio to ensure the required minimum gain.

2) Choose an L_ni/L- ratio that is suitable for the application. Typical values range from, but are not limited to, 3-10. Large values will result in higler peak currents in the tank, while small values will result in larger switching losses at low loads. 3) Generate voltage gain curves for various values of Q. On that plot, vlso graph the boundary curve separating LHS and RHS regions, similar to FIG. 25.

4) From the plot, select the Q value whose voltage gain curve intersects with boundary curve at the maximum voltage boost ratio, M_m„_KjmkI„,. This ensures the required maximum power can be transferred even wider maximum boosting conditions. Note the Q value and normali∑cd frequency (f_n) of this intersection point.

5) Using the Q and normalized frequency values found in step 4, calculate the I_T and C_r values using equations 9, 10 and 11.

6) Using the L. value calculated above and the desired L,„L_r ratio, calcu ate L_ra

The design process can be easily automated through software and can be app] led to any general form of the LLC circuit as shown in FIG. 26(a) with the interrupt switch and FIG. 26(b) without the interrupt switch.

Converter Design

This section will implement the design steps presented in the previous section to the convener constraints listed above.

1) Check if sufficient minimum gain conditions are met based on available control hardware. Here minimum gain is 2, which will allow high resolution power flow control using conventional control hardware.

2) Select an Ln, Lr rario of 5.

3) Zooming in on the voltage gain curves of FIG. 25 yields FIG- 27.

4) From the plot, choose a Q value of 0.123. This voltage gain cui ve intersects the resistive mode curve (the dashed line) at about 0.42 x fo-

5) Assigning f₀= fjwiictiine minimum and using equations 9, 10 and 1 ) , the Lr and Cr values can be determined to be:

• C_r = 28nF

• Ι^ = 1.8μΗ

6) Using the Lr value and the chosen L_m L,. ratio of 5, Lm = 9μΗ. The final converter design can then have LTX components with the following values:

• C_r = 28nF

• L_r = 1.8 H

· 1^ = 9μΗ

• <_ = 0.123

FIG. 28 shows the voltage gain curves of the designed converter, as well as the region of operation. Note how the region of operation remains in the "BUS Operation" region. The converter design described in the previous section is unique for the gi ven constraints and the selected Ln/Lr ratio. However, each time the designer selects new constraints, a new set of components must be calculated. As a consequence, there are in infinite number of different LLC converters that operate with high boosting and low Q. Table A shows a small sample of possible resonant Link component values for converters designed to operate at 300kHz and various Q and voltage boosting values. All of these converters may be successfully operated using frequency control to regulate load power.

TABLE A This design methodology is used to design resonant LLC converters with high voltage gain. Traditionally, resonant LLC converters arc designed with unity voltage gain, for voltage step down conversion. As a result, traditional designs will have larger Q values, and will operate near the resonant frequency f-n-

It will be appreciated by those skilled in the art that other variations of the embodiments described herein may also be practiced without departing from the scope of the invention. Other modifications are therefore possible. A skilled reader will recognize that there are numerous applications for the DC- DC converter technology described. The DC-DC converters of the present invem ion may provide an efficient, low cost alternative to numerous components providing high input-to-output voltage conversion. Moreover, DC-DC converters with high amplification ratios that are embodiments of the present invention may be use* I to create a fixed voltage DC bus in renewable/alternative energy applications.

A skilled reader will understand that the (A) method of operating a resonant DC - DC converter of the present invention; (B) the DC-DC converter disclosed herem; and (C) the method of designing a resonant DC-DC converter for high voltage boost ratio, may be used in connection with a range of different applications, including in connection with photovoltaic systems; a fuel cells; permanent magnet wind turbines; electric and hybrid vehicles; electric charge stations; aerospace systems; marine systems; power grids or smart grids including micro grids; and energy storage systems.

Claims

A method of operating a resonant DC-DC converter, the resonant DC- DC converter comprising a high voltage boost LLC circuit, characterized in that the method comprises :

(a) providing variable power flow control to the LLC circuit v ith externally determined input and output voltages using frequency control.

The method of claim 1, wherein the externally determined output volt ige is created by either a single externally determined output voltage, or a series connection of two externally determined output voltages to creale a bi-po^'Jar output.

The method of claim 1, wherein frequency control is applied such rha L ii emulates different loading conditions thus operating along horizo tal curves on the voltage gain compared to the switching frequency operating plane.

The method of claim 1, wherein the LLC circuit includes an LLC reson int tank, and wherein the LLC resonant tank operates with a minimum boosting having an effective value that is above unity over the entire operating range.

The method of claim 4, wherein the minimum boosting results in controllable transfer of power via change of switching frequency.

The method of claim 4, further comprising: maintaining an externally determined voltage gain and using frequency control to enable movemt nt between the load curves, and to control this movement within a frequency control region where there is horizontal separation amongst the load curves. The method of claim 1 , further comprising:

(a) operating the high voltage boost LLC circuit in a region close so a resonant frequency determined by a resonant inductor, magncti? ing inductor and a resonant capacitor, to achieve a high voltage bO'>sl; and

(b) utilizing unipolar or bipolar resonant tank excitation to improve converter efficiency in the high voltage boost circuit.

The method of claim 2, further comprising a balanced bipolar DC output wherein the output capacitor voltages are automatically balanced.

The rnethod of claim 2, wherein the DC-DC converter further includes a resonant inductor, a magnetizing inductor and a resonant capaciiOT, and 'he method comprises the further step of selecting these components such that ihe yield over the entire range of operation is an effective voltage gain tliat is greater than unity.

Tire method of claim 9, wherein the LLC converter is implemented with a transformer to allow decoupling of the resonant circuit gain from ihe externally determined voltage gain.

The method of claim 10, wherein the effective voltage gain value and ihe components are selected so as to minimize the effective voltage gain of i he resonant circuit, while being greater than unity, and provide controllability of the DC-DC converter via frequency.

The method of claim 2, further comprising operating at a range of input stage switching frequencies in an LLC circuit whereby a change in input voltage results in a change in load or transferred power, such that a decoupling between the input voltage and load is not required.

A resonant DC-DC converter for high voltage step-up ratio, characterized in that the resonant DC-DC converter for high voltage step-up rauo comprises: (a) a low voltage full-bridge or half-bridge DC-AC converter;

(b) an LLC resonant tank;

(c) a high voltage AC-DC converter or rectifier; and

(d) a high voltage controllable switch; wherein the high voltage controllable switch is controllable to regulate power flow from an input to an output of the DC-DC converter based on an externally determined voltage gain ratio, wherein the LLC resonant tank operates with a minimum boosting having an effective value above unity over the entire operating range.

The DC-DC converter of claim 12, designed to provide variable power flow control using frequency control.

The DC-DC converter of claim 14, wherein application of frequency control emulates different loading conditions thus enabling operation ak ng horizontal curves on a voltage gain compared to a switching frequency operating plane.

The DC-DC converter of claim 14, wherein the minimum boosting resi.lts in controllable transfer of power based on change of switching

.

The DC-DC converter of claim 14, that maintains an externally detemiired voltage gain, and uses frequency control to enable movement between t he load curves, and controls this movement within a frequency control region where there is horizontal separation amongst the load curves.

The DC-DC converter of claim 14, designed for:

(a) operation of a high voltage boost LLC circuit in a region close to a resonant frequency determined by a resonant inductor, magnetizing inductor and a resonant capacilOT, to achieve a high voltage boo>t; and (b) use of unipolar or bipolar resonant tank excitation to improve converter efficiency in the high voltage boost circuit.

The DC-DC converter of claim 14, further comprising a balanced bipolar DC output wherein output capacitor voltages are automatically balances,

The DC-DC converter of claim 14, wherein the DC-DC converter further includes a resonant inductor, a magnetizing inductor and a resonant capacitor, these components being selected such that the yield over the entire range of operation is an effective voltage gain that is greater dian unity.

The DC-DC converter of claim 14, comprising a transformer to allow decoupling of the resonant circuit gain from the externally determined voltage gain.

The DC-DC converter of claim 20, wherein the components are selected so as to minimize the effective voltage gain of the resonant circuit, while being greater than unity, and provide controllability of the DC-DC converter via frequency.

A method of designing a resonant DC-DC converter for high voltage boost ratio, the DC-DC converter comprising:

(a) a low voltage full-bridge or half-bridge DC-AC convener;

(b) an LLC resonant tank;

(c) a high voltage AC-DC converter or rectifier; and

(d) optionally, a high voltage controllable switch; wherein the high voltage controllable switch is controllable to regulnte power flow from an input to an output of the DC-DC converter based on a externally determined input to output voltage gain ratio maintained by t!ie high voltage controllable switch using frequency control, wherein the DO- DC converter includes (i) a resonant capacitor, (ii) a resonant inductor, and (iii) a magnetizing inductor; characterized in that the design method comprises:

(a) determining a minimum gain sufficient to enable high-resolution control of frequency using available control hardware;

(b) selecting an L„|/L, ratio that is suitable for an application for ihe DC-DC converter;

(c) generating voltage gain curves for various values of Q, and plotting these values so as to graph a boundary curve that defines LHS and HS regions, and selecting the 0 values whose voltage gain ctii ve intersects with boundary curve at the maximum voltage boost ratio, thereby defining a set of normalized frequency values; and

(d) using the Q values and (he normalized frequency values found to calculate values for the resonant capacitor, the resonant inductor, and the magnetizing inductor so as to enable selection of suitable components for the application.

The method of claim 1, wherein the output voltage is externally regulated.

The method of claim 24, further comprising externally regulating in output voUage and adjusting either current transfer or power transfer lor the externally regulated output voltage using a convener.

The method of claim 1, comprising applying the method in connection with operation of:

(a) a photovoltaic system;

(b) a fuel cell;

(c) a permanent magnet wind turbine;

(d) electric and hybrid vehicles;

(e) electric charge stations; (f) aerospace systems;

(g) marine systems;

(h) power grids or smart grids, including micro grids; or

(i) energy storage systems.