WO2011016854A1

WO2011016854A1 - Soft switching using a lossless snubber circuit in a power converter

Info

Publication number: WO2011016854A1
Application number: PCT/US2010/002152
Authority: WO
Inventors: Charlie Jourdan; John C. Elmes; Osama Abdel-Rahman
Original assignee: Advanced Power Electronics Corporation
Priority date: 2009-08-05
Filing date: 2010-08-03
Publication date: 2011-02-10

Abstract

A lossless snubber circuit that facilitates soft switching of a switching device in a power converter is provided. The present invention facilitates recovery of energy stored in the snubber capacitor and reroutes it back to the inductive load.

Description

SOFT SWITCHING USING A LOSSLESS SNUBBER CIRCUIT IN A POWER

CONVERTER

FIELD OF THE INVENTION

The invention disclosed here relates, in general, to the field of power converters. More specifically, the invention relates to a power converter that employs an inductive load with a switch.

BACKGROUND

Power converters are used in electrical engineering to convert electric power from one form to another. For example, power converters can be used to convert alternating current (AC) into direct current (DC) and vice versa. Yet another set of devices can be used to change the levels of AC or DC voltage or current.

One such type of converter is the DC-DC converter, which is used to change the levels of DC voltage. The DC-DC converter finds application in portable electronic devices such as cellular phones and laptops. Although these devices generally operate with a battery, they have multiple sub-circuits that require voltage at different levels other than that supplied by the battery. A DC-DC converter can be used to supply higher or lower voltages to these circuits. These converters can also aid power supply from the battery when the battery is running low by increasing the voltage supply from the battery, thus eliminating the need to have multiple batteries in the device and reducing the device size.

A certain converter within the class of DC-DC converters is the buck boost converter. A buck boost converter, as the name suggests, is a combination of a buck converter and a boost converter. A buck converter is essentially a step-down converter, whereas a boost converter is a step-up converter. A buck boost converter has an inductor inside it. Modifications in a buck boost converter design results in a flyback converter, which uses a transformer in the place of an inductor. The use of a transformer multiplies the voltage gain by the turns ratio, improving the efficiency and utility of the converter.

A flyback converter can operate in two modes— continuous conduction mode

(CCM) and discontinuous conduction mode (DCM). In the CCM, the current in the energy transfer inductor never reaches zero between switching cycles. In the DCM, the current reaches zero during part of the switching cycle. An advantage of the DCM operation is that the transformer's physical size can potentially be smaller than that in the CCM, as the primary inductance is smaller in the DCM. Another advantage of the DCM is that no energy is stored in the transformer between cycles, which can result in smaller transformer designs. Further, the switching device turns on with zero current, which reduces switching losses. However, the disadvantages of the DCM operation for the flyback converter are that there is a higher current swing in the transformer current, as well as higher peak currents, which could reduce the efficiency. In both modes of operation described above, there are significant switching losses in the converter, resulting in poor efficiency.

To overcome the drawbacks of the CCM and DCM operations, the flyback converter is sometimes operated in the boundary conduction mode (BCM). The BCM derives its name from the fact that the controller operates right on the boundary between the CCM and the DCM. In the BCM, the switch turns on and stores just enough charge to replenish the load during the time that the switch opens. The switch turns on again as soon as the entire charge is transferred to the output.

The boundary mode operation facilitates zero current turn on while minimizing the size of the transformer and the peak currents. However, even in the boundary mode operation, the flyback converter suffers from switching losses. During operation, the primary inductance develops a leakage inductance, known as primary leakage inductance, which attempts to continue the current flow in the primary winding when the switch is being turned off. This leads to resonance between the parasitic capacitances of the system which, if not damped, might damage the system. In previously known methods, flyback converters operated in the DCM, the CCM, or boundary modes suffered from power losses and reduced power density. Further, switching losses are present in any converter with a switch in series with an inductive load.

Therefore, in view of the shortcomings of methods known in the art described above, it is desirable to have a flyback converter having high efficiency, high power density, and low cost.

SUMMARY

The invention discloses a system and a method for achieving lossless switching in a converter.

The system comprises a snubber circuit which facilitates lossless switching in the circuit. The system is operated on the boundary conduction mode (BCM).

The invention also enables the utilization of energy generated in the snubber circuit by feeding it back to the input.

BRIEF DESCRIPTION OF THE DRAWINGS

Various embodiments of the invention will hereinafter be described in conjunction with the appended drawings provided to illustrate and not to limit the invention, wherein like designations denote like elements, and in which:

FIG. 1 illustrates a circuit diagram of a flyback converter design including a snubber circuit, in accordance with an embodiment of the invention;

FIG. 2 illustrates voltage and current waveforms of various components of the flyback converter of Fig. 1 ;

FIG. 3a illustrates a first mode of operation of the flyback converter;

FIG. 3b illustrates a second mode of operation of the flyback converter;

FIG. 3c illustrates a third mode of operation of the flyback converter; FIG. 3d illustrates a fourth mode of operation of the flyback converter;

FIG. 3e illustrates a fifth mode of operation of the flyback converter;

FIG. 3f illustrates a sixth mode of operation of the flyback converter;

FIG. 3g illustrates a seventh mode of operation of the flyback converter; and

FIG. 3h illustrates an eighth mode of operation of the flyback converter.

DETAILED DESCRIPTION OF THE INVENTION

In an embodiment, the invention provides a soft-switching variable frequency boundary mode flyback converter with lossless snubber. A flyback converter is derived from a buck-boost converter by replacing filter inductors with coupled inductors such as gapped core transformers. When the main switch turns on, the energy is stored in the coupled inductors as magnetic flux and is transferred to the output during the main switch off time. Since the flyback converter needs very few components, it is a very popular topology for low- and medium-power applications such as battery chargers, adapters, and consumer electronics.

Further, the flyback converter is able to regulate the output voltage at either negative or positive polarity with respect to the input voltage depending on the phasing of the output winding with respect to the primary. Most importantly, the transformer turns ratio in a flyback converter provides a way for achieving scaled voltages— step-up or step-down— with respect to the input voltage. The turns ratio is the ratio between the primary and secondary windings of the transformer.

FIG. 1 illustrates a circuit diagram of a flyback converter 100 with a snubber circuit, in accordance with an embodiment of the invention. The system comprises a voltage source 102. The voltage source 102 is a standard 220 volts source operating at 50 Hertz (Hz). In another embodiment, the voltage source can operate at 60 Hz. The circuit further comprises a magnetizing inductance L₃ 104, an auxiliary switch S₃ 106, diodes 108 and 110, a snubber capacitance 112, a MOSFET-based switching device S1 130 comprising a MOSFET 114, and a diode 116. The circuit further comprises a transformer having a primary winding 118 and a secondary winding 122. In one embodiment of the invention, the secondary winding 122 is a single winding. In another embodiment of the invention as shown in Fig 1 , the secondary winding 122 includes multiple stacked windings 122a to 122d. Various embodiments of the invention will hereinafter be described in conjunction with multiple windings stacked together to form the secondary winding 122. However, various embodiments of the present invention are equally applicable to the flyback converter 100 using the single secondary winding 122.

The secondary winding 122 is stacked to increase the output; the stacked windings on the secondary side are labeled 122a, 122b, 122c, and 122d. The secondary side further comprises diodes 120a, 120b, 12Oc₁ and 12Od. The secondary side also comprises capacitors 124a, 124b, 124c, and 124d. The equivalent capacitance 126 is connected at the output. Additionally, the flyback converter 100 includes a control circuit (not shown), which is responsible for the operation of the MOSFET-based switching device S1 130. The basic operation of the flyback converter 100 is well known in the art. The voltage source 102 is used to charge the primary winding 118. The reflected voltage across the secondary winding 122 charges the capacitor 124. The equivalent

capacitance of 124a, 124b, 124c, and 124d is the voltage generated at the output. The output voltage of the boundary mode flyback converter is dependent on the turns ratio of the transformer, the on-time of the MOSFET-based switching device S1 130, the input voltage, the converter design parameters, and the loading on the output voltage. As such, the on-time of the main MOSFET 114 is directly controlled to regulate the output voltage. Due to the nature of the boundary mode operation, it is more correct to consider this DC-DC converter as an energy converter, where the output voltage is dependent on the amount of energy transferred through the flyback transformer, as controlled by the on-time of S1 130.

The snubber capacitance 112 is placed across the MOSFET 114 through the diode 110. The snubber capacitance 112 is completely discharged while the MOSFET 114 remains on through an auxiliary switch S₃ 106. When the MOSFET-based switching device S1 130 is turned off, the current through the primary winding 118 flows through the diode 110 into the snubber capacitance 112 and thus, slowing the rate at which the voltage of the MOSFET-based switching device S1 130 rises. This process also, in-effect, by-passes the overvoltage generated at the junction of S1 130, thus protecting the MOSFET 114 from damage. This enables the MOSFET 114 to have zero- voltage turn off, which significantly reduces switching losses. Later in the cycle, when the MOSFET 114 is turned on, the auxiliary switch S₃ 106 discharges the snubber capacitance 112 back into the input voltage for use by the flyback converter 100. In effect, most of the soft-switching snubber energy is conserved and used in the next cycle, which is beneficial for a converter that has a high turns ratio and correspondingly high leakage inductance and parasitic capacitance.

The primary winding 118 is charged with a constant on-time current from the voltage source 102. The discharge time of the primary winding 118 is varied due to changes in the output voltage. This, in effect, results in a variable frequency operation of the converter. The operation of the flyback converter 100 will be explained in the following sections and will be considered in various timing modes.

FIG. 2 illustrates the voltage and current waveforms of the various components of the flyback converter 100 illustrated in FIG. 1. In Fig 2: 202 represents the operation waveform of MOSFET-based switching device S1 130; 204 is the switching operation of the auxiliary switch S₃ 106; 206 represents the current through the primary winding 118; 208 is the waveform for the voltage across the switching device S1 130; 210 represents the current through the secondary winding 122; 212 represents the voltage across the snubber capacitance 112; and 214 is the waveform of the current flowing through the magnetizing inductance 104. Further, the current through the switching device S1 130 is represented by 216. The various modes of operation of the flyback converter 100 will be explained in conjunction with FIGs. 2 and 3a-h.

FIG. 3a illustrates the first mode of operation of the flyback converter 100. The waveforms obtained during this mode can be referenced from FIG. 2. The first mode of operation begins at time instant to and terminates at time ti.

In this mode of operation, the primary winding 118 in series with the leakage inductance (not shown) resonates with the snubber capacitance 112 and a parasitic capacitance C_p 302. By slowing the rise rate of the voltage across the MOSFET-based switching device S1 130, zero voltage soft-switching (ZVS) is achieved. Zero

voltage/Current switching (ZVS/ZCS) can be defined as switching operations where each switch cycle delivers a quantized packet of energy to the converter output, and switch turn-on and turn-off occurs at zero current and voltage, resulting in a lossless switch. If the energy in the inductance is greater than the energy in the snubber capacitance 112, the peak primary inductance current is approximately the switch turn- off current. This can be shown mathematically in the following way:

c. + r

'pk = Jiqff² + ^v m ~ 'off

L ^Jm_m + ^τ L ^I_nk Vsi 130 (t) = Z_oi_pk sin(ω₀ (t - 1₀ )) + V_1n Vcs (ti) = V₀ ^ ₊ V_m

Where i_Ofr = Switch turn-off current

L_n,= Primary magnetizing inductance (from the primary winding 118 of the transformer) l_ik= Leakage inductance of the primary winding 118

Cp= Parasitic capacitance of the primary winding 118 Vsi ₁₃₀= Voltage across MOSFET-based switching device S1 130 ip_k= Peak primary inductance current Ni = Number of turns in the primary winding 118 N₂ = Number of turns in the secondary winding 122 Vo = Output voltage

C_s " Snubber capacitance Z₀ = Output impedance ω₀ = Angular frequency

Referring now to FIG. 2, as can be seen in the switching device voltage waveform 208, the voltage across the MOSFET-based switching device S1 130 rises when the primary inductance current is zero. It can be seen that the voltage rise rate across the MOSFET-based switching device S1 130 is low, resulting in Zero Voltage Switching (ZVS).

FIG. 3b illustrates the second mode of operation of the flyback converter 100. The waveforms obtained during this mode can be referenced from FIG. 2.

The second mode of operation begins at time instant ti and terminates at time instant t₂.

At the start of the second mode, the voltage across the primary side of the transformers is V₀ (Ni/N₂), where V₀ is the output voltage at the secondary side. During this cycle, the energy stored in the primary winding 118 is transferred to the secondary winding 122. In one embodiment of the invention, the current from the secondary inductance 122 is distributed among the stacked output windings (122a, 122b, 122c, and 122d) of the converter. In practical applications, stacked output windings enable lower voltage rating devices with higher output voltage and reduce the parasitic capacitance. The second mode of operation terminates when the current in the secondary winding drops to zero. This is shown in FIG. 2 by the secondary current waveform 210. The mathematical results for this mode of operation are provided below.

Where iu_sec is the current in the secondary winding 122 and

Ns_ta_cke_d is the number of turns in the stacked output windings (122a, 122b, 122c, and 122d).

FIG. 3c illustrates the third mode of operation of the flyback converter 100. The waveforms obtained during this mode can be referenced from FIG. 2. The third mode of operation begins at time instant t₂ and terminates at time instant X₃.

The third mode of operation begins when the energy stored in the flyback transformer is depleted, but a small amount of voltage across the parasitic capacitances 302 of the transformer exists. In this mode, the parasitic capacitance 302 resonates with the primary winding (Lpri) 118, eventually reducing the voltage across the MOSFET- based switching device S1 130 to 0 V and flowing negative current through the diode 116, allowing for ZVS and ZCS turn on. Referring now to FIG. 2, it can be seen in the waveform 206 that when t₂ < t < k, the current in the primary winding 118 is negative. The voltage across the MOSFET-based switching device S1 130 reaches zero at k, as shown in the switching device voltage waveform 208. The mathematical equations for this mode of operation are provided below.

l_LPΛt) - t₂))

V_sι(t) = V_Q -cos(ω_ι(t - t₂)) + V_i,

FIG. 3d illustrates the fourth mode of operation of the flyback converter 100. The waveforms obtained during this mode can be referenced from FIG. 2.

The fourth mode of operation begins at time instant t₃ and terminates at time instant U- The fourth mode of operation of the flyback converter 100 begins at the instant when the diode 116 of the MOSFET-based switching device S1 130 is conducting negative current (FIG. 2, switching device current waveform 216) from the previous resonance mode. This enables soft turn on of the MOSFET-based switching device S1 130 under ZVS and ZCS conditions. Mathematical equations are provided below. h_PΛO = (t - t₃) + i_Lpri(t₃)

^Lpri

FIG. 3e illustrates the fifth mode of operation of the flyback converter 100. The waveforms obtained during this mode can be referenced from FIG. 2. The fifth mode of operation begins at time instant U and terminates at time instant t-v.

The fifth mode of operation begins with the MOSFET-based switching device S1 130 turning on under ZVS and ZCS conditions (FIG. 2, switching device operation cycle waveform 202). When the current in the MOSFET-based switching device S1 130 is negative, the current will flow through the diode 116. Once the current is positive, it will flow through the MOSFET-based switching device S1 130. Mathematical equations are provided below: i_Si (t) = (t - t₃ ) ₊ i_Lpri (t₃ )

^Lpri FIG. 3f illustrates the sixth mode of operation of the flyback converter 100. The waveforms obtained during this mode can be referenced from FIG. 2.

The sixth mode of operation begins at time instant U- and terminates at time instant t₅.

The sixth mode of operation begins by turning on the auxiliary switch S₃ 106 for the snubber circuit (FIG. 2, auxiliary switch operation waveform 204), discharging the energy stored in the snubber capacitance 112 back to the voltage source 102. The sixth mode of operation ends when the voltage of the snubber capacitance 112 reaches 0 V. This causes the current in the magnetizing inductance Ls 104 to draw current from the primary inductance current (ipk), thus reducing the current through the switch S. This can be graphically seen in the waveform 214 of FIG. 2, where the inductor current rises between U- and t₅, reaching its peak at t₅. At the same time, it can be seen in the switching device current waveform 216 that the current through the MOSFET-based switching device S1 130 drops. Mathematical equations are provided below.

(O₂ =

IL_sC_p v_aM = v_m +vo N₁

N, h_s (U<) = 0

V_Cs(t) = V₀ - N÷₁ cos(ω₂(t -t₄,)) + V_in

N, i^{t) ^^m^ω₂ {t -t_v)) ₊ V_m

Z₂ N₂

FIG. 3g illustrates the seventh mode of operation of the flyback converter 100. The waveforms obtained during this mode can be referenced from FIG. 2. The seventh mode of operation begins at time instant ts and terminates at time instant fe.

The seventh mode of operation begins when the magnetizing inductance Ls 104 current begins drawing current from the primary winding 118. The equation for the primary inductor current (i_Lpπ) remains the same, but the current in the MOSFET-based switching device S1 130 reduces. This is shown in Fig. 2, switching device current waveform 216. To increase the efficiency, the delay time from when the switching device S1 130 is turned on until auxiliary switch S_a 106 is turned on can be optimized to allow the delay to be as long as reasonable without allowing the control to turn off the

MOSFET-based switching device S1 130 before the seventh mode is complete.

Mathematical equations are provided below.

»₁,(0 = 0

FIG. 3h illustrates the eighth mode of operation of the flyback converter 100. The waveforms obtained during this mode can be referenced from FIG. 2.

The eighth mode of operation begins at time instant fe and terminates at time instant to.

The final mode of operation begins when the magnetizing inductance 104 current has reached 0 A. At this instant, the current in the MOSFET-based switching device S1 130 is equal in magnitude to the current in the primary winding 118 (iι__Pri)- This mode of operation continues until the MOSFET-based switching device S1 130 is turned off. Thereafter, the cycle repeats from the start. Mathematical equations are provided below.

^Lpri

While the present invention has been described here in conjunction with DC-DC flyback converter 100, various embodiments of the present invention are equally applicable to any power converter such as DC-AC or AC-DC converters. Further, the soft-switching lossless snubber circuit is applicable to other topologies such as forward converters, boost converters, and other converters with an inductive load. Although the embodiments described above illustrate the present invention operating in the boundary mode of conduction, it will be readily apparent to those with ordinary skill in the art that the invention can also be practiced beneficially in the continuous conduction mode (CCM) and the discontinuous conduction mode (DCM). In accordance with the present invention, the soft-switching variable frequency boundary mode flyback converter with lossless snubber provides various advantages. The energy stored in the snubber capacitor is discharged back into the input current, while reducing the current that the switching device experiences, thus increasing efficiency. The new snubber circuit, coupled with optimal boundary mode operation, facilitates fully soft switching, high efficiency, and high power density. The new design proposed in the invention is also less complicated than previously known topologies, resulting in cheaper design.

While the preferred embodiments of the invention have been illustrated and described, it will be clear that the invention is not limited to these embodiments only. Numerous modifications, changes, variations, substitutions and equivalents will be apparent to those skilled in the art without departing from the spirit and scope of the invention.

Claims

What is claimed is: 1. A method for achieving lossless switching in a converter, the method comprising: a. operating a MOSFET-based switch to control operation of the converter; b. coupling the MOSFET-based switch to a snubber circuit;

c. by-passing an overvoltage generated at a junction of the MOSFET-based switch to the snubber circuit; and

d. feeding the energy discharged from the snubber circuit back in to an input voltage.

2. The method of claim 1 , wherein the snubber circuit comprises a snubber

capacitance and a diode.

3. The method of claim 1 further comprising discharging the energy in the snubber circuit through an inductor circuit coupled to the snubber circuit.

4. The method of claim 3, wherein discharging the energy in the snubber circuit is controlled by an auxiliary switch.

5. The method of claim 1 , wherein the converter is operated in a boundary

conduction mode.

6. A circuit for achieving lossless switching in a converter, the circuit comprising: a. a transformer with a primary winding and a secondary winding;

b. a MOSFET-based switch to control operation of the converter; c. a snubber circuit coupled to the MOSFET-based switch, wherein the snubber circuit is designed to store an overvoltage generated at a junction of the MOSFET-based switch; and

d. an inductor circuit coupled to the snubber circuit, wherein the coupled inductor circuit is used to discharge the snubber circuit.

7. The circuit of claim 6, further comprising an auxiliary switch, wherein the auxiliary switch controls the discharging of the snubber circuit.

8. The circuit of claim 6, wherein the snubber circuit comprises a snubber

capacitance and a diode.

9. The circuit of claim 6, wherein the MOSFET-based switch comprises a MOSFET and a diode.

10. The circuit of claim 6, wherein the secondary winding comprises one or more stacked secondary windings.