SI21983A - Circuit and procedure for the control of reluctance motors - Google Patents

Abstract

The invention deals with a circuit and procedure for the control of a reluctance motor within a drive, which is supplied from the mains network dealing with the method of operating the drive electronics, which requires less sub-assemblies and is consequently cheaper. The circuit consists of a RF filter (35), a rectifier bridge (36), a voltage gauge (42), an intermediate DC circuit (37), an output stage (38), a sensor (4) for measuring the rotor rotation angle or sensorless algorithm and commutation logic (43). The circuit is made in such a way that the voltage within the intermediate DC circuit (37) is of pulsating type therefore the electric current (iAC) from the mains network meets the regulations on electromagnetic compatibility (EN61000-3-2). The electric currents (i1, i2, i3) within the motor (39) phase coils have to be limited to prevent a blow-out of the output stage. This is achieved by limiting voltage within the motor (39) phase coils in such a way that the phase coils are switched on only, when the voltage within the intermediate DC circuit (37) or alternating mains voltage (uAC) regarding the motor (39) operational mode is adequately low.

Description

Circuit and process for controlling a reluctance motor

The invention relates to a circuit and control method for a reluctance motor, which is powered by a public network, and relates to a mode of operation of the drive electronics, which requires fewer sub-assemblies and is therefore less expensive.

Where household and other consumer goods require high speeds of rotation or where the speed of rotation must be adjustable, drives with universal collector motors are used. Universal collector motors are used mainly for their affordability. They consist of a statue of windings, a rotor with windings, and brushes and colecto, which commute the flow in the windings of the rotoi. As brushes wear off the collector, these motoes have a short life span. The brushes turn into dust that pollutes the environment. Rubbing the brushes against the manifold causes acoustic noise. Also, a spark appears on the edge of the brushes, which is a source of radio frequency interference.

Universal collector motors can be controlled by the simple and inexpensive electronic circuit shown in Figure 1. The universal collector motor 2 is connected in series with a triac 3 triggered by a triggering circuit 4. Part of the electronic circuit is also a radio interference filter 1. The universal collector motor is controlled by the opening angle α. During the opening angle α, before the voltage is transmitted through the zero trigger circuit 4, triac 3 opens through the current 3. At this point, current flows from the network / ac, which runs until the end of the half voltage period of the Mac network. · Triak 3 closes when the current through and s this causes the flow through the universal collector motor 2 to zero. The procedure is repeated every half-period of the voltage of the Mac- The diagrams 5 show an example when the opening angle α is small and thus the low power or speed of the universal collector moto is set 2. As the opening angle α increases, so does the power or speed universal collector motoi 2. The time diagrams 6 show an example where the opening angle α is large. Maximum power or velocity is obtained when the opening angle α is equal to the half-life of the Mac network ·

From a technical point of view, a brushless drive with a permanent brushless moto with a permanent magnet or a brushless drive with a reluctance moto could be used instead of universal drive motors. With these two types of motors, instead of brushes and colecto, it commutes current through the winding electronics, which is integral to the brushless drive, and which is complex and therefore expensive. The high price, compared to drives with universal collector moto, limits

-2Using brushless drives in consumer appliances. Brushless drive, on the other hand, does not have the technical limitations of drives with universal collector motors that result from the use of brushes and colecto.

A lower price would open up the possibility of using brushless drives in consumer goods. However, the price of the drive can be reduced by reducing the number of components on the electronics, that is, the electronic circuit components.

Powerless brushless actuators known and used from the public network have the subassemblies shown in Figure 2. Engine 12 may be a brushless permanent magnet motor with any number of phases or a reluctance motor with any number of phases. The phase windings of the motor 12 are connected to a power output stage 11 consisting of semiconductor switches that commute electrical current in the phase windings of the motor 12. This direction of flow must be in accordance with the angle of rotation of the motor rotor 12, which is measured by a sensor 15. V cheaper drives, sensor 15 is most often made with Hall sensors. Sensor 15 with optical position encoders is more expensive. Recently, the development of processor semiconductor circuits has made it possible to produce sensorless brushless drives, in which the motor 12 on the shaft lacks a sensor 15. The motor 12 is mechanically simpler in this case. The necessary information about the current angle of rotation of the motor shaft 12 in sensorless brushless drives is obtained from the measured electrical quantities and their conversion in processor semiconductor circuits. Information on the current angle of rotation of the shaft is required by the switching logic 14, which opens the semiconductor switches at power output stage 11.

The power output stage 11 receives electrical power from the intermediate DC circuit 10. To ensure that the voltage in the DC DC circuit 10 does not fluctuate too much, a smoothing capacitor 13. Provides a cooling capacitor 13. The cooling capacitor 13 is electrolytic. Its capacitance depends on the drive power and is usually between 100pF and some 1000pF. Because the smoothing capacitor 13 is large, it is an important part of the electronics price.

In front of the smoothing capacitor 13 there is a power factor corrector 9, which generates electrical current from the network / _A c- Unless there is a power factor corrector 9 in the electronics, the smoothing capacitor 13 would only be charged near the peak of the mains voltage w _A c and the charge amplitude of current and thus the flow from the network with ' _A c would be very large, as shown in the time diagrams 20 in the figure

4. All the rest of the time, the current from the grid with ' _A c would be zero. Since such a form of current from the network / _A c pollutes the public network severely, it is with the EN61000-3-2 electromagnetic compatibility standard.

-3 limited content of higher harmonic components in the electrical current drawn by the apparatus of the statement network. With the help of the power factor corrector 9, fill the smoothing capacitor 13 so that the current from the z'ac network contains fewer or more low-harmonic components than are allowed by EN61000-3-2.

The power factor corrector 9 can be implemented as a passive filter or as a switch converter. The passive filter requires a large choke which, by its reactance, greatly reduces the voltage in the DC dc circuit 10. Because the choke is large, it is heavy and also expensive. Due to the reduced voltage, weight and price, this version of the power factor corrector 9 is less suitable for use in consumer goods. The power factor corrector 9 with a switch converter consists of a damper, a semiconductor switch, a semiconductor diode, and a control that activates the semiconductor switch. In this embodiment, the choke is smaller than in the passive filter version and therefore the whole circuit is easier. Nevertheless, with all the necessary elements, the switch converter is still large and expensive.

The brushless actuator input has a bridge 8 and a radio interference filter 7. The bridge 8 directs the AC power to the Mac- In most cases, the Greatz bridge is used in bridge 8. The RF Interference Filter 7 detects interference from electronics due to rapid on and off currents and voltages. The RF filter 7 must be sized so that the apparatus complies with the applicable electromagnetic compatibility standards (EN55014-1).

A great deal of effort is being made to advise on reducing the number of components in the input part of the electronics, namely in the power factor corrector 9 and in the DC dc 10.

Figure 3 shows the circuit consisting of the Greatz bridge 16 and the load 17. The Greatz bridge 16 directs the voltage of the network "ac, so that a load pulsating voltage on the load 17 is obtained at the load 17" l- This voltage at the load m _{l is} followed by the current z'l. which flows through the load and which is also pulsating. When the current / l is reversed, the current from the z'ac network conforms to the electromagnetic compatibility standards (EN61000-3-2). Such a circuit was used by C. Larouci, J. P. Ferrieux, L. Gerbaud, J. Roudet, J. P. Keradec in the article Optimization of a PFC Flyback Converter and Discontinuous Conduction Mode, Proceedings PCIM 2002, Numberg, May 2002, where a switch 17 is used at the load site flyback type converter.

Typically, a capacitor is connected in parallel to the switch converter to block the radio frequency interference caused by the switch converter. In Fig. 4, the circuit of Fig. 3, where a capacitor 18 is added in parallel to the load. As long as the capacitor 18 is small, it only serves to block radio frequency interference caused by the load, such as a switch converter. Electrical sizes

-4 in this case they change as shown in the time diagrams 19. The voltage on the load ml is pulsating and each half-period drops to practically zero. The directed current / dc is the sum of the current z'l through the load and the condensate charge current 18. The z'dc current flows for most of the half-period so that when the z'dc current is reversed, the shape of the current from the z'ac network does not differ much from the sine and corresponds to electromagnetic compatibility standard (EN61000-3-2). As the capacitance 18 capacitance increases, the voltage on capacitor 18, or the load on ml, becomes smoother than the time diagrams 20 show. . The circuit pulls current from the z'ac network, which contains many higher harmonic components and therefore does not meet the electromagnetic compatibility standards (EN61000-3-2).

The permanent magnet brushless motor requires a fairly stable voltage in the DC DC circuit for its operation 10. The reason is the cutting voltage generated by the moto windings due to the rotation of the rotoi. The amplitude of the cutting voltage is equal to the constant speed of rotation. The voltage in the DC dc circuit 10 must be greater than the cutting voltage amplitude in order for the electric current 12 to flow into the motor 12, producing a positive torque. The electronics can provide a stable voltage in the DC dc circuit 10, provided that the smoothing capacitor 13 has a sufficiently high capacitance. Due to the large smoothing condensate 13, a power factor corrector 9 is required to allow electrical current to flow into the electronics from the z'ac grid of the proper shape.

The stator 21 and rotor 22 of the reluctance motoi in Figure 5 are designed such that the magnetic resistance for the magnetic flux or inductance L of the phase winding 23 varies with the rotation p. The inductance L changes periodically with period 25, otherwise in the case of the moto in Figure 5, the inductance L would have two periods.

The change in inductance due to the rotation p in the phase winding 23 in which the electric current z'f flows causes a torque 24 on the shaft. This applies to all phase windings in a reluctance motor. With the correct commutation of the electrical current in the phase windings, the torque 24 is positive. The voltage mf connected to the phase winding 23 should roughly be equal to the sum of the voltage drop 26 on the resistance of the phase winding 23, the voltage drop 27 on the inductance L of the phase winding 23, and the voltage drop 28 due to power generation on the shaft. Additionally, voltage drops occur in phase winding 23 as a result of the influence of other phase windings, but these drops are usually small and therefore not included in the equation in Figure 5. If the electric current z'f is small, the voltage drop 28, which is a consequence of generating power on the shaft, small even if the speed of rotation of the moto is high. Therefore, the voltage in the DC power circuit of a drive electronics with a reluctance motoq can be

-5pulsating, as in the time diagrams 19, in contrast to the voltage in the DC dc circuit 10 according to the known state of the brushless moto drive with permanent magnet according to Figure 2, which must be stable.

Following the interpretation of timing diagrams 19, we can see that electronics do not require power factor correction to control the reluctance moto if the voltage in the intermediate DC circuit is pulsating.

The authors of AC Clothier, S. Greetham, MH Haywood, NJ Leighton, NW Philips, in the patent application Power Conversion Apparatus, GB 2396491 A, use the fact that the shape of the current from the network with _A cv electronics complies with the electromagnetic compatibility standard (EN61000-3- 2) if the electronics in the intermediate circuit have a capacitor with a sufficiently low capacitance, as shown in timing diagrams 19 in Figure 4. In the patent application, said electronics is powered by a statement network and supplies power to the windings that this electronics switches on and off. The winding can be a transformer, as in the case of the flyback switch converter mentioned above. But windings can also be moto windings. The authors do not describe in patent application GB 2396491 A how they performed the start of the moto, or how they regulate the speed and power of the moto, but only describe the operation at full power.

When the reluctance motor rotates slowly, the voltage drop 28 due to power generation on the shaft is small, even if the electric current z'f is large. If a high voltage up were connected to the phase winding 23 at low speed, the electric current z'f would reach very high values. This occurs when the current value of ac power w _A c is at its peak value. At that point, the current z'f would increase so much in the currently turned on phase winding 23 that the semiconductor switches in the power output stage would be burnt, so we must limit it.

The z'p current can be limited by pushing a pulsating voltage on phase winding 23 'fs a frequency that is usually above the audible range, that is, a frequency higher than 20 kHz, which is generated by the circuit of Figure 6. The circuit consists of Greatz bridge 29, low capacitance capacitor 30, and semi-bridge 34 consisting of semiconductor switches VI and V2 and diodes Dl and D2. The phase winding 23 can be switched on softly, whereby the voltages and currents take the form shown in Fig. 7. If the phase winding 23 is rigidly switched on, the voltages and currents take the form shown in Fig. 8. The time axis in the diagrams in Fig. 7 and 8 is given at an angle p for which the rotor 22 rotates at some speed.

In the soft start shown in Figure 7, the semiconductor switches V i and V2 are switched on at moment t \, so that the current z'p through the phase winding 23 increases. This stream z'p runs along Route 32, or is there

-6currently equal to current fr and current z'dc in Figure 6. Then the semiconductor switch Vi is switched off and current fr begins to fall. At time i ₂ , current fr terminates along path 33 inside crescent 34. During this time, current fr and current frc are equal to 0. Interrupting current / dc, and therefore current from the network frc, causes major radio frequency interference.

In the solid state shown in Fig. 8, the semiconductor switches V i and V ₂ are instantaneously switched on, so that the current fr through the phase winding 23 increases. This current fr runs along path 32, or at the moment is equal to current fr and current frc in Figure 6. Then we turn off both semiconductor switches Vj and V ₂ and the current fr begins to fall. At time fr, current fr terminates along path 31 and charges the capacitor 30 so that the voltage on it begins to increase. The current frc equals 0. When the semiconductor switches Vi and V ₂ switch on, the current flows along path 32 again. At the moment fr, the capacitor 30 empties, so the current z'dc is still equal to 0. The same as in the case of soft switching, interrupting the current with ' dc and therefore the frc current also causes major radio frequency interference.

In known circuits, where the capacitance of the tensile condensate 13 is large in the DC dc 10, the current spikes in the fr circuit are smoothed out so that they do not occur in the current / dc.

Another approach to making cheaper electronics was chosen by L. Helle, G. K. Andersen, F. Blaabjerg, P. A. Rasmussen in the article An Integrated Single-Phase Power-Factor-Switched Reluctance Motor Drive, Proceedings PCIM '99, Ntimberg, June 1999. Automobiles are able to reduce the cost of propulsion by reducing the required capacitance of the smoothing condensate by controlling the reluctance moto 13. The semiconductor switches in power output stage 11 control such that the current power consumption in the motor is the same as that provided by the power factor corrector 9 Therefore, the role of the condensatoil 13 for energy storage is reduced and may have very low capacitance. The drive price is further reduced by winding the power factor correction choke 9 on the moto stator.

Known circuits and control procedures for reluctance moto in public-powered drives that comply with electromagnetic compatibility regulations are costly in terms of components and therefore unsuitable for installation in relatively inexpensive consumer goods or cause radio frequency interference at certain operating points requiring high filter for radio frequency interference.

-7The object and object of the invention is to provide such a circuit and process for controlling a reluctance moto in a publicly powered plant that complies with the regulations on electromagnetic compatibility and which is cheaper in components and therefore suitable for installation in relatively inexpensive consumer goods.

According to the invention, the problem is solved by a circuit and process for controlling a reluctance moto according to independent claims.

The following will describe the circuit and the process of controlling the reluctance moto using pictures showing:

Figure 1: Known circuit and control procedure for universal collector motor,

Figure 2: Known brushless motor control circuit,

Figure 3: Known non-smoothing condensation circuit and its operation,

Figure 4: Known smoothed capacitor circuit and its operation,

Figure 5: Example of a known three-phase reluctance motoi,

Figure 6: Directions of currents in the camshaft with the known pulse power supply of the moto phase winding,

Figure 7: Timing diagram of known pulse power supply for moto phase winding,

Figure 8: Flowchart of known pulse power supply for moto phase winding,

Figure 9: circuit for controlling the reluctance moto according to the first embodiment of the invention,

Figure 10: a circuit for controlling a reluctance moto according to another embodiment of the invention,

Figure 11: a circuit for controlling a reluctance moto according to the third embodiment according to the invention,

Figure 12: an example of a power output stage for powering a reluctance moto according to the method of the invention,

Figure 13: time diagram of the output current stages according to the invention,

Figure 14: Timing diagram of the currents and voltages of the control according to the first method according to the invention,

Figure 15: Timing diagram of the currents and voltages of the control according to the second method of the invention.

-8 According to the invention and the circuit of FIGS. 9, 10 and 11, electronics from a public power supply network with a reluctance moto in an intermediate DC circuit 37 located between the bridge bridge 36 and the power output stage 38, a capacitor 40 with very low capacitance, such that the voltage on it every half-period drops below 25% of the highest voltage on it in the same half-period and the voltage-dependent circuit 41 or the electronics in the DC-DC circuit 37 have only a voltage-dependent circuit 41 without a capacitor 40. At the voltage-dependent circuit 41, the magnetic is converted to heat energy recovered from the engine 39. According to the invention and circuit of Figures 9, 10 and 11 from a public network, a powered reluctance moto drive does not require a power factor corrector. According to the invention and the method of Figures 12 and 13, in which the electrical output of the form 38, which causes less radio frequency interference, flows into the power output stage 38. According to the invention and the method of Figures 14 and 15, where we limit the current through the windings of the motor 39 by switching on the phase currents of the moto 39 synchronously with the AC power of the Mac network.

Although the explanation of the invention is made on a three-phase reluctance motor, it is also applicable to a two-phase or multi-phase reluctance motor.

Circuits for controlling a reluctance moto 39 with two or more phases according to embodiments in Figures 9, 10 and 11 include a filter for radio frequency interference 35, a bridge bridge 36, an intermediate DC circuit 37 with a capacitor 40 and a voltage-dependent circuit 41, or only with a voltage-dependent circuit 41, power output stage 38, sensor 44, switching logic 43, and voltage meter 42. Sensor 44 can be replaced by a sensorless algorithm. A voltage meter 42 is understood to mean an electronic circuit that provides information about the passage of voltage across a Mac network through zero.

The on and off currents and voltages in power output stage 38 are very fast and therefore cause radio frequency interference. Therefore, it is desirable to give as little switching as possible. The smallest number of switches occurs when switching current only from one phase winding 46 to the next phase winding 47 and then phase winding 48. During one period of change of inductance 25 due to rotation, each of the phase windings of moto 39 is switched on only once and once switched off with the exception of very low speeds and starts when the period of one inductance change period 25 is longer than the half voltage period of the Mac mains. , but only once during the Mac half-life ·

-9Figure 12 shows a circuit of power output stage 38 to which a phase winding 46, a phase winding 47 and a phase winding 48 of a reluctance moto 39 are connected. The circuit of the power output stage 38 consists of diodes Di to D6 and semiconductor switches Vi to Ve, which can be bipolar transistors, MOSFETs or IGBTs. Through phase winding 46 flows a stream zj, through phase winding 47 flows a stream zj and through phase winding 48 flows current z / 3. The flow paths zj, zj and zj are presented in the time diagrams in Fig. 13, which shows the occurrence of a single revolution, where the time of one revolution is much shorter than the voltage half-period «ac ·

At the moment these in the timing diagrams in Figure 13, semiconductor switches V are enabled! and V2 such that the current zj through phase winding 46 flows along path 50. The current zj at this point is equal to the current zj.

The process of switching the current from phase winding 46 to phase winding 47 begins with the activation of semiconductor switches V ₃ and V4 or phase winding 47. The moment of switching on phase winding 47 changes depending on the speed and load of the moto 39. In a moment / 2 the current zj increases and flows along path 52. The current zj is now the sum of the currents zj and zj. Phase winding 46 is switched off by first switching off only the semiconductor switch Vi, so that at instant / 3 the current zj terminates along path 51 via the semiconductor switch V ₂ and the diode Dj inside the semiconductor 34 in the power output stage 38. The inductance of the phase winding 46 must in during this time it is increasing that the current zj in the phase winding 46 generates torque and thus decreases. Instead of the semiconductor switch Vi, the semiconductor switch V2 could be switched off by terminating the current through the semiconductor switch Vi and diode D2. The time when the current zj terminates inside the semicircuit 34 may be longer than 5% of the period of change of inductance 25. When the current zj becomes greater than the current zj, the semiconductor switch Vi is switched off. The current zj then at instant G flows from the camshaft 34 along the path 49 and is subtracted from the current zj flowing along the path 52. Since the current zj is greater than the current zj, the current zj is greater than zero. When the current zj drops to zero, the commutation current from phase winding 46 to phase winding 47 is complete.

When the motor 39 is rotated forward, the commutation of the current from the phase winding 47 to the phase winding 48 begins, thus rotating forward.

During switching, the current did not drop to zero or start to flow back to the DC link

37. Therefore, capacitor 40 is not required during operation. If both semiconductor switches Vi and V2 were to be switched off at time h, the current zj would start to flow back or the energy would return from the magnetic field in the motor 39 to the intermediate DC 37. This energy can be stored in the capacitor 40, or change to heat on the voltage-dependent element 41.

-10 When starting and controlling the power or speed, it is necessary to limit the currents through the moto windings

39. This is achieved by reducing the voltage on the windings of the reluctance motor 39 by switching on the phase windings of the moto 39 synchronously with the voltage of the Mac network. When the motor is rotating very slowly, the phase windings of the motor 39 are switched on only near the passage of the Mac voltage through zero. The aperture angle a, as indicated in Figures 14 and 15, is small in slow rotation. At the opening angle a, the electronics switch the currents zj, z ' ₂ and z ₃ in the motor phases 39 as required by the rotation of the rotoi. As the speed of the moto 39 increases, it can increase as well as the openings a, without causing too much current through the phase windings of the motoi 39. When the motor 39 reaches the rated speed, the opening angle a is equal to π. At that time, power output stage 38 commutes the current in phase windings of moto 39 without interruption. Figures 14 and 15 also indicate the time of one rotation 53 of the moto 39 insofar as the motor 39 would be identical to the moto in Fig. 5.

There are two ways of controlling the power or speed of the reluctance moto 39 at start-up and during operation with the opening angle a. In the first mode, the electronics switch the phase windings of moto 39 on and off so that the electrical currents zj, z ' ₂ , z ₃ in moto phase 39, start to flow at the moment of angle β near the voltage crossing' ac through 0 and then flow only during the angle openings a, as shown in Figure 14. In another way, the electric currents zj, z, ₂ , and ₃ in moto stages 39 only flow at the time of the opening angle a and cease to flow at the moment of angle β n near the passage of voltage across the Mac through zero, as is shown in Fig. 15. This second mode is similar to the method of controlling the universal collector motor 2 with triac 3, as shown in Fig. 1. In both modes, the opening angle time a can take values from zero to half the grid voltage period «ac ·

The angle β denotes how much the currents zj, z ₂ , z ' ₃ in moto stages 39 overtake or delay the moment when the voltage of the Mac network passes through zero. Like β, when the electric currents zj, z ' ₂ , z ₃ start or stop flowing, less than 20% of the half-life of the Mac network voltage is transmitted at zero over zero time ·

According to the first embodiment of the circuit of Figure 9, the voltage meter 42 provides the switching logic 43 with information about the current voltage value of the Mac network or the time when the voltage across the Mac network passes through zero. Thus, the switching logic 43 can be controlled synchronously with the voltage of the Mac network as the openings of a .. In addition, the switching logic 43 also takes care of the commutation of currents zj, z, ₂ , ₃ depending on the rotation of the rotoi moto 39. For its operation, the switching logic 43 also needs information about the desired speed or torque.

In the second embodiment, the circuit of Fig. 10 interrupts the flow through the phase windings of the motor 39 synchronously with the public network, thereby controlling the openings with an additional semiconductor switch 45 in the intermediate DC circuit 37 to which information about the current value of the voltage u ^ c or about the moment when the voltage of the Mac network passes through zero. The semiconductor switch 45 may be in the positive or negative branch of the DC link 37.

According to the third embodiment of the circuit of Fig. 11, we synchronously with the public network interrupt the flow through the phase windings of the motor 39 and thereby control the openings a in the bridge bridge 36, where controlled semiconductor switches are used. In this case, we keep information about the current value of the Mac network voltage, or the moment when the Mac network voltage passes through zero to the bridge bridge 36.

In the last two cases, the power output stage 38 only directs the current relative to the rotation of the rotor to the corresponding motor winding 39.

Claims (12)

Patent claims A method for controlling the power or speed of a reluctance moto (39) at start-up and in operation, characterized in that the electronics switch on and off the moto winding phase (39) so that the electric currents in the moto stages (39) begin to flow momentarily at an angle (β ) near the voltage grid crossing («ac) through zero and then only flow at the opening angle (a). 2. A method of controlling the power or speed of a reluctance moto (39) at start-up and during operation, characterized in that the electronics switch on and off the phase winding moto (39) so that the electric currents in the moto stages (39) flow only during the opening angle ( a) and cease running at the instant of angle (β) near the voltage crossing of the network (« _A c) through zero. Method according to claims 1 and 2, characterized in that the opening angle time (a) can take values from zero to half the grid voltage period («ac) · Method according to Claims 1 and 2, characterized in that the angle (β) is less than 20% of the network voltage half-cycle («ac) over a time interval (« ac) · Method according to Claims 1 and 2, characterized in that each phase winding of the moto (39) is only switched on and off once for a period shorter than one period of change of inductance (25) due to rotation and at the same time shorter than the half voltage of the grid ( «Ac) · Method according to Claims 1 and 2, characterized in that, after switching off the phase winding, the electrical current is first terminated inside the semiautomatic switch (34) in the power output stage (38) for a period longer than 5% of the inductance change period (25), then however, the off-phase electric current flows from the camshaft (34). Method according to Claims 1, 2, 5 and 6, characterized in that the switching logic (43) controls the opening angle (a) synchronously with the voltage of the network («ac). Method according to claims 1, 2, 5 and 6, characterized in that the semiconductor switch (45) in the intermediate DC circuit (37) controls the openings angle (a) in synchrony with the voltage. Method according to claims 1, 2, 5 and 6, characterized in that the pier bridge (36) containing the controlled semiconductor switches controls the opening angle (a) in synchrony with the mains voltage («ac). Method according to claims 7 to 9, characterized in that the synchronization of the opening angle control (a) is achieved by means of a voltage meter (42). -1311. A circuit for carrying out the process according to claims 1 to 10 with electronics for controlling the reluctance device comprising a bridge bridge (36), an intermediate DC circuit (37) and a power output stage (38), characterized in that a capacitor ( 40) and voltage dependent circuit (41). A circuit for performing the process according to claims 1 to 10 with electronics for controlling the reluctance device comprising a bridge bridge (36), an intermediate DC circuit (37) and a power output stage (38), characterized in that they are located in the intermediate DC circuit only voltage dependent circuit (41) without capacitor (40). Circuit according to claim 12, characterized in that the voltage-dependent circuit (41) is transformed into heat by the magnetic energy that returned from the motor (39).