MXPA02000016A

MXPA02000016A - Switching power supply circuit.

Info

Publication number: MXPA02000016A
Application number: MXPA02000016A
Authority: MX
Inventors: Yasumura Masayuki
Original assignee: Sony Corp
Priority date: 2000-05-11
Filing date: 2000-05-11
Publication date: 2002-07-02
Also published as: WO2001086796A1; KR20020029904A; CA2379089A1; EP1281233A1; TR200200035T1; AU4431900A; CN1360750A; US6587358B1

Abstract

A switching power supply circuit, which is small, lightweight, and highly efficient in power conversion, includes an insulating converter transformer, which is formed by a core with no gap and a primary and secondary winding wound on the core such that the mutual inductance between the windings exhibits an additive mode. A half wave rectifier circuit is provided on the secondary side of the circuit and performs a rectification operation in the additive mode to obtain a secondary side DC output voltage. In a constant voltage control circuit system for stabilizing the secondary side output voltage, the switching frequency of a switching element is varied in response to the secondary side output voltage level to control the resonance impedance of a primary side parallel resonance circuit and the continuity angle of the switching element compositely.

Description

(( CIRCUIT OF SUPPLY OF SWITCHING ENERGY Technical Field This invention ST relates to ur. switching power supply circuit that can be incorporated as a power supply in various electronic devices.

BACKGROUND ART A switching power supply circuit can adopt a switching converter such as a return converter or a feed converter. Since these switching converters use a rectangular waveform signal for a switching operation, a switching power supply circuit adopting said converter can also be called a hard switching power supply. Figure 7 illustrates a hard switching power supply circuit 700 adopting a Call Throttle Converter ("RCC") system. The power supply circuit 700 is used as a supply d? standby power provided separately from a main power supply and is constructed in order to satisfy, for example, a low load condition where the charging power (Po) is 50 W or less or another condition where the charging power Po is 0.5 W or less. As shown in Figure 7, the power supply circuit 700 includes a CVT converter transformer having a drive winding NB, a primary winding NI on one primary side, and a secondary winding N2 on a secondary side. In this way, the power supply circuit 700 is divided into a primary side 710 and a secondary side 715. The power supply circuit 700 includes a rectifier filtering circuit 705 for receiving a commercial Alternating Current ("AC") power supply with an input AC voltage VAC and producing a Direct Current input voltage Ei (" DC "). The rectifier filtering circuit 705 is a full wave voltage multiplication rectifying circuit composed of a bridge rectifier Di circuit and a filtering capacitor Ci. The rectifier smoothing circuit 705 produces rectified filtered DC input voltage E i that is substantially equal to the AC input voltage VAC. In addition, an incoming current limiting resistor Ri is interposed in a rectifier current path of the rectifier smoothing circuit 705 in order to suppress any initial incoming current peak flowing to the filtering capacitor Ci, for example, when the AC power supply is initially provided to circuit 705. Switching element Ql receives and switches DC input voltage Ei to produce a switching output. Illustratively, a bipolar transistor is used for switching element Ql. The collector of the switching element Ql is connected to a positive electrode terminal of the filtering capacitor Ci through a series connection to the primary winding NI of the CVT converter transformer. The base of the switching element Ql is connected to receive the rectified filtering voltage Ei through a starting resistor RS so that a base current can be supplied thereto upon initiation. In addition, a series circuit connection of the base current limiting resistor RB, a diode D4, and drive winding NB is connected to the base of the switching element Ql. One end of the drive winding NB is grounded. A CB capacitor is connected in parallel to the diode D, the base current limiting resistor RB, the diode D4, the drive winding BN, and the CB capacitor cooperatively form a self-excited oscillation drive circuit that oscillates and drives the element Ql of switching in a self-excited manner The emitter of the switching element Ql is connected to ground through a resistor R7. The converter CVT transformer is provided to transmit a switching output obtained by the primary side 710 of the power supply circuit 700 to the secondary side 715 and has the primary winding NI and the secondary winding N2 wound therein. Likewise, the drive winding NB for self-excited oscillation described above is wound on the primary side of the converter CVT transformer A half-wave rectifier circuit formed by a rectifying diode D01 and a filter capacitor COI is connected to the secondary winding N2 of the CVT transformer of converter and produces and outputs a secondary side DC output voltage EOl, the secondary side DC output voltage EOl is supplied to a load (not shown) and further input is given as a detection voltage to a circuit 7 control for constant voltage control.

The control circuit 7 includes a PC photocoupler to isolate, in DC, parts thereof on the secondary side 715 of parts on the primary side 710. On the secondary side 715, the control circuit 7 comprises a pair of resistors R3 and R4 which divide the DC output voltage EO1 of the secondary side, and the divided voltage is input to a detection input of a detection element Q3. . One end of the sensing element Q3 is connected to receive the DC output voltage EO1 from the secondary side through a series connection of a resistor R1 and a photodiode PD of the photocoupler PC. The other end of the detection element Q3 is connected to ground. A series circuit connection of a capacitor Cll and a resistor R2 is connected in parallel to the resistor R4. Another series circuit connection of a capacitor C12 and a resistor R5 is connected through a joint between the resistors R4 and R3 and a joint between the detection element Q3 and the photodiode PD. On the primary side 710, the control circuit 7 includes a phototransistor PT of photocoupler PC. A half wave rectifier circuit formed by a diode D3 and a capacitor C3 for rectifying and filtering an alternating voltage excited in the drive winding NB is connected to the collector of the phototransistor PT so that a low DC voltage obtained by the rectifier circuit of Half wave (D3 and C3) can be supplied as power supply to PT phototransistor operation. The emitter of phototransistor PT is connected to the base of a transistor Q, which serves as an amplifier. A series circuit connection of a resistor R8 and a ZD Zener diode is inserted between the emitter of the phototransistor PT and a joint between the drive winding NB and the diode D4. The collector of transistor Q4 is connected to the base of switching element Ql, and the emitter of transistor QA is connected to ground. The base of the transistor QA is connected to the emitter of the switching element Ql through a parallel circuit connection of a resistor R6 and a capacitor C13, and is thus connected to ground through the resistor R7. A reset circuit 10 is formed by connecting the DRS diode in series to a parallel circuit connection of a RES resistor and a CRS capacitor. The reset circuit 10 is connected in parallel to the primary winding Ml. A protection circuit 11 includes a capacitor Csn connected in series to a resistor Rsn. The collector of the switching element Ql is connected to ground through the protection circuit 11, the reset circuit 10 and the protection circuit 11 are required to suppress a peak voltage that appears when the switching element Ql is switched off, The switching operation is initiated by applying current to the switching element Ql through the starting resistor RS, thereby connecting the switching element Ql. When the switching element Ql is connected, the magnetic energy is stored towards the primary winding NI of the CVT converter transformer. When the switching element Ql is disconnected, the magnetic energy stored in the winding or primary NI is discharged to the secondary side of the converter CVT transformer. This operation is repeated to produce an output voltage on the secondary side of the transformer CVT transformer. The control circuit 7 varies the amount of current that passes through the sensing element Q3 in response to the DC output voltage EO1 of the secondary diode. The PC photocoupler controllably varies the base current supplied to the transistor QA in response to the amount of current flowing through the sensing element Q3, thereby varying the collector current of the transistor QA. Since the collector of transistor QA is connected to the base of the switching element Ql, the base current (amount of driving current) to flow from the self-excited oscillation drive circuit (resistor RB), diode D4, drive NB, and capacitor CB) to the base of the switching element Ql varies in accordance with the collector current of transistor Q4, consequently, the connected time of the switching element Ql is varied, and as a result, the frequency of switching is varied in a controllable manner, thereby realizing constant voltage control, the power supply circuit 700 having the construction shown in Figure?, a constant voltage effect can be obtained by controlling a switching frequency (fs) to increase in response to a rise in AC input voltage VAC or a decrease in load Po energy. The control scale of the switching frequency fs is set to a wide scale of 25 KHz to 250 KHz because the control sensitivity is low. The waveform diagrams in Figures 8A to 8C illustrate an operation of power supply circuit 700. A voltage Vcp through the switching element Ql and resistor R7 (between the collector of the switching element Ql and earth) has a waveform such as that shown in Figure 8A. As shown by Figure 8A, the voltage Vcp exhibits a level U during a TON period when the switching element Ql is on, but exhibits a shape of a rectangular pulse through a TOFF period when the switching element Ql is disconnected As can be seen from the impulse waveform of the voltage Vcp shown in FIG. 8A, a peak voltage is generated by disconnecting the switching element Ql by a leakage inductance component of the CVT converter transformer and a distributed capacitance. (electrostatic capacity) between the windings NI, N2 and NB in the converter CVT transformer. The reset circuit 10 and the protection circuit 11 are provided in order to suppress a portion of the voltage waveform Vcp when the peak voltage appears. A collector current Icp flows into the collector of the switching element Ql in response to a switching operation of the switching element Ql. The collector current Icp flows in a waveform as shown in Figure 8B during the TON period. A rectified current 12 flows from the secondary winding N2 to the rectifying diode D01 during the TOFF period when the switching element Ql is switched off, corresponding to the switching operation of an RCC. The switching (reverse or forward) converters used in a hard switching power supply, such as circuit 700, are limited in energy conversion efficiency and the amount of switching noise they can suppress. In this way, various soft switching energies, which employ resonance type switching converters, can be used. A resonance type switching converter is advantageous because it can easily obtain a high energy conversion efficiency. Said converter also generates lower noise than a converter used in a hard switching power supply because the waveform of a switching operation of a resonance type converter is a sinusoidal waveform. The type of resonance is also advantageous because it can be formed from a comparatively small number of parts, a switching power supply circuit 900 satisfying the low load condition of having a face Po energy of 50 W or less or 0.5 W or less is shown in Figure 9. For simplicity, similar portions in Figure 9 to those in Figure 7 are denoted by similar reference numbers and their description is omitted. Referring to Figure 9, the power supply circuit 900 includes a self-excited converter of the current resonance type wherein a rectified filtered voltage Ei is used as a supply of operating power. The switching converter of the power supply circuit 900 includes a pair of switching elements Ql and Q2 connected in a half-bridge connection, as shown in Figure 9, and interposed between the positive electrode side seal of a Ci capacitor of filtration and ground. The starting resistors RS1 and RS2 are interposed between the collector and the base of the switching elements Q1 and Q2, respectively. A pair of holding diodes DD1 and DD2 are interposed between the base and the emitter of the elements Q1 and Q2 of switching, respectively. A series circuit connection of a resonance capacitor CB1, a resistor RB1 of base current limitation, and a drive winding NB1 (having an inductance LB1) is interposed between the base of the switching element Ql and the collector of the switching element Q2, thereby cooperatively forming a series resonance circuit for self-excited oscillation and determination of the switching frequency of the switching element Ql. Similarly, another series circuit connection of a resonance capacitor CB2, a resistor RB2 of base current limitation., And a pulse winding 1SIB2 (having an inductance LB2) is interposed between the base of the switching element Q2. and ground, forming a series resonance circuit for self-excited oscillation and determining the switching frequency of the switching element Q2. The partial resonance capacitors CC1 and CC2 are connected between the collector and the emitter of the switching elements Q1 and Q2, respectively . The partial resonance capacitors CC1 and CC2 are provided in order to absorb the switching noise of the switching elements Q1 and Q2, respectively. They also act to obtain a zero-voltage switching operation by disconnecting the switching elements Ql and Q2, respectively, in accordance with the switching frequencies which are controlled by a constant voltage control operation performed in a manner such as is describe later. The operation reduces the switching loss, A PRT drive transformer (Energy Regulation Transformer) is provided to drive the switching elements Ql and Q2 and controllably vary the switching frequencies to perform constant voltage control. The drive in the power supply circuit 900 is an orthogonal saturable reactor in which the driving windings NBl and NB2 and a resonant current detection winding ND are wound, and in which a control NC winding is wound in a direction orthogonal to the windings NBl, NB2, and ND One end of the driving winding NBl of the driving transformer PRT is connected to the base of the switching element Ql through a series connection to the resonant capacitor CB1 and the resistor RB1 of base current limiting, and the other end of the NBl drive winding is connected to the emitter of the switching element Ql, one end of the driving winding NB2 is connected to ground, and the other end of the driving winding NB2 is connected to the base of the switching element Q2 through a connection in series to the capacitor CB2 of resonance and the RB2 icon of base current limitation. The drive winding NBl and the driving winding NB2 are coiled so that they can generate voltages that have opposite polarities, An isolation converter transformer (Energy Isolation Transformer j transmits switching outputs of the switching elements Ql and Q2 to In this case, one end of the primary winding NI of the isolation converter transformer PIT is connected to a junction (switching output point) between the emitter of the switching element Ql and the collector of the switching element Q2. through the ND winding of resonance current detection The other end of the primary winding NI is connected to ground through a capacitor Cl series resonance so that a switching output signal can be obtainedIn this way, a series resonance circuit for a current resonance type switching converter operation is formed of a capacitance capacitor Cl of series resonance and a leakage inductance component of the PIT transformer of isolation converter including the primary winding NI (series resonance winding), On the secondary side of the transformer PIT of the isolation converter, an alternating voltage with a switching period is excited in the secondary winding N2 by the switching output signal supplied to the primary NI winding. A bypass is provided for the secondary winding N2, and the rectifier diodes DOl, D02 and D03, and the filtering capacitors COI and C02 are connected as shown in Figure 9. A full-wave rectifying circuit is formed by the diodes DOl and D02 of rectifier and capacitor COI of filtration, and a rectifier circuit of half wave is formed by diode C03 of rectifier and capacitor C02 of filtration. The full wave rectifier circuit formed by the rectifier diodes DOl and D02 and the filtering capacitor COI performs a full wave rectification operation to produce an output voltage EO1 of DC and supplies electrical power to a load (not shown) in the next stage. The DC output voltage EO1 is also admitted to a control circuit 9 and is used as a detection voltage by the control circuit 9. In addition, to protect the load from short circuit, a fuse, for example, an integrated circuit ("IC") link, IL interposes - íe - between the DC output voltage EOl and the < The control circuit 9 supplies a DC current, the level of which is varied, for example, in response to the level of the DC output voltage EO1 of the secondary side, as a control current to control the NC winding of the transformer PRT of drive to perform constant voltage control in the manner as described below. The control circuit 9 includes a pair of resistors R3 and R4. between which the DC output voltage EOl is divided from the secondary side. The divided voltage is input to a detection element Q3. The cathode of the detection element Q3 is connected to the positive electrode of the filtering capacitor C02 through a series connection to control the NC winding, and the anode of the detection element Q3 is connected to ground. A series circuit connection of a capacitor C12 and a resistor P5 is interposed between the positive electrode of the filtering capacitor COI and a joint between the resistors R3 and R4, another serial circuit connection of a capacitor Cll and a resistor R2 of interposes between the cathode of the sensing element Q3 and the joint between the resistors R3 and R4. In a switching operation of the power supply circuit 900, the starting current is supplied to the bases of the switching elements 01 and Q2. through the initial resistors RS1 and RS2, respectively, when the AC power supply is provided first. For example, if the switching element Ql is switched on first, then the switching element Q2 is controlled so that it is switched off. Then, as one. output of the switching element Ql, a resonance current II flows through the resonance current detection winding ND, the primary winding NI, and the capacitor Cl of resonance in series. The switching elements are controlled such that, when the resonant current II decreases to zero, the switching element Q2 is switched on and the switching element Q1 is switched off, then the resonant current II flows in the reverse direction to through the switching element Q2, Next, a self-excited switching operation, wherein the switching elements Ql and Q2 are connected alternately, is performed, As the switching elements Ql and Q2 repeat disconnect connection operations using the terminal voltage of the filtering capacitor Ci as an operating power supply, the driving current having a waveform close to a current waveform The resonance is supplied to the primary winding NI of the PIT transformer of the isolation converter while an alternate output is obtained in the secondary winding N2. The constant voltage control by the PRT drive transformer is done in the following way. The control circuit 9 controls the level of a control current flowing through the control winding NC to increase in response to an increase in the DC output voltage EOl of the secondary side. While the PRT drive transformer is inclined to approach saturation due to the influence of the magnetic flux variation generated in the driving PRT transformer and thus dropping the inductance of the NB1 and MB2 drive windings, the condition of the self-excited oscillation circuits is varied by the control current so that the switching frequency can be increased, while the switching frequency in the power supply circuit 900 is adjusted on a frequency scale higher than the frequency of resonance of the series resonance circuit of series capacitor Cl and the primary winding NI (upper side control), if the switching frequency is raised as described above, then the switching frequency is spaced far from the frequency resonance of the series resonance circuit., the resonance impedance of the resonance circuit in series with respect to the switching output increases. As the resonance impedance increases in this manner, the driving current to be supplied to the primary winding NI of the primary-side series resonance circuit is suppressed. As a result, the secondary side output voltage EOl is suppressed, and consequently, constant voltage control is achieved (switching frequency control system), Figures 10A to 10H are waveform diagrams illustrating circuit operations. 900 power supply. Particularly, Figures 10A to 10D show operating waveforms of different portions of the power supply circuit 900 when the charge Po energy is at a minimum charge energy (Pomin) and the input VAC dump AC is at a maximum AC input power (VACmax), and Figures 10E to 10H show operation of waveforms of the same portions as those of Figures 10A to 10D when the charge Po energy is at a maximum charge energy (Pomax) and the AC input voltage VAC is at a minimum AC input voltage (VAC'min), As the switching element Q2 performs a switching operation, the voltage Vcp, obtained between the collector and the emitter of the element Switching Q2, has a waveform that exhibits the level of zero during a TON period within which the switching element Q2 is connected, but which exhibits a pulse of a rectangular waveform during a period TOFF d in which the switching element Q2 is disconnected, as shown in Figures 10A and 10E. In addition, as can be recognized from a comparison of the voltages Vcp shown in Figures 10A and 10E, the switching frequency is controlled by the constant voltage control operation described above so that it is higher when the charging power Po is at Minimum load Pomin energy and AC input voltage VAC is at maximum AC input voltage VACmax than when load Po energy is at Pomax maximum load power and AC input voltage FAC is at VACmin power minimum AC input, In this case, the collector current Icp flowing to the collector of the switching element Q2 exhibits a waveform such that it flows to the collector of the switching element Q2 during the TON period, but has the level of zero during the TOFF period, as shown in Figures 10B and 10F. In the meantime, the switching output current (primary-side series resonant current) II flowing through the primary winding NI and the series resonance capacitor Cl exhibits a current waveform substantially corresponding to the switching frequency. , as shown in Figures 10C and 10G. As shown in Figures 10B and 10C, the collector current Icp and the primary resonant current GT primary current II have waveforms of a sine wave corresponding to the type of current resonance when the switching frequency is low. As the switching frequency increases, the waveforms of the collector Icp current and the primary resonance current II of the primary side approach waveforms of a sawtooth wave, as shown in the Figures 10F and 10G. It is noted that the switching element Q2 provides waveforms having phases shifted by 180 degrees from the waveforms shown in Figures 10A to 10C and 10E to 10G, on the other hand, on the secondary 915 side, the rectifier diode D02 it becomes substantially conductive in a TON period timing within which the switching element Q2 is connected.

Consequently, the rectification current 12 flowing from the secondary winding N2 to the rectifying diode D02 exhibits a waveform as shown in Figure 10D or 10H. In addition, the operation of the rectifier DOl diode provides a waveform having a displacement of 180 degree phase of the waveform shown in Figure 10D or 10H. With the power supply circuit 700 in the RCC system. The switching frequency rs is controllably varied as a constant voltage control operation as described above. However, since the control sensitivity for the constant voltage control is low, the scale of variation of the switching frequency fs is comparatively wide from 25 KHz to 250 KHz as described above. Therefore, when the charging energy Po is converted into minimum load Pomin energy and the switching frequency fs becomes low, the switching loss increases and the energy conversion efficiency drops significantly. In addition, the energy loss is increased by the reset circuit 10 and the protection circuit 11, which are connected in order to suppress a peak voltage upon disconnection of the switching element. Furthermore, as seen in Figure 8A, the alternating voltage that is generated by a switching operation is a pulse signal of a rectangular waveform, and the switching noise is generated on and off. Therefore, in order to allow the power supply circuit 700 of the RCC system to be used practically as the supply d? energy, for example, for a video device, the charging power condition is at about 1 W or less, while the application of the power supply circuit 700 of the RCC system is limited to one application as a power supply of wait, whose charge Po energy is approximately 0.5 W or less. On the other hand, in the switching power supply circuit 900, a current resonance converter via a half-bridge connection in which the capacitors for partial resonance are connected between the collector and the emitter of two switching elements is provided to In order to perform a zero volt switching operation (zero voltage) when disconnecting a switching element. Therefore, the switching power supply circuit 900 generates less noise than the circuit 700 d? power supply and has a higher energy conversion efficiency. However, the switching power supply circuit 900 does not overcome all the disadvantages of the 700 circuit, ie, the invalid power increases when the charging power Po approaches minimum load Pomin energy and the switching frequency fs. falls, and the energy conversion efficiency drops signifi- cantly to, for example, approximately 60%. Further, since the power supply circuit 900 adopts a construction of a self-excited current resonance converter in which two switching elements are connected in a medium connection, it requires the formation of a switching circuit system that includes two self-excited oscillation drive circuits. Consequently, the number of components increases so much, resulting in a limitation in the reduction in size and weight of a power supply circuit board. Additionally, the power supply circuit 900 is not provided with a protection function of short circuit charging. In particular, during the short-circuit of the load, the control current (amount of DC current) flowing through the control winding NC of the drive transformer PRT is substantially reduced to zero. Consequently, the switching frequency fs decreases almost to a lower limit of the control scale, and also the current II of resonance in series d? primary side flowing through the primary side series resonance circuit is inclined to increaseIn this state, the heat generation due to the switching loss in the switching elements Ql and Q2 increases to a level that can not be ignored, and in accordance with the circumstances, there is the possibility that a thermal and thermal leakage may occur. destroy the switching elements Ql and Q2. Therefore, for example, as shown in Figure 9, it is necessary to interpose the IL fuse to cut the DC output voltage EOl from the secondary side and the load from the other when the short circuit occurs with the load. The IL fuse also increases the size of the circuit board and decreases the energy conversion efficiency.

Statement of the Invention An object of the present invention is to provide a switching power supply circuit that is small, d? light weight, and highly efficient in energy conversion while satisfying a comparatively low load condition such as the charge Po energy, for example, is 50 W or less. In order to achieve the object described above, a switching power supply circuit in accordance with the present invention comprises a rectifier filtering circuit for receiving a commercial AC power supply, which produces a rectified filtering voltage of an equal level. at a commercial AC power supply level, and outputting the rectified filtered voltage as a DC input voltage, an isolation converter transformer that includes a core that has no space formed therein such that an efficiency of desirable coupling can be obtained, and a primary winding and a secondary winding wound on the core with polarities such that an additive mode of operation is provided; a switching circuit including a switching element for switching the DC input voltage on and off in such a way as to output to the primary winding of the isolation converter transformer, a primary side parallel resonance circuit formed of a leakage inductance component of the primary winding of the insulating converter transformer and a capacitance of a parallel resonance capacitor to operate a switching element of the voltage resonance type, a DC output voltage production circuit to receive an alternating voltage obtained in the secondary winding of the insulating converter transformer and performing a half-wave rectification operation for the alternating voltage to produce a secondary side DC output voltage substantially equal to the level of the DC input voltage; and a constant voltage control circuit for varying the switching frequency of the switching element in response to a level in the secondary side DC output voltage for controlling a resonance impedance of the primary side parallel resonance circuit and an angle of continuity of the switching element in order to perform a constant voltage control of the secondary side output voltage. Preferably, in order to satisfy a higher load energy condition than a particular level, a secondary side parallel resonance capacitor is connected in parallel to the secondary winding of the insulator converter transformer so that a secondary side parallel resonance circuit is formed of a leakage inductance component of the secondary winding of the insulating converter transformer and a capacitance of the secondary side parallel resonance capacitor. To satisfy a load energy condition less than a particular level, the secondary side parallel resonance capacitor is omitted. Preferably, the switching circuit includes a series resonance circuit formed of a series connection of at least one drive winding and a resonance capacitor. The switching power supply circuit comprises a self-excited oscillation drive circuit for driving the switching element in a self-excited manner in response to a resonance output of the series resonance circuit. The constant voltage control circuit includes an orthogonal control transformer as a saturable reactor in which the sensing winding and the drive winding connected in series to the primary winding of the insulating converter transformer and a control winding whose winding direction TS orthogonal to that of the sensing winding and the drive winding are wound up The constant voltage control circuit supplies a control current, which varies in response to a secondary side DC output voltage level, to the control winding to vary an inductance of the drive winding to controllably vary the switching frequency.

Preferably, the detection coil and the drive coil are formed of the same type of material as that used for the control coil. The switching power supply circuit can be constructed so that the circuit d? commutation includes a separately excited drive circuit for driving the switching element in a separately energized manner, and a constant voltage circuit controllably varies a switching element's switching period while maintaining a switch-off period of the fixed switching element in response to a level of output voltage d? Secondary side DC to controlably vary the switching frequency. The switching element of the switching circuit can be formed of a Darlington circuit that includes a bipolar transistor, a MOS field effect transistor, a bipolar isolated gate transistor or an electrostatic induction thyristor. In the switching power supply circuit, a voltage resonance converter is provided on the primary side, and the isolation converter transformer has a loose coupling. On the secondary side, the secondary side DC output voltage is produced by a half-wave rectifier circuit to supply power to a load. Furthermore, in the construction for constant voltage control, the resonance impedance of the primary-side parallel resonance circuit and the continuity angle of the switching element are simultaneously controlled by varying the switching frequency in response to the DC output level of the ladc. In this way, the increase in control sensitivity is achieved by said compound control operation. More particularly, when the switching power supply circuit is formed, for example, in order to satisfy the comparatively low charging condition of having a charging power of about 50 W or less, it includes a voltage resonance switching converter. provided on the primary side and an isolation converter transformer, including a primary winding and a secondary winding which are wound so that the mutual inductance therebetween can provide an additive mode of operation (- + M, forward system) . In addition, a half-wave rectifier circuit is provided on the secondary side so that a secondary side DC output voltage can be obtained from an alternating voltage (excited voltage) obtained in the secondary winding by a half-wave rectification operation of the half wave rectifier circuit in the additive operation mode. Further, as a construction for constant voltage control to stabilize the secondary output voltage, the switching frequency of the primary side is varied in response to the output voltage level of the secondary output, thereby controlling in this way the impedance of the secondary voltage. resonance of the power supply circuit and the continuity angle of the switching element. From the construction described above, the following advantages can be achieved, since the switching converter of the switching power supply circuit is of the voltage resonance type, a switching operation that produces less noise than a power supply RCC system switching is performed. Consequently, as opposed to a switching power supply of the RCC system, a power supply circuit according to the present invention does not require a reset circuit or a protection circuit to suppress a peak voltage. Accordingly, when compared to a switching power supply of the RCC type, the switching power supply circuit of the present invention achieves significant improvement in energy conversion efficiency. The switching power supply circuit of the present invention also exhibits significant improvement in energy conversion efficiency, when operating with maximum load energy, on a current resonance converter, which is considered to have a comparatively high energy conversion efficiency of its characteristics. A current resonance converter is formed of a half-bridge connection of two switching elements. Since the switching power supply circuit of the present invention can be constructed so that substantially equal charge energy is obtained using a single switching element because it is a resonance converter d? voltage, the number of parts is greatly reduced, and the reduction in size, weight and cost of the circuit can be promoted. Additionally, since the switching power supply circuit d of the present invention is constructed so as to vary the switching frequency to control both the resonance impedance for the switching output and the continuity angle of the switching element, performing This way constant voltage control, the control sensitivity is improved and the controllable scale is expanded, Consequently, the stabilization of the output voltage of the secondary side can be achieved on a narrower control scale of the switching frequency. This reduction in the control scale of the switching frequency contributes to the reduction of the number of turns that need to be wound up in the transformer that forms the power supply circuit and the miniaturization in size of various components and devices. When a self-excited circuit system for driving the switching element is provided in the constant voltage control circuit, an orthogonal control transformer is provided in which a control winding, an impulse winding, and a detection winding are wound. . In this case, if the detection winding and the drive winding are formed from the same type of material as is used for the control winding, then the production efficiency of the orthogonal control transformer is improved, when a resonance capacitor secondary side parallel is connected in parallel to the secondary winding to form a parallel resonance circuit, the half-wave rectifier circuit of the secondary side receives an alternating voltage, which is a resonance output of the parallel resonance circuit, to obtain a voltage of DC output from the secondary side, consequently, the load energy increases. In other words, the switching power supply circuit can deal with a higher Charge energy than a particular level by merely inserting the secondary side parallel resonance capacitor. When the switching power supply circuit is applied to an application such as, for example, a power supply d? If you wait and only require to deal with a charge energy lower than the particular level, the secondary side parallel resonance capacitor can be omitted. In this way, adjustments in accordance with a required charge energy condition can be made by simply inserting or removing the secondary side parallel resonance capacitor, in addition, since the parallel resonance circuit is provided on the secondary side, the voltage Parallel resonance is obtained on the secondary side even when the short circuit of the load occurs. In this way, the switching frequency does not fall even during load short-circuiting. Briefly, the switching power supply circuit has a protection function against load short circuit. Consequently, with the switching power supply circuit of the present invention, the need to insert an IC link fuse or the like into the secondary side output is eliminated. Consequently, the improvement in energy conversion efficiency and reduction in size and weight of the circuit can be promoted. In the switching power supply circuit, the switching element can be formed of a Darlington circuit that includes a bipolar transistor, a MOS field effect transistor, a bipolar isolated gate transistor, or an electrostatic induction thyristor. Therefore, the energy conversion efficiency can be further improved over, for example, a single bipolar transistor switching element. In this way, in accordance with the present invention, the reduction in cost. size, and weight, and improvement of such characteristics as energy conversion efficiency are promoted for a power supply circuit that includes a voltage resonance converter on the primary side and is ready for a comparatively low load.

The foregoing and other objects, features and advantages of the present invention will become apparent from the following description and the appended claims, taken in conjunction with the accompanying drawings in which like parts or elements are denoted by like reference symbols.

BRIEF DESCRIPTION OF THE DRAWINGS For a more complete understanding of the invention, reference is made to the following description and accompanying drawings, in which Figure 1 is a circuit diagram showing a construction of a power supply circuit. to which the present invention is applied; Figure 2 is a perspective view showing a structure of an orthogonal control transformer provided in the power supply circuit of Figure 1; Figure 3 is a perspective view showing a structure of an insulating converter transformer provided in the power supply circuit of Figure 1; Figures 4A and 4B are circuit diagrams illustrating operations of the insulating converter transformer shown in Figure 3, when the mutual inductance between the windings is + M and -M, respectively. Figures 5A through 5J are waveform diagrams illustrating the operation of various components of the power supply circuit of Figure 1; Fig. 6 is a diagrammatic view illustrating a relation between a switching framing and a sequential-sided DC output voltage of the power supply circuit of Fig. 1; Figure 7 is a circuit diagram showing a construction of a conventional power supply circuit; Figures 8A to 8C are waveform diagrams illustrating multi-component operations of the power supply circuit of Figure 7, Figure 9 is a circuit diagram showing a construction of another conventional power supply circuit, and Figures 10A to 10H are waveform diagrams illustrating multi-component operations of the power supply circuit of Figure 9, BEST MODE FOR CARRYING OUT THE INVENTION Figure 1 shows a construction of a switching power supply circuit 100 in accordance with the present invention, as shown in Figure 1, the power supply circuit 100 includes various components in common with those of the 700 and 900 power supply circuits. It is noted that the description of said common components is omitted to avoid redundancy. Referring to Figure 1, the power supply circuit 100 includes, on its primary side 105, a self-excited switching converter of the voltage resonance type including a switching element Ql. A bipolar transistor (BJT: junction transistor) having a high voltage holding property ST adopts for the switching element Ql. The base of the switching element Ql is connected to the positive electrode of a filtration capacitor Ci (rectified filtering voltage Ei) through d? a series connection to a limiting resistor RB d? base current and a starting resistor RS so that the base current can be obtained from a rectifier filtering line during startup. In addition, a resonance circuit for self-excited oscillation drive (self-excited oscillation drive circuit) is connected between the base of the switching element Ql and ground and is formed of a series circuit connection that includes the current limiting resistor PB of base, a capacitor CB of resonance, and a winding NB of drive of detection. The switching element Ql is driven for switching with the driving current applied to the base thereof from the self-excited oscillation drive circuit after it starts with a starting current. A clamping diodes DD is interposed between the base of the switching element Ql and the negative electrode (primary side ground) of the filtering capacitors Ci and forms a path for damping current flowing when the switching element Q1 is disconnected. The collector of the switching element Ql is connected to one end of the primary winding NI of a transformer PIT of insulating converter, and the emitter of the switching element Ql is connected to ground. Consequently, a switching output of the switching element Ql is transmitted to the primary winding NI. A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Ql. The parallel resonance capacitor Cr forms, based on a capacitance of parallel resonance capacitor Cr and a leakage inductance Ll of the primary winding of the transformer PIT of the isolating converter, a primary-side parallel resonance circuit for the voltage resonance switching of the Ql switching element. Even when the detailed description is omitted here, when the switching element Ql is disconnected, the voltage resonance operation is obtained by the action of the parallel resonance circuit which causes the voltage seen through the resonance capacitor Cr to actually display an impulse wave of a sinusoidal waveform, An orthogonal control transformer PRT, as shown in Figure 1, is a saturable reactor including a detection winding ND, driving winding NB, and a control winding MC, As shown in Figure 2, the orthogonal control transformer PRT has a three-dimensional core 200 which is formed by joining two cores 201 and 202 of double channel shape, each having four magnetic legs, with each other at the ends of the magnetic legs thereof. The detection winding ND and the driving winding NB are wound in the same winding direction around two predetermined magnetic legs' of the three-dimensional core 200, and the control winding NC is wound in a direction orthogonal to that of the detection winding ND and drive winding NB In this case, the detection winding ND of the orthogonal control transformer PRT is interposed in series between the positive electrode of the filtering capacitor Ci and the primary winding NI of the PIT transformer of the insulating converter so that the output of switching of the switching element Ql transmits to the detection winding ND through the primary winding NI. In the orthogonal control transformer PRT, the drive winding NB is excited by a switching output obtained in the detection winding ND so that an alternating voltage is generated in the drive winding NB. The alternating voltage is output as a source of driving voltage to the self-excited oscillation drive circuit. A control circuit 1 shown in Figure 1 operates to vary the level of a control current (DC current) supplied to the NC winding of control in response to the level of a voltage EOl DC output voltage from secondary side to it. It is noted that the control circuit 1 may have said internal construction, for example, as that of the control circuit 9 described above with reference to the Figure, as the level of the control current (DC current) that to be supplied to the control winding NC is varied in response to a variation of the secondary side DC output voltage level by the operation of the control circuit 1, the inductance LB of the drive winding NB is wound up in the PRT transformer Orthogonal control is varied in a controllable manner. Consequently, the resonance condition of the circuit d? Series resonance in the self-excited oscillation drive circuit for the switching element Ql, including the inductance LB of the drive winding NB, varies. This is an operation of varying the switching frequency of the switching element Ql described below with reference to Figure 5, and this operation acts to stabilize the secondary side DC output voltage EOl. The PIT transformer of insulating converter of the power supply circuit 100 is shown in Figure 3. As illustrated in Figure 3, the transformer PIT of insulating converter includes a core 300 of EE shape formed d? two nuclei 301 and 302 of configuration in E made of a ferrite material and combined so that the magnetic poles thereof are opposite each other. The primary winding NI and the secondary winding N2 (and another secondary winding N2A) are wound separately from each other on the central magnetic legs of the EE-shaped core 300 using a split coil whose winding portion is divided for a primary side and a secondary side. According to one embodiment, the space between the central magnetic legs of the E-shaped cores 301 and 302 is not formed. Consequently, said loose coupling condition is established that a required saturation condition is obtained. The coupling coefficient k is, for example, k = 0.90. In this case, in the isolating converter PIT transformer, an alternating voltage is excited in the secondary winding N2 in response to a switching output transmitted to the primary winding NI. In the PIT transformer of the insulating converter of the power supply circuit 100, a bypass is provided for the secondary winding N2 as shown in Figure 1, and the anode of a rectifier diode DOl is connected in series to the bypass output of the rectifier. N2 secondary winding. The cathode - A A - The DOI diode rectifier is connected to the positive electrode of a COI filter capacitor, and the negative electrode of the filter capacitor COI is connected to ground. Briefly, the rectifier diode DOl and the filtering capacitor COI form a half wave rectifier circuit that receives an alternating voltage obtained from the bypass output of the secondary winding N2 and performs half wave rectification for the alternating voltage to obtain the voltage EOl of secondary side DC output. The DC output voltage EOl d? The secondary side is supplied to a load (not shown) and is also input as the detection voltage to the control circuit 1 described above. In addition, the anode of the rectifier diode D02 is connected to a winding start end of the secondary winding N2 while the cathode of the rectifying diode D02 connects the positive electrode of the filtration capacitor C02, thereby forming a rectifier circuit of medi. 3 composite wave of the DOl diode rectifier and the filtering COI capacitor. The half-wave rectifier circuit formed by the diode DOl rectifier and the capacitor COI of filtration produces another voltage E02 of secondary side DC output and supplies it as a power supply d? operation to control circuit 1 In the PIT transformer of insulating converter shown in Figure 3, a mutual inductance M between the inductance Ll of the primary winding NI and the inductance L2 of the secondary winding N2 could have a value + M (additive mode: system of advance) or another value -M (subtractive mode: recoil system) depending on the relationship between the polarities (winding directions) of the primary winding NI and the secondary winding N2 and the connection of the diodes DOl and D02 rectifier. For example, in an operation where the aforementioned components assume a connection configuration as shown in Figure 4A, the mutual inductance M is + M, but in another operation where the components assume said connection configuration as shown in FIG. Figure 4B, the mutual inductance M is -M, In circuit 100, the polarities of the primary winding NI and the secondary winding N2 exhibit the additive mode. Also, in the circuit 100 d? power supply, a secondary side parallel resonance capacitor C2 is provided for the secondary winding N2. In this way, a parallel resonance circuit of the leakage inductance L2 of the secondary winding N2 and capacitance of the secondary side parallel resonance capacitor C2 is formed. The parallel resonance circuit converts an excited alternating voltage in the secondary winding N2 into a resonance voltage. Consequently, the voltage resonance operation is performed on the secondary side 110. In this way, the power supply circuit 100 includes a parallel resonance circuit provided on the primary side 105 to perform a switching operation of the voltage resonance type, and another parallel resonance circuit provided on the secondary side and formed of the secondary winding N2 and the capacitor C2 of parallel resonance. It is observed that. In the present specification, a switching converter of a construction that includes resonance circuits for both the primary side and the secondary side in this manner is appropriately termed as "composite resonance switching converter". When the capacitor C2 of parallel resonance of the secondary side is provided for the secondary winding N2 in the manner described above, since the energy on the secondary side 110 during the rectification operation is increased by a resonance operation of the side parallel resonance circuit Secondary, the charge energy available with the power supply circuit can be increased For example, the power supply circuit 100 can deal with a charge power of 1 W to 50 W as a result of the insertion of the resonance capacitor C2 parallel of secondary side. Nevertheless, when the charging power condition is 1 W or less, such as when the power supply circuit 100 is used as a standby power supply, the secondary side parallel resonance capacitor C2 is not inserted to adjust the power of cargo. Figures 5A through 5J are waveform diagrams illustrating the operation of the power supply circuit 100. More particularly, Figures 5A to 5E show operating waveforms in different portions of the power supply circuit 100 when the charging power is at a maximum load power (Pomax = 50 W) and AC input voltage VAC. is at minimum AC input power (VACmin = 80 V), and Figures 5F to 5J show operating waveforms of the same portions as those of Figures 5A to 5E but when the energy d load is at a power of Minimum load (Pomin (Po = 0)) and AC input voltage VAC is at a maximum AC input voltage (VACmax). In addition, the indicated waveforms - 40 broken lines in Figures 5B, 5C, 5D and 5E illustrate operations where secondary side parallel resonance capacitor C2 is omitted. The waveforms indicated by solid lines in Figures 5B, 5C, 5D and 5E illustrate operations in which the secondary side parallel resonance capacitor C2 is connected. Since the switching element Ql performs a switching operation, a resonance voltage Ver appearing between the collector and the emitter of the switching element Ql has a waveform which exhibits, as seen in FIGS. 5A or 5F, the zero level during a TON period within which the switching element Ql is on but exhibits a pulse of a sinusoidal waveform during a TOFF period within which the switching element Ql is switched off. Furthermore, as can be recognized by comparison between the resonance See voltages illustrated in Figures 5A and 5F, the switching frequency is controlled to a higher value when the load energy is at minimum load energy (Pomin = 0) and the AC input voltage VAC is at maximum AC input voltage (VACmax = 288 V) qae when the charging power is at maximum load power (Pomax = 50 W) and the AC input voltage VAC is at power minimum AC input (VACmin = 80 V) for the constant voltage control operation described above. After this, the collector current Icp flowing to the collector of the switching element Ql exhibits a waveform so that it flows to the collector of the switching element Ql during the TON period but exhibits a level of zero during the TOFF period, as it is shown in Figures 5B and 5G. In addition, the switching output current, which flows through the primary winding NI, has an alternating current shape substantially corresponding to a switching frequency and has a waveform approximated to a sine wave by an action of the resonance circuit primary side parallel, The rectifying operation of the secondary side 110 of the power supply circuit 100 is an additive mode (advance system), as described above with reference to Figure 4. It is illustrated as an operation when the diodes DOl and D02 rectifiers are made conductive and rectification current 12 flows substantially corresponding to the period TON during which switching element Ql is connected, as shown in Figures 5E and 5J, meanwhile, a voltage V2 of parallel resonance secondary side generated in the secondary side parallel resonance circuit exhibits a mode waveform what. when the rectifying diodes DOl and D02 are not conducting (disconnected), it is a negative sine wave, but when the rectifying diodes DOl and D02 are conducting (connected), they are subject to the DC output voltage EO (EOl or E02) with positive polarity. It is observed that, when the capacitor C2 of parallel resonance is not inserted, when the load energy is at maximum load power Pomax is (Pomax = 50 W) and the AC input voltage VAC is at an AC input power. minimum VAC in = 80v, the switching operation waveforms (Icp, II, II, V2, and 12) vary as shown by the broken line waveforms of Figures 5B through 5E. As can be recognized from the above description, the power supply circuit 100 is formed as a composite resonance switching converter that includes a voltage resonance converter (parallel resonance circuit) provided on the primary side 105 and a circuit parallel resonance and a half wave rectifier circuit provided on the secondary side 110. Figure 6 illustrates a relationship between a switching frequency fs and the secondary side DC output voltage E0 (Epl) in the power supply circuit 100. In Figure 6, the abscissa axis indicates the commutation frequency and the ordinate indicates the EO voltage level of the secondary side DC outputAs can be seen from the d-resonance curves indicated by the solid lines, with the power supply circuit 100, for example, in order to stabilize the voltage EO the DC output gives the secondary side at a desired level (e.g. approximately 5 V) against a variation of the load or a variation of the AC input voltage VAC, the frequency fs d? Switching is controlled to be within a range of 100 KHz to 200 KHz (ie, a scale of 100 KHz). In contrast, for example, when the power supply circuit 700 is used, in order to convert the secondary side DC output voltage EOl to a constant voltage, the switching frequency fs must be controlled to be within the scale of, for example. 25 KHz to 250 KHz (that is, a 225 KHz scale), as described above. The reason that the frequency control scale fs of switching is reduced for the power supply circuit 100 will now be described. In the power supply circuit 100, the constant voltage control action is provided by controllably varying the switching frequency of the switching element Ql by an operation of the constant voltage control circuit system composed of the control circuit 1 and orthogonal control PRT transformer, as described above. This operation is also illustrated in Figures 5A to 5J. For example, as can be recognized from the comparison between the waveforms (See and Icp) of Figures 5A and 5B and Figures 5F and 5G, in order to vary the switching frequency, the power supply circuit 100 varies from controllable way the period TON during which the switching element Ql is connected while the period TOFF during which the switching element Ql is disconnected remains fixed, In other words, it can be considered that, as a control operation dß constant voltage of the power supply circuit 100, operates to controllably vary the switching frequency to effect the resonance impedance control for the switching output, and to simultaneously perform the continuity angle control (PWM control) of the Ql element of switching in a switching period, This composite control operation is performed by a single control circuit system. Actually, when Pomin = 0 and VAC = 288 V to which the operating waveforms shown in Figures 5F and 5G correspond, the TON period decreases in response to the switching frequency when Pomax = 50 W and VAC = 80 V to which correspond the operating waveforms shown in Figures 5A and 5B. Consequently, also the amount of current of current II flowing to the capacitor d- resonance converter of the capacitor Ci of filtering is limited as can be seen from a current transition II of that of Figure AC to that of Figure 4H. As a result, the ST control sensitivity improves. Also, in Figure 6, a parallel resonance frequency fol of the parallel resonance circuit d? The primary side and a parallel resonance frequency fo2 of the secondary side parallel resonance circuit with respect to the switching frequency fs are shown. Here, for example, if the inductances and the capacitances are selected such that the parallel resonance frequency ff and the parallel resonance frequency fo2 can be equal to each other at or around 80 KHz, as shown in Figure 6, then a operation that the resonance impedances of the two parallel resonance circuits are controlled simultaneously to controllably vary the secondary side output voltage is obtained by the switching frequency control operation (constant voltage control operation) described above. This operation also significantly improves the control sensitivity. Improving the control sensitivity in the manner described above, the substantial control scale is expanded with the power supply circuit 100. Consequently, the variation width of the switching frequency can be reduced when compared with those in the power supply circuits 700 and 900. An impulse obtained with the voltage See resonance, illustrated in Figure 5A or 5F. Within the period TOFF is generated because the pedagogy of the parallel resonance circuit on the primary side of the voltage resonance converter acts on the DC input voltage (rectified filtering voltage) Ei. An Lvcr level of the voltage impulse View resonance is represented by Lvcr = Ei. { l + (p / 2 (TON / TOFF).}. (1 wherein Ei is the rectified filtering voltage level and TOFF and TON are periods of time of a disconnected period and a period connected within the switching period of the switching element Ql, respectively. It is here assumed that the power supply circuit 100 is commonly used with 100 V AC and 200 V AC as commercial AC power supplies. When using 100 V AC (VAC = 80 V), the DC input voltage Ei (rectified filtering voltage) at 100 V, and where AC 200 V is used (VAC = 228 V), the input voltage DC (rectified filtered voltage) E is 400 V. In this way, the DC input voltage (rectified filtered voltage) Ei for 200 V AC varies within a scale of approximately 3.6 times that of the input Ei voltage of DC (rectified filtered voltage for 100 V AC, As described above, the constant voltage control of the power supply circuit 100 varies the switching frequency by controllably varying the period TON during which the switching element Ql is connected , while the TOFF period during which the switching element Ql is switched off ST remains fixed.In other words, the voltage supply circuit operates so that, as the DC input voltage Ei increases (rectified filtered voltage). ), decrease uy the TON period in the same amount. If this operation is made to correspond to the expression (1) provided above, even if the filtered Ei voltage rectified for AC of 200 V has a variation width of 3.6 times that for AC of 100 V, the Lvcr level of Resonance voltage does not increase in proportion to an elevation in the rectified filtering voltage, but the elevation ratio is suppressed, in fact, as seen in Figures 5A and 5F, when the AC input voltage VAC varies from VAC = 80 V to VAC = 288 V (that is, variation of rectified filtering Ei voltage), the Lvcr level of the resonance voltage See varies from Lvcr = 550 Vp to Lvcr = 715 Vp. In this way, the increasing level ratio Lvcr is suppressed to approximately 1.3 times. Therefore, for the switching element Ql and the parallel resonance capacitor Cr to which a resonance voltage pulse is applied, a device having a voltage support property against, for example, 900 V can selectively be used. Consequently, an economical device can be selectively used for the switching element Ql and the parallel resonance capacitor Cr. Particularly for the switching element Ql which is a bipolar transistor, a device having better characteristics in relation to the saturation voltage VCE (SAT), the time d? tSTG storage, decay time tf, hFE factor of current amplification, and so on can be used selectively, in addition if the same wire material is used for the NC control winding, the detection ND winding, and the NB winding of driving the orthogonal control transformer PRT provided in the power supply circuit 100, then the parts management and the production process are simplified and the production efficiency is improved. Additionally, since the power supply circuit 100 includes the secondary side parallel resonance circuit, a state where secondary side parallel resonance voltage V2 is produced can be obtained by a parallel resonance operation of the parallel resonance circuit of secondary side even when load short circuit occurs. Therefore, even when the secondary side DC output voltage EOl falls, for example, at 10 V of 15 V which may occur during ordinary operation, the secondary side output voltage EO1 to the control circuit 1 is can maintain In this way, the power supply circuit 100 is constructed so that, during the load short-circuit, an IC for error amplification in the control circuit 1 is short-circuited so that the DC current supplied to the control NC winding of the orthogonal control transformer PRT is maintained to prevent a drop in the switching frequency. . As a result, an increase of the primary resonant current II of the primary side and the collector current Icp flowing to the collector of the switching element Ql is suppressed and, thus, the thermal leakage of the switching element Ql is prevented. Briefly, the power supply circuit 100 has a load short-circuit prevention function in it and can continue its stable switching operation even during load short-circuiting. Consequently, the power supply circuit 100 eliminates the need for a protection part such as an IC link fuse, In an experiment to test the operation of the power supply circuit 100, an energy conversion efficiency of about 90% was obtained when the energy Po dß charge was Pomax = 50 W and the voltage VAC do AC input was VAC = 100 V. When the power P charge was Pomin = 10 W and the input voltage VAC AC was VAC = 240 V, an energy conversion efficiency of approximately 80% was obtained. In this way, the experiment shows that the energy conversion efficiency of the power supply circuit 100, particularly when the charge energy at the minimum charge energy, exhibits an approximately 20% improvement over the alternative power supply circuits. . While the power supply circuit 100 is constructed such that a self-excited voltage resonance converter is provided on the primary side 105, it can be modified, for example, to include an oscillation drive circuit in the form of an IC (integrated circuit) instead of the self-excited oscillation drive circuit so that the switching element of a voltage resonance converter is driven by the oscillation drive circuit. In this case, as a constant voltage control, a waveform drive signal produced by the oscillation drive circuit is controllably varied in response to the secondary side output voltage level EOl. In the control, the impulse signal waveform may be produced so that the TOFF period during which the switching element is disconnected is fixed, while the TON period during which the switching element is connected is decreased in response to an EOl level rise of secondary side output voltage. By the control just described, the power supply circuit 100 operates in a similar manner as described above with reference to Figures 5A to 5J. It should be noted that, when this separate excited construction just described is adopted, the orthogonal control transformer PRT is omitted. Furthermore, when the separately excited construction described above is adopted, it is possible to adopt, instead of the simple bipolar transistor (BJT) as an element Ql switching, a Darlington circuit where two bipolar transistors (BJT) are connected in a Darlington connection. It is also possible to use, instead of the simple bipolar transistor (BJT), a MOSFET (MOS field effect transistor); metal oxide film semiconductor), an IGBT (insulated gate bipolar transistor) or a SIT (electrostatic induction thyristor). When the Darlington circuit or one of the above mentioned devices is used as the switching element Ql, superior efficiency can be achieved. When any of the devices is used as switching element Ql, even when not shown, the construction of the drive circuit for the switching element Ql ST would be modified so as to satisfy a characteristic of the device which is actually to be adopted in place of the BJT. For example, if a MOSFET is used as the switching element Ql, then the drive circuit for the switching element Ql can be constructed to drive the switching element Ql in a separately excited manner. In this way it will be seen that the objects set forth above, between those facts evident from the above description, are efficiently achieved and, because certain changes can be made in the exposed constructions without abandoning the spirit and scope of the invention, it is intended that all matter contained in the above description and shown in the accompanying drawings be construed as illustrative and not in a limiting sense. It should also be understood that the following claims are intended to cover all the generic and specific features of the invention described in the present and all the declarations of the scope of the invention which, as a matter of language, can be said to remain therein, Industrial Application As described above, a switching power supply circuit in accordance with the present invention includes an insulating converter transformer, which is formed by a core with no space and a primary and secondary winding wound on the core so that the mutual inductance between the windings exhibits an additive mode. A half wave rectifier circuit ST provides on the secondary side of the circuit and performs a rectification operation in the additive mode to obtain a secondary side DC output voltage. In a constant voltage control circuit system for stabilizing the secondary side output voltage, the switching frequency of a switching element is varied in response to the secondary side output voltage level to control the resonance impedance of a parallel resonance circuit of the primary side and the continuity angle of the switching element in a composite manner. In this way, the switching power supply circuit that is small, d? Light weight, and highly efficient in energy conversion can be provided.

Claims

1. - A switching power supply circuit, comprising: a filtering rectifier for receiving an AC power supply, which produces and outputs rectified filtered DC input voltage having a level equal to that of the power supply of AC; an insulating converter transformer having a core with no space to obtain a predetermined coupling efficiency and a primary winding and a secondary winding wound on the core with polarities such that an additive mode of operation is provided; a switching circuit for switching the DC input voltage on and off and outputting the switched voltage to the primary winding of the isolating converter transformer, thereby creating an alternating voltage in the secondary winding of the isolating converter transformer; a primary-side parallel resonance circuit formed of a leakage inductance component of the primary winding of the isolating converter transformer and a capacitance of a parallel resonance capacitor for operating the switching circuit in a voltage resonance-type operation 1; a DC output voltage circuit to receive the alternating voltage in the secondary winding of the insulating converter transformer and perform a half wave rectification in the alternating voltage to produce a secondary side DC output voltage having a substantially equal level to that of rectified filtering DC input voltage; and a constant voltage control circuit for varying a switching circuit switching frequency in response to a secondary side DC output voltage level for controlling the resonance impedance of the primary side parallel resonance circuit and varying an angle of continuity of the switching circuit in order to perform constant voltage control of the secondary side DC output voltage.

2. A switching power supply circuit according to claim 1, further comprising a secondary side parallel resonance capacitor connected in parallel to the secondary winding of the insulator converter transformer so as to form a parallel resonance circuit secondary side of a leakage inductance component of the secondary winding of the insulating converter transformer and a capacitance of the secondary side parallel resonance capacitor in order to satisfy a load energy condition above a predetermined level. 3 - A switching power supply circuit according to claim 1, wherein the constant voltage control circuit comprises an orthogonal control transformer such as a saturable reactor having a detection winding and a drive winding connected in series to the primary winding of the insulating converter transformer and a control winding whose winding direction is orthogonal to those of the detection winding and the drive winding, and controllably varies the switching frequency by varying a control current to the control winding in response to a level of the secondary side DC voltage output in order to vary - an inductance of the drive winding, the switching circuit comprises a series resonance circuit formed of a series connection of at least the drive winding and a resonance capacitor, and the ci The power supply network includes .6 - furthermore a self-excited oscillation drive circuit for driving the switching circuit in a self-excited manner in response to a resonance output of the series resonance circuit 4. A switching power supply circuit according to claim 3, where the detection winding and the impulse winding are formed from the same type d? material used for the control winding. 5. A switching power supply circuit according to claim 1, wherein the switching circuit includes a separately excited driving circuit for driving the switching circuit in a separately excited manner, and the voltage control circuit The constant varies controllably in a period of the switching circuit while maintaining a period disconnected from the switching circuit set in response to a secondary side DC output voltage level to controllably vary the switching frequency. 6. A switching power supply circuit according to claim 1, wherein the switching circuit comprises a Darlington circuit, which includes a bipolar transistor, 7.- A switching power supply circuit in accordance with the Claim 1, wherein the switching circuit comprises a field MOS transistor. 8. A switching power supply circuit according to claim 1, wherein the switching circuit comprises a bipolar isolated gate transistor. 9. A switching power supply circuit according to claim 1, wherein the switching circuit comprises an electrostatic induction thyristor.