KR102409792B1 - Control device of permanent magnet synchronization electric motor, microcomputer, electric motor system, and driving method of permanent magnet synchronization electric motor - Google Patents

Abstract

The control device of the embodiment includes a current change amount calculating unit that calculates the amount of change in the current passed through the stator of the electric motor, a magnetic pole position estimation unit that estimates the magnetic pole position of the rotor from the estimated rotation speed of the rotor, and rotates the estimated magnetic pole position A position correction unit that corrects based on a parameter that determines the output torque of the former, a coordinate transformation unit that converts the current into a current on the dq axis based on the estimated magnetic pole position, d, q axis current command value and d, q axis A current control unit that controls the current to generate d and q-axis voltage command values based on each difference of current, and a modulation control unit that generates PWM signal command values of a plurality of phases based on the d and q-axis voltage command values and the estimated magnetic pole positions , a low-speed side error calculating unit that calculates a low-speed side position error based on the amount of current change on the dq axis obtained by coordinate transformation using the corrected magnetic pole position, a high-speed based on the induced voltage or rotor flux obtained based on the voltage equation of the motor A high-speed side error calculating unit for outputting the position error of the side is provided, and an adding unit for weighting and adding the position errors of the low-speed side and the high-speed side at a predetermined ratio, respectively, wherein the magnetic pole position estimating unit includes the speed of the rotor based on the addition result of the addition unit to estimate

Description

CONTROL DEVICE OF PERMANENT MAGNET SYNCHRONIZATION ELECTRIC MOTOR, MICROCOMPUTER, ELECTRIC MOTOR SYSTEM, AND DRIVING METHOD OF PERMANENT MAGNET SYNCHRONIZATION ELECTRIC MOTOR

An embodiment of the present invention relates to a control device for a permanent magnet synchronous motor, a microcomputer provided with the device, a permanent magnet synchronous motor and a system including the device, and a method of operating the permanent magnet synchronous motor.

For permanent magnet synchronous motors, a position sensor such as a resolver or an encoder is generally used because it is necessary to change the energization signal of the inverter to flow the current according to the magnetic pole position of the rotor. However, a magnetic pole position estimation method for driving a synchronous motor without a position sensor is desired from requests for system downsizing, cost reduction, energy saving and maintenance, and the like.

Conventionally, as a method of estimating the position of the magnetic pole of a permanent magnet synchronous motor, a method of estimating the position based on the difference in dq-axis inductance, that is, the salient polarity, is used in the medium-speed region from the stationary state. On the other hand, from the medium speed range to the high speed range, for example, a method of calculating an induced voltage or rotor magnetic flux proportional to the speed of the motor from the input voltage and current to the motor and estimating based on the induced voltage is widely used. .

Since these two types of estimation methods are applied in different speed ranges, in a system for driving an electric motor, it is necessary to switch these estimation methods according to the rotational speed. Japanese Patent Laid-Open No. 2002-51580 proposes a method of calculating an error amount according to a magnetic pole position for a low speed and a high speed, respectively, and estimating the magnetic pole position and speed based on the weighted sum of them. Japanese Patent Laid-Open No. 2003-299381 also proposes a method of estimating rotational speed by weighting and adding estimated error amounts for low speed and high speed under the name of a frequency hybrid machine.

In the method of combining the two estimation methods as described above, when each estimation method operates ideally, the position can be estimated without any problem. Patent Document 1 mentions a problem that occurs when the estimated angles obtained by the low-speed and high-speed estimation methods are weighted and added. As a countermeasure, there is proposed a method in which the weighted addition is performed at the time of the error amount prior to calculating the angle, rather than weighted addition of the estimated angle. However, even in this case, if each error amount is not an ideal characteristic, a problem may occur.

As an example, ideally, it is assumed that the error amount Err _low in the low-speed magnetic pole position estimation is expressed by Equation (1), and the error amount Err _high in the high-speed magnetic pole position estimation is expressed by Equation (2). Δθ is the axial error between the true magnetic pole position θ of the electric motor and the estimated magnetic pole position.

When these are added, for example, by weight of 0.5, it becomes like Formula (3).

This error amount Err _sum becomes zero at the point where the axial error Δθ becomes zero, as shown in FIG. 13 . Therefore, if the position is estimated so that the error amount Err _sum becomes zero using, for example, the PI controller, the magnetic pole position can be estimated with high accuracy.

만큼 어긋나서 검출되어 버린다고 하자.Here, it is considered that the error amounts Err _low and Err _high themselves contain errors with respect to the true magnetic pole position. For example, the error amount Err _low is higher than the axis error Δθ.

Assume that the deviation is detected as much.

In such a case, as shown in FIG. 14 , the angle at which the error amount Err _sum becomes zero and the angle at which the axial error Δθ becomes zero do not match. Therefore, when an angle is estimated based on Formula (5), an error will be included in the estimated position.

만큼 어긋나 있는 경우의 도 13 상당도이다.BRIEF DESCRIPTION OF THE DRAWINGS It is 1st Embodiment, Comprising: It is a functional block diagram which shows the structure of the control apparatus of a permanent magnet synchronous motor.
Fig. 2 is a functional block diagram showing the configuration of a magnetic pole position estimation unit.
3 is a diagram showing the relationship between the position estimation error Δθ and the d-axis induced voltage Edc.
Fig. 4 is a diagram showing a waveform of a three-phase PWM signal and a waveform of the U and V-phase currents.
5 is a diagram showing the relationship between the position estimation error Δθ and the amount of change in the q-axis current.
6 is a diagram showing the relationship between the output torque of the motor, the q-axis current Iq, and the correction angle θadd.
7 is a diagram showing the relationship between the motor speed ? and the low-speed side weight Klow and the high-speed side weight Khigh.
8 is a diagram showing the relationship between the position estimation error Δθ and the amount of change in the q-axis current during high-load operation.
9 is a diagram showing the configuration of the angle correction unit.
Fig. 10 is a diagram showing a change in each value when the motor performs a forward and reverse operation.
Fig. 11 is a functional block diagram showing the configuration of a control device for a permanent magnet synchronous motor according to the second embodiment.
Fig. 12 is a diagram showing the configuration of a low-speed side axial error calculating section.
13 is a diagram showing the relationship between the position estimation error Δθ and the error amount Err _sum in the prior art.
14 shows that the error amount Err _low is higher than the axial error Δθ.

It is a figure equivalent to FIG. 13 in the case where it shift|deviates by this.

According to each embodiment, the control apparatus of a permanent magnet synchronous motor which can estimate the magnetic pole position without an error, a microcomputer, an electric motor system, and the driving method of a permanent magnet synchronous motor are provided.

A control device for a permanent magnet synchronous motor according to an embodiment includes: a current change amount calculating unit for calculating a change amount of a current passed through a stator of the permanent magnet synchronous motor;

a magnetic pole position estimator for estimating the rotational speed of the rotor and estimating the magnetic pole position of the rotor from the estimated rotational speed;

a position correction unit for correcting the estimated magnetic pole position based on a parameter for determining the output torque of the rotor;

a coordinate conversion unit that converts the current into a current on the dq axis based on the estimated magnetic pole position;

a current control unit that controls the current to generate d and q-axis voltage command values based on the respective differences between the d and q-axis current command values and the d and q-axis currents;

a modulation control unit for generating PWM signal command values of a plurality of phases based on the d and q-axis voltage command values and the estimated magnetic pole positions;

a low-speed side error calculating unit for calculating a low-speed side position error based on the amount of current change on the dq axis obtained by coordinate transformation using the corrected magnetic pole position;

a high-speed side error calculating unit for outputting a high-speed side position error based on the induced voltage or rotor magnetic flux obtained based on the voltage equation of the synchronous motor;

an adding unit for weighting and adding the position error on the low-speed side and the position error on the high-speed side at a predetermined ratio, respectively;

The magnetic pole position estimation unit estimates the speed of the rotor based on the addition result of the addition unit.

The microcomputer of the embodiment is equipped with a control device for the permanent magnet synchronous motor of the embodiment.

The electric motor system of embodiment is equipped with a permanent magnet synchronous motor and the control apparatus of the permanent magnet synchronous motor of embodiment.

(First embodiment)

EMBODIMENT OF THE INVENTION Hereinafter, 1st Embodiment is demonstrated with reference to FIGS. 1-10. BRIEF DESCRIPTION OF THE DRAWINGS It is a functional block diagram which shows the structure of the control apparatus of a permanent magnet synchronous motor. The DC power supply 1 is a power source for driving the permanent magnet synchronous motor 2, and may be substituted for the one produced by performing AC → DC conversion from the AC power supply. The permanent magnet synchronous motor 2 is hereinafter referred to as a motor 2 . The inverter 3 uses, for example, an N-channel MOSFET 4 as a switching element, and is configured by connecting six FETs 4 to a three-phase bridge. The inverter 3 generates a voltage for driving the motor 2 based on six switching signals generated by the PWM generator 10 to be described later.

The current detection unit 5 is generally constituted by a circuit for processing a sensor signal output from the current sensor 6 using a shunt resistor, Hall CT, or the like, and detects three-phase currents Iu, Iv, and Iw. The three-phase/dq coordinate conversion unit 7 includes three-phase currents Iu, Iv, and Iw to the rotor of the motor 2 ; The coordinates are converted into the d-axis current Id and the q-axis current Iq by the angle estimated value θc according to the magnetic pole position of the rotor.

The current control unit 8 calculates d and q-axis voltages Vd and Vq so that the input current commands IdRef and IqRef of the d and q axes coincide with the currents Id and Iq of the d and q axes, respectively. The d-axis current command IdRef is set, for example, according to the case where the electric field operation or the weak field operation is performed from a higher level control device (not shown). In addition. The q-axis current command IqRef is generated, for example, according to the difference between the speed command ωRef given from the upper level control device and the estimated rotor speed ωc as will be described later.

The modulation control unit 9 coordinates transforming the d and q-axis voltages Vd and Vq into the three-phase voltages Vu, Vv, and Vw by the angle estimation value θc, and normalizes them by the voltage Vdc of the DC power supply 1 to modulate the three-phase Calculates the instructions Du, Dv, and Dw. The PWM generator 10 generates PWM signal pulses for each phase by comparing the three-phase modulation commands Du, Dv, and Dw with the carrier. A dead time is added to each pulse, and the switching signals U+, U-, V+, V-, W+, and W- of the three-phase upper and lower elements are generated, respectively. In addition, the PWM generator 10 shifts the phases of the PWM signal pulses of each phase, if necessary, so that a phase current change amount calculating unit described later can reliably detect changes in a plurality of phase currents.

The phase current change amount calculating section 11 detects the corresponding phase currents at two timings within the PWM period in order to obtain the three types of change amounts of the phase currents. In the present embodiment, the U-phase current change amount dIu_V5 while the voltage vector V5(001) is applied, the V-phase current change amount dIv_V5, and the V-phase current change amount dIv_V1 during the voltage vector V1 (100) application are obtained. The low-speed side axis error calculating unit 12 calculates the phase current change amounts dIu_V5, dIv_V5 and dIv_V1 obtained by the phase current change amount calculating unit 11 based on the estimated angle θc2 corrected by the later-described angle correction unit, the d-axis current change amount dId, the q-axis Coordinates are converted into current change amount dIq. Then, the change amount dIq is output as the low-speed side axis error Err _low . The high-speed side axial error calculating section 13 calculates the high-speed side axial error Err _high using the d and q-axis current/voltages Id, Iq, Vd, and Vq.

The weighting adding unit 14 adds the high-speed side axial error Err _high and the low-speed side axial error Err _low by a predetermined weighting, respectively, and calculates an error summation value Err _sum . The magnetic pole position estimating unit 15 uses the error summation value Err _sum output from the weighting and adding unit 14 to estimate the magnetic pole position, for example, as shown in FIG. 2 , PLL (Phase Locked) Loop) circuit or the like calculates the angle estimated value θc. Here, PI control calculation is performed using the proportional gain Kp and the integral gain Ki in the PI control unit 15a for the error addition value Err _sum to obtain the speed ωc. Then, the angle estimation value θc is calculated by performing an integration operation on the speed ωc by the integrating unit 15b. The angle correction unit 16 calculates an angle θc2 obtained by correcting the angle θc estimated by the magnetic pole position estimation unit 15 , and outputs it to the low-speed side axis error calculation unit 12 through the coefficient multiplier 17 .

In addition, in the structure shown in FIG. 1, the thing except the motor 2, the inverter circuit 3, and the current sensor 6 is the control apparatus 20, and the control apparatus 20 is comprised by the microcomputer. . In addition, the motor 2 and the control device 20 constitute an electric motor control system.

Here, the principle of the magnetic pole position estimation method in the present embodiment will be described. First, the principle of estimating the magnetic pole position of each of the low-speed region and the high-speed region will be described. In the position estimation in the high-speed region, an induced voltage generated with the rotation of the motor is used. Equation (6) is a voltage expression of the dq-axis of the permanent magnet synchronous motor in a steady state.

Ld, Lq: dq-axis inductance [H]

R: winding resistance [Ω]

φf: armature flux linkage by permanent magnet [Wb]

ω: motor speed [rad/s]

Here, the second term on the right side is an induced voltage term due to the rotation of the motor, and is generated only on the q-axis as shown in Equation (7).

Equation (7) corresponds to the true magnetic pole position of the motor, but when it is considered as a coordinate system θc shifted by the axial error Δθ here, the induced voltage term becomes as shown in Equation (8).

As shown in Fig. 3, the d-axis-side induced voltage Edc has a relationship of approximately monotonically increasing/decreasing about the zero point with respect to the axial error Δθ. From equation (9) modified from equation (6), the motor It is obtained from the constant and the detected voltage/current and used for position estimation.

ωc is the estimated speed, and Iqc is the current converted to the coordinate axis recognized by sensorless control.

Edc calculated in Equation (9) monotonically increases/decreases with respect to the axial error Δθ as the center of zero as described above. Therefore, position estimation becomes possible by determining the estimated speed ?c so that the induced voltage Edc becomes zero, and configuring a PLL (Phase Locked Loop) to obtain the estimated position ?c from the integral. FIG. 2 shows a PLL configured inside the magnetic pole position estimation unit 15 . In this embodiment, the induced voltage Edc is set to the error amount Err _high on the high-speed side.

Next, position estimation in the low-speed region will be described. The position estimation in the low-speed region is performed using the salient polarity in which the inductance changes according to the position of the magnetic pole of the motor. Equation (10) shows the characteristics of the three-phase inductance of the motor.

L0: constant inductance value regardless of angle [H]

L1: Displacement value of inductance that changes according to angle [H]

Since each phase inductance changes according to the position of the magnetic pole, the position is estimated using this characteristic.

Equation (11) expresses the characteristics of the change amount dIv_V1 of the V-phase current during application of the voltage vector V1 (100), the amount of change of the V-phase current dIv_V5 during the application of V5 (001), and the amount of change dIu_V5 of the V-phase current during the application of V5 (001), respectively. is indicating

dt: detection time of current change [s]

Vdc: DC voltage [V]

Paying attention to the right side, although the amplitude is different, the aspect of the phase change is the same as the three-phase inductance in Equation (10). Accordingly, these current variations are detected by the calculating unit 11 to obtain the magnetic pole position.

In this embodiment, in order to detect the amount of current change during application of these voltage vectors, for example, a PWM signal pattern as shown in FIG. 4 is used. In this case, the modulation control unit 9 shifts the pulse phase of the three-phase PWM signal, for example, as shown in the figure, in order to improve the detection rate of the current change amount. That is, using the peak value of the V-phase carrier, which is a triangular wave, as a reference phase, the upper pulse of the U phase extends the pulse to the slow side, and the upper pulse of the W phase extends the pulse to the leading side. The pulse on the upper side of the V phase uses the bottom value of the V phase carrier as the reference phase, so that the pulse is extended to both the slow and the leading side. If the current sensor 6 detects each phase current individually, such a phase shift process is unnecessary.

DELTA t shown in FIG. 4 is a period for sampling the amount of current change. First, in the voltage vector V1 (100) in which only the upper side of the U-phase is turned on, the amount of change in the current of the V-phase is sampled. Next, in the voltage vector V5 (001) in which only the upper side of the W phase is turned on, the amount of change in the U-phase current and the V-phase current is sampled. These sampled three types of current change amounts in Equation (11) are coordinate-converted into dId/dt and dIq/dt by Equation (13) at an angle that is twice the estimated angle θc.

Here, it is assumed that the estimated angle θc is shifted from the true position θ by an error Δθ as shown in equation (14).

By substituting the right side of Equation (11) for the change amount of three-phase current on the right side of Equation (13), and expanding the equation, dId/dt and dIq/dt expressed by Equation (15) are obtained.

As shown in Fig. 5, the q-axis term dIq/dt in Equation (15) has a relationship of approximately monotonically increasing/decreasing centered on zero with respect to the axial error Δθ. Therefore, similarly to the high-speed side, the magnetic pole position can be estimated by configuring the PLL as shown in Fig. 2 . In the present embodiment, dIq/dt is an error amount Err _low on the low-speed side. Further, as shown in FIG. 6 , the rate at which the q-axis current Iq increases as the output torque of the motor 2 increases and the rate at which the correction angle θ _add increases are approximately equal. Therefore, in the region where the driving state of the motor 2 is low-speed rotation and high torque, a reasonable amount of error is obtained by setting Err _low = dIq/dt.

Next, in the present embodiment, a method for estimating the magnetic pole position from the axial error amount on the high-speed side and the low-speed side will be described. The low-speed side error amount Err _low and the high-speed side error amount Err _high calculated as described above are added by using the high-speed side weight Khigh and the low-speed side weight Klow in the weighting adding unit 14, and expressed in Equation (16) Calculate the error amount Err _sum with characteristics. Since the polarity with respect to the shaft error is negative, the high-speed side error amount Err _high is multiplied by a coefficient "-1" and added.

The error amount Err _sum is input to the PI controller with zero as a command value, and the estimated position ?c is obtained from the estimated speed ?c and the integral thereof.

Here, in the characteristic with respect to the axial error Δθ of the error amount Err _sum shown on the right side of Equation (16), since the first term is an induced voltage term, it is proportional to the rotational speed ωc. For this reason, this term is set small in the low-speed region, and since the error and noise are large, the high-speed side weight Khigh is set small. When stopped, Khigh=0. On the other hand, the paired slow-side weight Klow is set to "1.0", which is the maximum value at the time of stopping. Thereafter, as shown in Fig. 7, as the speed ?c increases, Khigh increases and Klow decreases. In addition, the total value of both weights is always set to "1.0". By changing both weights in this way, sensorless control in the entire area from stationary to high-speed is possible.

In addition, in FIG. 7 , the detection rate of the current change amount is greatly improved after the point in time when Klow=Khigh=0.5 in the process of increasing the speed ?c. Accordingly, the modulation control unit 9 may stop the phase shift processing of the PWM signal pulses shown in FIG. 4 .

Next, the processing of the angle correction unit 16 for generating an angle used by the low-speed side axial error calculating unit 12 in order to calculate the low-speed side axial error Err _low will be described. On the low-speed side, the position is estimated using the salient polarity, which is the magnetic pole position dependence of the inductance explained in equations (10) and (11). In general, when the motor is operated with a light load, the relationship between voltage, current, and inductance described in equations (10) and (11) appears. However, it is known that the characteristic of the inductance due to the salient polarity changes due to magnetic saturation, interaxial interference of the dq axis, or the like during high load operation in which a large current is applied. Due to these influences, the phase characteristic of a sine wave with a frequency twice that of the magnetic pole position shown in equation (10) changes, and as shown in equation (17), there is a case where the phase shifts only by θe.

As a result, the characteristic of the q-axis current change amount dIq/dt calculated by the low-speed side axis error calculating unit also shifts as shown in Equation (18) and FIG. 8 .

If the magnetic pole position is estimated by the PLL based on the deviation dIq/dt shown in Equation (18), an error of θe will similarly occur in the estimated angle. An increase in the estimated angle error causes various problems such as an increase in the energized current, a decrease in efficiency, and a decrease in control stability.

For this reason, the angle correction unit 16 corrects the estimated angle θc estimated by the PLL in the magnetic pole position estimation unit as shown in Equation (19) and FIG. 9 . The corrected angle ?c2 is further multiplied by a coefficient "-2", and is input to the low-speed side axis error calculating section 12.

Here, since the correction value θe is an error generated when a large current is supplied, it is necessary to obtain the correction value θe in advance by measuring it by a test or the like. As a result of the coordinate transformation to the angle θc2 corrected by the angle correction unit 16, the characteristic of the q-axis current change amount dIq/dt becomes the same as in Equation (15), and the error due to the influence of magnetic saturation or the like is corrected.

Fig. 10 shows the change of each value in the case where the motor 2 is subjected to the forward reversal operation by the control of the present embodiment. The speed command is changed from negative to positive, and it is shifted from reverse to forward with a stop in the center of the drawing. In the low-speed region including the stop, the high-speed side weight Khigh is zero, and thereafter, it increases with the increase of the speed. The change pattern of the slow side weight Klow is the opposite. The true magnetic pole position θ and the estimated position θc almost coincide, and there is only a very minute difference before and after the rotation is stopped. According to the effect of the present embodiment, smooth switching from a stationary to a high-speed region and high-accuracy position estimation are realized.

As described above, according to the present embodiment, the current change amount calculating unit 11 calculates the change amounts dIv_V1, dIv_V5, and dIu_V5 of the three-phase current passed through the stator of the motor 2 , and the magnetic pole position estimating unit 15 rotates The rotational speed ωc of the electron is estimated, and the magnetic pole position θc of the rotor is estimated from the estimated rotational speed ωc. The angle correction unit 16 corrects the magnetic pole position θc based on a parameter that determines the output torque of the rotor. The three-phase/dq coordinate conversion unit 7 converts the three-phase current into a current on the dq-axis based on the magnetic pole position θc, and the current control unit 8 provides the d and q-axis current command values Idref, Iqref and d , and q-axis currents are controlled to generate d and q-axis voltage command values Vd and Vq based on the respective differences.

The modulation control unit 9 generates three-phase PWM signal command values Du, Dv, and Dw based on the magnetic pole position θc estimated by the d and q-axis voltage command values Vd and Vq, and generates a signal according to the PWM signal command value of each phase. The phase of the pulse is shifted so that the current change amount calculating section 11 can calculate the three-phase current change amount.

The slow-side axis error calculating unit 12 calculates the low-speed side position error Err _low based on the current change amount dIq/dt on the dq axis obtained by coordinate transformation of the change amount of the three-phase current by the corrected magnetic pole position θc2, , The high-speed side axis error calculating unit 13 outputs the d-axis induced voltage Edc obtained based on the voltage equation of the motor 2 as the high-speed side position error Err _high . The weighting and adding unit 14 weights and adds the position errors Err _low and Err _high at a predetermined ratio, respectively, and the magnetic pole position estimating unit 15 calculates the rotor speed ωc based on the addition result Err _sum estimate With this configuration, the magnetic pole position θc2 without error can be estimated. In particular, in the present embodiment, the magnetic pole position θc2 in which the error is excluded as much as possible can be estimated even if the error amount between the low speed side and the high speed side includes the error with the true position at different angles.

Further, since the weighting adder 14 changes the predetermined ratio according to the rotational speed ωc of the rotor, the ratio of the weight values of the position errors Err _low and Err _high is appropriately adjusted with the change of the speed ωc. can change

In addition, the modulation control unit 9 controls the phase of the signal pulse after the point in time when the weighting and adding unit 14 changes the predetermined ratio from the low-speed side to the high-speed side, the weight on the low-speed side and the weight on the high-speed side become equal. By stopping the shift, the control becomes simpler.

(Second embodiment)

Hereinafter, the same code|symbol is shown to 1st Embodiment and the same part, description is abbreviate|omitted, and the different part is demonstrated. The control device 21 of the second embodiment shown in FIG. 11 includes a three-phase/dq coordinate conversion unit 23 replacing the high-frequency voltage application unit 22 and the low-speed axis error calculating unit 12 and the low-speed side axis. An error calculating unit 24 is provided. In the second embodiment, the dq-axis high-frequency voltage application method is used for the position estimation method in the low-speed region.

The high-frequency voltage application unit 22 adds, to the d and q-axis voltage command values Vd and Vq output from the current control unit 8, the dq-axis high-frequency voltages Vd _h and Vq _h shown in Equation (20), an adder 25d, 25q) to overlap. V _h is the high frequency applied voltage amplitude and ω _h is the applied frequency.

At this time, the dq-axis currents Id2 and Iq2 coordinate-transformed by the three-phase/dq coordinate conversion unit 23 include information on the magnetic pole position under the influence of the salient polarity of the motor 2 . That is, the ω _h components Idh and Idq of Id2 and Iq2 have the characteristics shown in Formula (21).

By using this characteristic, the magnetic pole position θc can be estimated. Then, as shown in FIG. 12, the low-speed side axial error calculating part 24 of 2nd Embodiment performs a detection process using the band-pass filter 24a and the low-pass filter 24c. First, the dq-axis currents Id2 and Iq2 input from the three-phase/dq coordinate conversion unit 23 are passed through the bandpass filter 24a of the center frequency _ωh to extract the component of the frequency _ωh . Thereafter, the current Id2 is multiplied by cos(ω _h t) and the current Iq2 is multiplied by sin(ω _h t) by the multipliers 24bd and 24bq. Then, it passes through the low-pass filter 24c in which the cutoff frequency which can sufficiently remove the component of the frequency 2 omega _h is set. The axial error Err _low obtained by taking the difference between the two filter outputs by the subtractor 24d has the characteristic shown in Equation (22).

Although the axial error Err _low in Equation (20) has different coefficients, the axial error characteristic is the same as the q-axis current change amount dIq/dt in the first embodiment, so in the second embodiment, it is used for estimation. . Others are the same as in the first embodiment.

As described above, according to the second embodiment, the high frequency voltage application unit 22 superimposes the dq axis high frequency voltages Vdh and Vqh on the d and q axis voltage command values Vd and Vq output by the current control unit 8 . Then, the low-speed side error calculating unit 24 performs synchronous detection of the dq-axis currents Id2 and Iq2 coordinate-converted by the three-phase/dq coordinate conversion unit 23 with the high-frequency voltages Vd _h , Vq _h of the low-speed side from the result of synchronous detection. Generates position error Err _low . Therefore, the effect similar to 1st Embodiment is acquired.

(Other embodiments)

Although the method for estimating the position of the low-speed region and the high-speed region has been exemplified, methods other than those exemplified may be used.

In the second embodiment, the high-frequency voltage may be applied to only one of the d-axis and the q-axis.

The current detection unit may be a shunt resistor or a CT.

As the switching element, a MOSFET, an IGBT, a power transistor, a wide bandgap semiconductor such as SiC or GaN, or the like may be used.

Although several embodiments of the present invention have been described, these embodiments have been presented by way of example and are not intended to limit the scope of the invention. These novel embodiments can be implemented in other various forms, and various abbreviations, substitutions, and changes can be made in the range which does not deviate from the summary of invention. These embodiments and their modifications are included in the scope and gist of the invention, and the invention described in the claims and their equivalents.

Claims (9)

A current change amount calculating unit for calculating a change amount of the current passed through the stator of the permanent magnet synchronous motor;
a magnetic pole position estimator for estimating the rotational speed of the rotor and estimating the magnetic pole position of the rotor from the estimated rotational speed;
a position correction unit for correcting the estimated magnetic pole position based on a parameter for determining the output torque of the rotor;
a coordinate conversion unit that converts the current into a current on the dq axis based on the estimated magnetic pole position;
a current control unit that controls the current to generate d and q-axis voltage command values based on the respective differences between the d and q-axis current command values and the d and q-axis currents;
a modulation control unit for generating PWM signal command values of a plurality of phases based on the d and q-axis voltage command values and the estimated magnetic pole positions;
a low-speed side error calculating unit for calculating a low-speed side position error based on the amount of current change on the dq axis obtained by coordinate transformation using the corrected magnetic pole position;
a high-speed side error calculating unit for outputting a high-speed side position error based on the induced voltage or rotor magnetic flux obtained based on the voltage equation of the synchronous motor;
an adding unit for weighting and adding the position error on the low-speed side and the position error on the high-speed side at a predetermined ratio, respectively;
The magnetic pole position estimation unit estimates the speed of the rotor based on the addition result of the addition unit,
The low-speed side error calculating section outputs the q-side current change amount as a low-speed side position error, a control device for a permanent magnet synchronous motor. The control apparatus for a permanent magnet synchronous motor according to claim 1, wherein the modulation control unit shifts a phase of a signal pulse according to a PWM signal command value of each phase so that the current change amount calculating unit can calculate the change amount of the current. 3. The method of claim 2,
When the constant of the synchronous motor is '3', the modulation control unit shifts the phase of the signal pulse of the first phase in a direction with respect to the center of the PWM period, and shifts the phase of the signal pulse of the second phase to the center of the PWM period A control device for a permanent magnet synchronous motor, which shifts in a direction opposite to the direction as a reference, and bilaterally shifts a phase of a signal pulse of a third phase with respect to a starting point or an ending point of a PWM period. 4. The phase shift of the signal pulses according to claim 3, wherein the modulation control unit, when the addition unit changes the predetermined ratio from the low speed side to the high speed side, after a point in time when the weight on the low speed side and the weight on the high speed side become equal to each other, the phase shift of the signal pulse A control device for a permanent magnet synchronous motor that stops the motor. The method according to claim 1, further comprising: a high-frequency signal applying unit for applying a high-frequency signal to at least one of the d and q-axis voltage command values;
The low-speed side error calculating unit generates a low-speed side position error from a result of synchronous detection of at least one of the currents on the dq axis with the high-frequency signal. The control apparatus for a permanent magnet synchronous motor according to claim 1, wherein the adding unit changes the predetermined ratio in accordance with a rotation speed of the rotor or a rotation speed command value. A microcomputer in which the control device of the permanent magnet synchronous motor according to any one of claims 1 to 6 is mounted. permanent magnet synchronous motor,
The electric motor system provided with the control apparatus of the permanent magnet synchronous motor in any one of Claims 1-6. A current detection step of detecting a current flowing through a stator of the permanent magnet synchronous motor;
a current change amount calculation step of calculating the change amount of the current;
a magnetic pole position estimation step of estimating the rotational speed of the rotor and estimating the magnetic pole position of the rotor from the estimated rotational speed;
a position correction step of correcting the estimated magnetic pole position based on a parameter determining the output torque of the rotor;
a coordinate conversion step of converting the current into a current on the dq axis based on the estimated magnetic pole position;
a current control step of controlling the current to generate d and q-axis voltage command values based on the respective differences between the d and q-axis current command values and the d and q-axis currents;
a modulation control step of generating a 3-phase PWM signal command value based on the d and q-axis voltage command values and the estimated magnetic pole position;
A low-speed side error calculation step of calculating a low-speed side position error based on the amount of change in the q-axis current on the dq axis obtained by coordinate transformation using the corrected magnetic pole position, and an induced voltage or circuit obtained based on the voltage equation of the synchronous motor a high-speed side error calculation step of outputting a high-speed side position error based on the electromagnetic flux;
an adding step of weighting and adding the position error on the low-speed side and the position error on the high-speed side at a predetermined ratio, respectively;
and wherein the magnetic pole position estimation step estimates the speed of the rotor based on an addition result of the addition step.

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