KR102068180B1 - Apparatus for driving motor - Google Patents

Abstract

The present invention relates to a motor drive device. The motor drive device according to the embodiment of the present invention, the motor drive device includes a rectifier for rectifying the input AC voltage, an inverter for converting the rectified voltage to drive the motor, a capacitor disposed between the rectifier and the inverter, and in the inverter And a control unit for controlling the operation of the switching element, wherein the control unit includes a vector of the output current flowing through the motor and the output voltage when the output voltage command value vector for driving the motor exceeds an output voltage limit boundary region that can be output from the inverter. The final output voltage command value vector is calculated to be equal to the output power of the inverter based on the command value vector. Accordingly, overmodulation control can be performed in a capacitorless method.

Description

Apparatus for driving motor

The present invention relates to an electric motor driving apparatus, and more particularly, to an electric motor driving apparatus capable of performing overmodulation control in a capacitorless method.

Recently, in consideration of environmental problems, various methods for efficient power consumption in home appliances have been devised. Accordingly, the use of synchronous motors (SM), in particular, permanent magnet synchronous motors (PMSM) to which permanent magnets are applied, is increasing.
Recently, research on a capacitorless method using a small-capacity capacitor has been conducted in consideration of a manufacturing cost in an electric motor driving apparatus. There are disadvantages.

It is an object of the present invention to provide an electric motor driving apparatus capable of performing overmodulation control in a capacitorless method.

Motor driving apparatus according to an embodiment of the present invention for achieving the above object, the rectifier for rectifying the input AC voltage, an inverter for converting the rectified voltage to drive the motor, a capacitor disposed between the rectifier and the inverter, the inverter And a control unit for controlling the operation of the switching element in the control unit, wherein the control unit includes a vector of the output current flowing through the motor and the output when the output voltage command value vector for driving the motor exceeds an output voltage limit boundary region that can be output from the inverter. The final output voltage command value vector is calculated so that the output power of the inverter based on the voltage command value vector is not changed.

According to the embodiment of the present invention, when the generated first voltage command value exceeds the allowable voltage value, the electric motor drive device calculates an effective voltage command value based on the current command value and the first voltage command value, and calculates the calculated effective voltage command value. By outputting a switching control signal for controlling the inverter switching element on the basis of this, it is possible to maintain the magnitude of the inverter output power even when an overmodulation situation occurs.

On the other hand, according to this overmodulation technique, since the inverter can output the voltage, stable inverter control is possible.

In addition, the distortion of the inverter input current can be prevented and power factor deterioration can be minimized.

On the other hand, according to an embodiment of the present invention, the motor drive device generates an inverter output voltage command value based on the instantaneous power value based on the AC power source, and generates a switching control signal for driving the inverter based on the inverter output voltage command value. By outputting, in the capacitorless method, the input current control of the inverter can be performed in consideration of the pulsating dc terminal voltage.

In particular, when controlling a capacitorless inverter using a small capacitor, the inverter output voltage command value can be generated in consideration of the dc terminal voltage component even when the dc terminal voltage variation is large. do.

On the other hand, even in the field weakening control, the field weakening current command value is input to the voltage command generating section, and then the inverter output voltage command value is generated in consideration of the dc terminal voltage component, so that the field weakening control can be stably performed.

On the other hand, since it is possible to use a small capacity film capacitor with a long life, the reliability of the motor drive device is improved, the production cost and its volume is reduced.

1 is a circuit diagram showing an example of an electric motor drive device according to an embodiment of the present invention.
FIG. 2 is an internal block diagram of the controller of FIG. 1.
FIG. 3 is a diagram illustrating a dc terminal voltage according to the capacity of the capacitor of FIG. 1.
4 is a block diagram inside the controller illustrated for comparison with an embodiment of the present invention.
FIG. 5 is a block diagram of the control unit of FIG. 1 according to an exemplary embodiment of the present invention.
6 is an internal block diagram of the power command generation unit of FIG. 5.
7A to 7B are views referred to for describing an operation inside the controller of FIG. 5.
FIG. 8 is a diagram referred to describe the operation of the power controller of FIG. 5.
9A to 9B are diagrams for describing an overmodulation scheme according to the operation of the controller of FIG. 5.

Hereinafter, with reference to the drawings will be described the present invention in more detail.

The suffixes "module" and "unit" for components used in the following description are merely given in consideration of ease of preparation of the present specification, and do not impart any particular meaning or role by themselves. Therefore, the "module" and "unit" may be used interchangeably.

The motor drive device described in the present specification mainly describes a capacitorless type motor drive device. Capacitorless may mean using a small capacitance in a DC terminal.

1 is a circuit diagram showing an example of an electric motor drive device according to an embodiment of the present invention.

Referring to the drawings, according to an embodiment of the present invention, the motor driving apparatus 100 of FIG. 1 includes a rectifier 110, an inverter 120, a controller 130, an input current detector A, and a dc terminal voltage. The detector B, the DC terminal capacitor C, the motor current detector E, and the input voltage detector F may be included. In addition, the motor driving apparatus 100 of FIG. 1 may further include reactors L1 and L2.

The reactors L1 and L2 may be disposed between the commercial AC power supplies 105 and v _g and the rectifier 110 to perform an operation such as noise removal.

The input current detector A can detect the input current i _g input from the commercial AC power supply 105. To this end, a CT (current trnasformer), a shunt resistor, or the like may be used as the input current detector A. FIG. The detected input current i _g is a discrete signal in the form of a pulse and may be input to the controller 130 for power control.

The input voltage detector F can detect the input voltage v _g input from the commercial AC power supply 105. To this end, the input voltage detector F may include a resistor, an amplifier, or the like. The detected input voltage v _g is a discrete signal in the form of a pulse and may be input to the controller 130 for power control.

The rectifier 110 rectifies and outputs the commercial AC power supply 105 which has passed through the reactors L1 and L2. For example, the rectifier 110 may include a full bridge diode connected to four diodes, but various modifications are possible.

The capacitor C stores the input power. In the figure, one device is exemplified as the DC-terminal capacitor C, but a plurality of devices may be provided to ensure device stability.

Meanwhile, the DC terminal capacitor C in the present specification may be a capacitorless type capacitor as a small capacitor. That is, the capacitor C may be a film type capacitor, not an electrolytic capacitor. By using such a capacitorless type capacitor, the reliability of the electric motor drive device is improved, and the production cost and the volume thereof are reduced.

On the other hand, since the DC power is stored at both ends of the capacitor C, this may be referred to as a dc end or a dc link end.

The dc end voltage detector B may detect a dc end voltage Vdc that is both ends of the capacitor C. To this end, the dc terminal voltage detector B may include a resistor, an amplifier, and the like. The detected dc terminal voltage Vdc is a discrete signal in the form of a pulse and may be input to the controller 130 to generate the inverter switching control signal Sic.

The inverter 120 includes a plurality of inverter switching elements, and converts the rectified power supply Vdc into three-phase AC power supplies va, vb and vc of a predetermined frequency, and outputs them to the three-phase electric motor 150.

The inverter 120 has a pair of upper arm switching elements Sa, Sb, Sc and lower arm switching elements S'a, S'b, S'c, which are connected in series with each other, respectively, and a total of three pairs of upper and lower arms The switching elements are connected in parallel with each other (Sa & S'a, Sb & S'b, Sc & S'c). Diodes are connected in anti-parallel to each of the switching elements Sa, S'a, Sb, S'b, Sc and S'c.

The switching elements in the inverter 120 perform on / off operations of the respective switching elements based on the inverter switching control signal Sic from the controller 130. As a result, three-phase AC power supplies va, vb and vc having a predetermined frequency are output to the three-phase electric motor 150.

The controller 130 may control the switching operation of the inverter 120. To this end, the controller 130 may receive an output current i _o flowing through the motor detected by the motor current detector E. FIG.

The controller 130 outputs the inverter switching control signal Sic to the inverter 120 in order to control the switching operation of the inverter 120. The inverter switching control signal Sic is a PWM switching control signal and is generated and output based on the output current value i _o detected from the motor current detector E. FIG. Detailed operation of the output of the inverter switching control signal (Sic) in the control unit 130 will be described later with reference to FIG.

The motor current detection unit E detects the output current i _o flowing between the inverter 120 and the three-phase motor 150. That is, the current flowing through the three-phase motor 150 is detected. The motor current detector E may detect all of the output currents ia, ib, and ic of each phase, or may detect the output current of two phases by using three-phase equilibrium.

The motor current detector E may be located between the inverter 120 and the three-phase motor 150. For the current detection, a CT (current trnasformer), a shunt resistor, or the like may be used.

When a shunt resistor is used, three shunt resistors are located between the inverter 120 and the three-phase motor 150, or the three lower arm switching elements S'a, S'b, S'c of the inverter 120. It is possible to connect one end to each. On the other hand, it is also possible to use two shunt resistors using three-phase equilibrium. On the other hand, when one shunt resistor is used, the corresponding shunt resistor may be disposed between the above-mentioned capacitor C and the inverter 120.

The detected output current i _o is a discrete signal in the form of a pulse and may be applied to the controller 130, and the inverter switching control signal Sic is generated based on the detected output current io. do. In the following description, it is assumed that the detected output current i _o is the three-phase output currents ia, ib and ic.

On the other hand, the three-phase electric motor 150 includes a stator and a rotor, and an alternating current power of each phase of a predetermined frequency is applied to the coils of the stators of the phases (a, b, and c phases) so that the rotor rotates. Will be

Such three-phase motor 150 may be, for example, a surface-mounted permanent magnet synchronous motor (SMPMSM), an embedded permanent magnet synchronous motor (IPMSM), and a synchronous motor. Synchronous Reluctance Motor (Synrm) and the like.

Of these, SMPMSM and IPMSM are permanent magnet synchronous motors (PMSMs) with permanent magnets, while synrms have no permanent magnets.

Meanwhile, in accordance with an embodiment of the present invention, the controller 130 may include an input current i _g detected by the input current detector A, an input voltage v _g detected by the input voltage detector F, and dc. Power control is performed based on the dc end voltage Vdc detected by the end voltage detector B and the output current i _o detected by the motor current detector E.

FIG. 2 is an internal block diagram of the controller of FIG. 1.

Referring to FIG. 2, the controller 130 may include an axis converter 210, a position estimator 220, a current command generator 230, a voltage command generator 240, an axis converter 250, and The switching control signal output unit 260 may be included.

The axis converter 210 receives the three-phase output currents (ia, ib, ic) detected by the output current detector E, and converts the two-phase currents i α and i β of the stationary coordinate system.

On the other hand, the axis conversion unit 210 can convert the two-phase current (iα, iβ) of the stationary coordinate system into the two-phase current (id, iq) of the rotary coordinate system.

)에 기초하여, 추정된 속도(

)를 추청할 수 있다.The position estimating unit 220 receives the two-phase currents iα and iβ of the stationary coordinate system and the two-phase voltages vα and vβ of the stationary coordinate system, and estimates the rotor position θ. In addition, the position estimator 220 estimates the estimated position value (

Based on the estimated velocity (

) Can be requested.

At this time, the two-phase currents iα and iβ of the still coordinate system may be input from the axis conversion unit 210, and the two-phase voltages vα and vβ of the still coordinate system may be input from the dc stage from the dc stage voltage detector B. The voltage Vdc and the switching operation state of the inverter 120 may be calculated. For example, the three-phase output voltages (va, vb, vc) are calculated by a predetermined relation according to the dc end voltage (Vdc) from the dc end voltage detection unit (B) and the switching operation state of the inverter (120). In addition, the axis transformation unit 210 may be converted back to the two-phase voltage (vα, vβ) of the still coordinate system.

)와 추정된 속도(

)를 출력할 수 있다.On the other hand, the position estimating unit 220 estimates the position (

) And estimated speed (

) Can be printed.

)를 연산할 수 있다. 즉, 위치 신호에 기반하여, 시간에 대해, 나누면, 속도를 연산할 수 있게 된다. 이하에서는, 위치 추정부(220)를 중심으로 기술한다.Meanwhile, FIG. 2 illustrates a sensorless type position estimator 220 without a separate sensor for detecting a rotor position. However, in the case where a position sensing sensor such as a hall sensor is used, The government 220 can be replaced by a speed calculator (not shown). That is, when the position signal detected by the hall sensor is input to the speed calculator (not shown), the speed (

) Can be calculated. That is, based on the position signal, when divided over time, the velocity can be calculated. In the following description, the position estimation unit 220 will be described.

)와 연산된 속도(

)를 출력할 수 있다.On the other hand, the position estimator 220 calculates the position (calculated based on the input position signal H of the rotor)

) And computed speed (

) Can be printed.

)와 속도 지령치(ω^* _r)에 기초하여, 전류 지령치(i^* _q)를 생성한다. 예를 들어, 전류 지령 생성부(230)는, 연산 속도(

)와 속도 지령치(ω^* _r)의 차이에 기초하여, PI 제어기(235)에서 PI 제어를 수행하며, 전류 지령치(i^* _q)를 생성할 수 있다. 도면에서는, 전류 지령치로, q축 전류 지령치(i^* _q)를 예시하나, 도면과 달리, d축 전류 지령치(i^* _d)를 함께 생성하는 것도 가능하다. 한편, d축 전류 지령치(i^* _d)의 값은 0으로 설정될 수도 있다. On the other hand, the current command generation unit 230 has a calculation speed (

) And the current command value i ^* _q based on the speed command value ω ^* _r . For example, the current command generation unit 230 has a calculation speed (

) Based on the difference between the speed command value (ω ^* _r ) and the PI controller 235, the PI control may be performed, and the current command value i ^* _q may be generated. In the drawing, although the q-axis current command value i ^* _q is illustrated as a current command value, it is also possible to generate | generate a d-axis current command value i ^* _d unlike a figure. On the other hand, the value of the d-axis current command value i ^* _d may be set to zero.

On the other hand, the current command generation unit 230 may further include a limiter (not shown) for limiting the level so that the current command value i ^* _q does not exceed the allowable range.

Next, the voltage command generation unit 240 includes the d-axis and q-axis currents i _d and i _q which are axis-converted in the two-phase rotation coordinate system by the axis conversion unit, and the current command value (such as the current command generation unit 230). Based on i ^* _d , i ^* _q ), the d-axis and q-axis voltage command values v ^* _d and v ^* _q are generated. For example, the voltage command generation unit 240 performs PI control in the PI controller 244 based on the difference between the q-axis current i _q and the q-axis current command value i ^* _q , and q The axial voltage setpoint v ^* _q can be generated. In addition, the voltage command generation unit 240 performs PI control in the PI controller 248 based on the difference between the d-axis current i _d and the d-axis current command value i ^* _d , and the d-axis voltage. The setpoint (v ^* _d ) can be generated. On the other hand, the value of the d-axis voltage command value v ^* _d may be set to 0, corresponding to the case where the value of the d-axis current command value i ^* _d is set to zero.

On the other hand, the voltage command generation unit 240 may further include a limiter (not shown) for restricting the level so that the d-axis and q-axis voltage command values (v ^* _d , v ^* _q ) do not exceed the allowable range. .

On the other hand, the generated d-axis and q-axis voltage command values v ^* _d and v ^* _q are input to the axis conversion unit 250.

)와, d축, q축 전압 지령치(v^* _d,v^* _q)를 입력받아, 축변환을 수행한다.The axis transform unit 250 may calculate the position calculated by the position estimator 220 (

), And the d-axis and q-axis voltage command values (v ^* _d , v ^* _q ) are input, and axis conversion is performed.

)가 사용될 수 있다.First, the axis transformation unit 250 performs a transformation from the two-phase rotation coordinate system to the two-phase stop coordinate system. At this time, the position calculated by the position estimator 220 (

) Can be used.

In addition, the axis transformation unit 250 performs a transformation from the two-phase stop coordinate system to the three-phase stop coordinate system. Through this conversion, the axis conversion unit 1050 outputs the three-phase output voltage command values v ^* a, v ^* b, v ^* c.

The switching control signal output unit 260 generates an inverter switching control signal Sic based on the pulse width modulation (PWM) method based on the three-phase output voltage command values (v ^* a, v ^* b, v ^* c). To print.

The output inverter switching control signal Sic may be converted into a gate driving signal by a gate driver (not shown) and input to a gate of each switching element in the inverter 120. As a result, each of the switching elements Sa, S'a, Sb, S'b, Sc, and S'c in the inverter 120 performs a switching operation.

FIG. 3 is a diagram illustrating a dc terminal voltage according to the capacity of the capacitor of FIG. 1.

First, FIG. 3A illustrates the dc short voltage waveform Vdc1 when the electrolytic capacitor having a large capacity is used as the dc short capacitor C. FIG.

Next, FIG. 3B illustrates a dc short voltage waveform Vdc2 when a small film type capacitor is used as the dc short capacitor C. FIG.

As shown in FIG. 3B, when a capacitorless type capacitor is used, the capacitance is small, so that the smoothing function is lowered, and the variation of the dc terminal voltage is significantly larger than that in FIG. 3A.

Equation 1 below shows the relationship between the input power Pg, the dc stage power Pdc, and the inverter output power Pinv.

,

는 각각 전동기에 인가되는 전압 벡터와 전류 벡터를 의미하며, θ_vi는 전압 벡터(

)와 전류 벡터(

)의 위상 차이를 나타낸다.Where Cdc denotes a dc terminal capacitor,

,

Denotes a voltage vector and a current vector applied to the motor, respectively, and θ _vi denotes a voltage vector (

) And the current vector (

) Shows a phase difference.

In the case of a typical inverter, since the capacity of the dc terminal capacitor C of FIG. 3a is large enough to compensate for the difference in power, the desired inverter output power Pinv can be output without considering the input power Pg. .

However, as shown in FIG. 3B, when the small-capacity DC stage capacitor is used, the difference between the input power Pg and the inverter output power Pinv is because the size of the dc end power Pdc is limited by the capacity of the dc end capacitor C. Can't compensate enough.

In particular, in the case of using a single-phase input power supply, a large pulsation component of the input power Pg generated thereby remains largely in the inverter output power Pinv, which may cause pulsation of the motor output.

In the embodiment of the present invention, in a motor drive apparatus using a capacitorless type capacitor, a method of driving the motor in consideration of a large dc voltage (Vdc) having a high variability is proposed.

4A to 4C are diagrams for explaining an example of the overmodulation technique.

First, FIG. 4A is a diagram illustrating a pulse width modulation (PWM) scheme based on a space vector.

The a-axis, the b-axis, and the c-axis of the space vector area 310 correspond to the vector components ((1,0,0), (0,1,0), (0,0,1)) corresponding to three phases of the electric motor, respectively. Indicates.

The output voltage output from the inverter 120 can be output within the hexagonal space vector region 310. A voltage exceeding the space vector region, that is, a voltage exceeding a voltage output from the inverter 120 may be referred to as an overmodulation voltage.

As a technique for preventing such overmodulation, FIG. 4B or 4C may be illustrated.

First, the overmodulation prevention technique of FIG. 4B is an in-phase technique, while maintaining the same phase with respect to the overmodulation vector V_reference based on the first voltage vector V1 and the second voltage vector V2. Illustrates a technique for reducing the size of.

That is, the maximum output voltage of the inverter 120, that is, the voltage corresponding to the boundary point P1 of the space vector region 310 is output as the effective vector V_new.

In this case, since the effective vector is calculated by considering only the output voltage among factors related to the output power of the inverter 120, the output power control of the inverter 120 may not be accurately controlled, and thus the output power output from the inverter 120 may be distorted. It becomes possible.

Next, the overmodulation prevention technique of FIG. 4C is a minimum distance technique. For the overmodulation vector V_reference based on the first voltage vector V1 and the second voltage vector V2, the overmodulation vector V_reference is used. A method of setting the minimum distance point P2 of the space vector region 310 as the effective vector V_new is illustrated.

Vector phases are different, but voltage magnitude errors can be minimized. However, as shown in FIG. 4B, since the effective vector is calculated by considering only the output voltage among factors related to the output power of the inverter 120, the output power of the inverter 120 is not precisely controlled and the output output from the inverter 120 is performed. Power may be distorted.

Hereinafter, as an embodiment of the present invention, a method for controlling inverter output power in a capacitorless inverter control method having a large dc terminal voltage variation as a technique for preventing overmodulation will be described.

In addition, as an inverter control method based on a capacitor-less, power control is possible in consideration of the weak field control as well as the dc terminal voltage component, thereby reducing the power factor and limiting the motor operating area. It describes an inverter control method that can solve the problem.

FIG. 5 is a block diagram of the controller of FIG. 1 according to an exemplary embodiment of the present invention, FIG. 6 is a block diagram of the power command generation unit of FIG. 5, and FIGS. 7A to 7B illustrate operations of the controller of FIG. 5. FIG. 8 is a diagram referred to for describing the operation of the power controller of FIG. 5.

Referring to the drawings, the controller 130 of FIG. 5 includes a torque command generator 510, a power command generator 520, a power controller 525, a current command generator 530, and a voltage command generator 540. ), An adder 555, and a switching control signal output unit 560. Meanwhile, the axis converter 210, the axis converter 250, the position estimator 220, and the like described with reference to FIG. 2 may be further provided. In the following description, the units described in FIG. 5 will be described.

)와 속도 지령치(ω^* _r)에 기초하여, 전동기의 회전을 위한, 토크 지령치(T^*)를 출력할 수 있다. 특히, 토크 지령 생성부(510)는, 평균 토크 지령치를 출력할 수 있다. 한편, 연산 속도(

)는, 상술한, 전동기(150)에 흐르는 출력 전류(i_o), 또는 위치 신호에 기반하여, 연산될 수 있다.The torque command generation unit 510 has a calculation speed (

) And the torque command value T ^* for the rotation of the motor based on the speed command value (ω ^* _r ). In particular, the torque command generation unit 510 can output the average torque command value. On the other hand, computation speed (

) May be calculated based on the output current i _o flowing through the electric motor 150, or the position signal.

The current command generation unit 530 can generate the current command value I ^* based on the torque command value T ^* . Here, the current command value I ^* may mean a meaning including a d-axis current command value and a q-axis current command value of the fixed coordinate system.

The voltage command generation unit 540 may generate the first voltage command value V ^* ₁ based on the current command value I ^* and the output current flowing in the actual motor. Here, the first voltage command value V ^* ₁ may mean a value including a d-axis voltage command value and a q-axis voltage command value of the fixed coordinate system.

On the other hand, in relation to the field weakening current, the current command generating unit 530 can output the field weakening control current command value to the voltage command generating unit 540 during the field weakening control.

), 및 dc 단 전압 검출부에서 검출된 dc 단 전압(Vdc)에 기초하여, 출력 전력 지령치(P^*)를 생성하여 출력한다. 특히, 인버터(120)에서 출력 가능한 인버터 출력 전력 지령치(P^*inv)를 생성하여 출력한다. 본 명세서에서는, 전력 지령 생성부(520)에서 생성되는 출력 전력 지령치에 대해, P^* 와 P^*inv 를 혼용하여 사용하나, 그 의미는 동일하다.On the other hand, the power command generation unit 520 includes an input voltage Vg, a torque command value T ^* , and a calculation speed (

) And an output power command value P ^* based on the dc end voltage Vdc detected by the dc end voltage detector. In particular, the inverter output power command value (P ^* inv) that can be output from the inverter 120 is generated and output. In the present specification, with respect to the output power command value generated by the power command generation unit 520, P ^* and P ^* inv Is used interchangeably, but the meaning is the same.

)와, 계통 전압(Vg)의 위상을 이용하여, 입력 전력의 순시치(P^*g)를 산출한다.Specifically, the power command generation unit 520 includes a torque command value T ^* which is the output of the torque command generation unit 510 and the calculated current motor speed (

) And the instantaneous value P ^* g of input electric power is computed using the phase of system voltage Vg.

On the other hand, in order to drive the capacitorless inverter, the power change caused by the dc stage voltage variation should be taken into consideration, so that the power command generation unit 520 has an instantaneous value of the dc stage power in addition to the instantaneous value of input power P ^* g. Calculate (P ^* dc) Then, electric power command generation unit 520, by using the instantaneous value (P ^* g) and the instantaneous value (P ^* dc) of the dc terminal power of the input power, and generates the inverter output power command value (P ^* inv) .

Referring to FIG. 6, the input voltage Vg detected by the input voltage detector F is transmitted to the zero crossing detection unit 605 and the phase detection unit PLL 607 in the power command generation unit 520. Can be entered.

The zero crossing point detected by the zero crossing detection unit 605 may be input to the torque command generation unit 510 and used.

The phase detection unit PLL 607 detects the phase θg using the input voltage Vg detected by the input voltage detection unit F. FIG. The detected phase θg is used in the first unit 609.

)를 입력받아, 이를 승산한다. 이에 의해, 입력 전력(P'g)이 연산될 수 있다. 제3 유닛(612)에 입력된다.On the other hand, the second unit 611 has a torque command value T ^* and a calculation speed (

) And multiply it. Thereby, the input power P'g can be calculated. It is input to the third unit 612.

The third unit 612 multiplies the calculated input power P′g by the sinusoidal function 2sin ² (θg) output from the first unit 609. As a result, the input power instantaneous value P ^* g is calculated.

On the other hand, the fourth unit 614 generates a dc short power instantaneous value P ^* dc calculated by the capacitance Cdc of the capacitor C and the dc short voltage Vdc detected by the dc short voltage detector. To print.

The fifth unit 616 subtracts the dc terminal voltage command value P ^* dc from the instantaneous power instantaneous value P ^* g. Then, the difference between the input power instantaneous value P ^* g and the dc short power instantaneous value P ^* dc, that is, the inverter output power command value P ^* inv is output. The inverter output power command value P ^* inv can also be called inverter power instantaneous value.

Equation 2 below illustrates a method of calculating the inverter output power command value P ^* inv calculated in the power command generation unit 520 of FIG. 6 described above.

Here, Vg denotes an input voltage, Ig denotes an input current, Vdc denotes a dc terminal voltage, and Cdc denotes a capacitance of a capacitor (C). And wgt corresponds to the phase (theta) g mentioned above.

The power command value P ^* output from the power command generation unit 520 of FIG. 5 and the inverter output power command value P ^* inv of FIG. 6 mean the same value.

Next, the power controller 525 performs power control based on the input inverter output power command value P ^* inv.

The power controller 525 may generate the second voltage command value V ^* ₂ based on the inverter output power command value P ^* inv. Here, the second voltage command value V ^* ₂ is a compensation voltage command value for compensating the first voltage command value V ^* ₁ for which the dc terminal voltage is not considered.

The adder 555 adds and outputs the 1st voltage command value V ^* ₁ and the 2nd voltage command value V ^* ₂ . That is, the inverter output voltage command value V ^* 3 is output as the third voltage command value.

The switching control signal output unit 560 generates and outputs a switching control signal based on the third voltage command value V ^* 3.

In order to explain a specific operation of the power controller 525, a description will be given with reference to FIGS. 4A and 7A to 7B. First, FIG. 4A is a diagram illustrating a pulse vector modulation method based on a space vector. to be.

The a-axis, the b-axis, and the c-axis of the space vector area 310 correspond to the vector components ((1,0,0), (0,1,0), (0,0,1)) corresponding to three phases of the motor, respectively Indicates.

Meanwhile, the magnitude of the inverter power Pinv may be determined as an inner product of the motor output current vector and the inverter output voltage vector. Accordingly, the motor current vector or the inverter output voltage vector can be adjusted to obtain the desired inverter power.

Among these methods, the method of adjusting the motor output current vector may not quickly follow the inverter output power command value due to the delay occurring in the voltage command generation unit 540. In addition, in a given motor output current situation, in order to match the magnitude of the required inverter output voltage and the magnitude of the inverter power, the magnitude of the required voltage is different, so that accurate inverter power control may not be achieved.

Accordingly, the present specification proposes a method of adjusting the inverter output voltage vector.

On the other hand, since FIG. 4 does not perform power control, distortion of the inverter power occurs.

However, according to an embodiment of the present invention, the power controller 525 generates a second voltage command value V ^* ₂ for compensating the first voltage command value V ^* ₁ , corresponding to the dc terminal voltage. . Accordingly, the power controller 525 enables the control of the inverter output voltage.

)와의 내적 값이 같은 인버터 출력 전압 벡터들(

,

,)

은, 점선으로 표시된 전동기 전류 벡터(

)와 직교하는 선 위에 위치한다. 이는 주어진 전동기 전류 벡터에서 원하는 인버터 출력 전력을 얻기 위해 출력해야 할 인버터 출력 전압 지령 벡터의 해는 여러 가지가 존재할 수 있음을 의미한다. 인버터 출력 전압 벡터는 인버터 출력 전력 뿐만 아니라 전동기 전류 벡터의 변화에도 영향을 줄 수 있으므로 적절한 전압 벡터의 선정을 통해 전압 지령 생성부(540)에 주는 영향을 최소화하는 것이 필요하다. Referring now to FIG. 7A, a given motor current vector (

Inverter output voltage vectors with the same dot product as

,

,)

Is the motor current vector (indicated by the dotted line)

Located on a line perpendicular to). This means that there can be several solutions to the inverter output voltage command vector that need to be output to obtain the desired inverter output power from a given motor current vector. Since the inverter output voltage vector may affect not only the inverter output power but also the change of the motor current vector, it is necessary to minimize the influence on the voltage command generator 540 by selecting an appropriate voltage vector.

As illustrated in FIG. 7B, the power controller 525 selects any one of the various inverter output voltage command vectors in consideration of the inverter output power command value.

)에 평행하며, 최종 전압 지령치의 벡터(

)가, 제1 전압 지령치의 벡터(

)에 가장 가까운 벡터가 되도록 하는, 벡터를, 보상 전압 지령치의 벡터(

)로 산출한다. 그리고, 전력 제어기(525)는, 산출된 보상 전압 지령치의 벡터(

)를 제2 전압 지령치의 벡터(

)로 출력한다.That is, the power controller 525 is a vector of the output current flowing to the motor as shown in FIG.

Parallel to) and the vector of final voltage setpoint (

) Is the vector of the first voltage setpoint (

Let the vector be the closest vector to the vector of the compensation voltage setpoint (

Calculate The power controller 525 then calculates a vector of the calculated compensation voltage command value (

) Is the vector of the second voltage setpoint (

)

)의 크기는, 하기의 수학식 3에 의해 연산될 수 있다. Vector of compensation voltage setpoint

) May be calculated by Equation 3 below.

,

는, 전동기에 흐르는 출력 전류의 벡터, 출력 전압의 벡터를 나타내며, P^*inv는 전력지령 생성부(520)에서 출력되는 전력 지령치(P^*)를 나타내며, P^*x는 제1 전압 지령치(V^*1)에 기초한 출력 전력 지령치를 나타낸다.here,

,

Denotes a vector of output current flowing through the motor and a vector of output voltage, P ^* inv denotes a power command value P ^* output from the power command generator 520, and P ^* x denotes a first voltage command value V. ^* Indicates the output power command value based on 1).

The inverter output voltage vector, that is, the third voltage command value V ^* 3, may eventually be calculated by the sum of the first voltage command value V ^* ₁ and the second voltage command value V ^* ₂ .

Meanwhile, in FIG. 7B, the size of the third voltage command value V ^* 3 is smaller than the size of the first voltage command value V ^* _{1 due} to the dc short voltage compensated second voltage command value V ^* ₂ . It can be seen that.

)와 전동기에 흐르는 출력 전류의 벡터(

)를 승산하고, 유닛(715)은, 유닛(710)의 출력 값에, 1.5를 승산한다. 이에 따라, 유닛(715)의 출력값은, 제1 전압 지령치(V^*1)에 기초한 출력 전력 지령치( P^*x)일 수 있다.Referring to FIG. 8, the power controller 525 operates according to Equation 3 below. That is, the unit 710 is a vector of the first voltage command value (

) And vector of output current flowing to the motor (

), And the unit 715 multiplies the output value of the unit 710 by 1.5. Accordingly, the output value of the unit 715 may be an output power command value P ^* x based on the first voltage command value V ^* 1.

The unit 720 subtracts the output power command value P ^* x from the power command value P ^* from the power command generation unit 520.

)의 절대값을 출력하며, 유닛(730)은, 유닛(725)의 출력 값에, 1.5를 승산한다. 그리고, 유닛(735)은, 유닛(720)의 출력 값을 유닛(730)의 출력 값으로 제산한다. On the other hand, the unit 725 is a vector of output currents flowing to the motor (

), And the unit 730 multiplies the output value of the unit 725 by 1.5. The unit 735 divides the output value of the unit 720 by the output value of the unit 730.

)와 유닛(725)의 출력 값을 연산하고, 유닛(745)은, 유닛(740)의 출력값과 유닛(735)의 출력값을 승산하여, 최종적으로, 보상 전압 지령치의 벡터(

)의 크기를 출력한다.On the other hand, the unit 740 is a vector of the output current flowing to the motor (

) And the output value of the unit 725, the unit 745 multiplies the output value of the unit 740 and the output value of the unit 735, and finally, the vector of the compensation voltage command value (

) Prints the size of.

Accordingly, according to the embodiment of the present invention, for controlling the desired inverter output power, real-time compensation of the output voltage is possible, and the inverter output power command value can be effectively estimated.

In addition, the proposed control method is capable of controlling the inverter output power without considering the manifold of the motor and thus is not affected by the manifold error of the motor. In addition, since there is no time delay element inside the controller, there is an advantage of reducing the error of the inverter output power due to the delay.

Meanwhile, according to the embodiment of the present invention, even in the field weakening control, the field weakening current command value is input to the voltage command generating unit 530. After that, the inverter output voltage command value V is considered in consideration of the dc terminal voltage component. ^* 3) is generated, so that the field weakening control can be stably performed.

9A to 9B are diagrams for explaining an overmodulation scheme according to the operation of the controller of FIG. 5.

9A illustrates the minimum distance overmodulation technique described in FIG. 4C.

As described in FIG. 5, the adder 555 adds and outputs the first voltage command value V ^* ₁ and the second voltage command value V ^* ₂ . That is, the inverter output voltage command value V ^* ₃ is output as the third voltage command value. The inverter output voltage command value V ^* ₃ is represented by V ^* _a in FIG. 9A.

)가, 인버터(120)에서 출력 가능한 전압을 초과하는 경우, 즉, 공간 벡터 영역(310)의 출력 전압 제한 경계 영역을 초과하는 경우, 최소 거리 기법에 따라, 공간 벡터 영역(310)과의 최소 거리 지점(P3)을 유효 벡터(V_new)로서 설정하는 기법을 예시한다.At this time, the vector of the inverter output voltage command value (

) Exceeds the voltage that can be output from the inverter 120, i.e., exceeds the output voltage limiting boundary region of the space vector region 310, according to the minimum distance technique, A technique of setting the distance point P3 as the valid vector V_new is illustrated.

)에서, 공간 벡터 영역(310)으로의 수직선(Lv1)에 의한, 최소 거리 지점(P3)에 대응하는, 유효 전압 지령치의 벡터(

)가 산출되는 것을 예시한다.In FIG. 9A, the vector of the inverter output voltage command value (

), A vector of the effective voltage command values corresponding to the minimum distance point P3 by the vertical line Lv1 to the space vector region 310

) Is calculated.

However, as shown in FIG. 9A, since the effective vector is calculated by considering only the output voltage among the factors related to the output power of the inverter 120, the output power of the inverter 120 is not precisely controlled and the output output from the inverter 120 is performed. Power may be distorted.

9B illustrates an overmodulation technique in accordance with an embodiment of the present invention.

As described in FIG. 5, the adder 555 adds and outputs the first voltage command value V ^* ₁ and the second voltage command value V ^* ₂ . That is, the inverter output voltage command value V ^* 3 is output as the third voltage command value. The inverter output voltage command value V ^* ₃ is represented by V ^* _a in FIG. 9B.

)가, 인버터(120)에서 출력 가능한 전압을 초과하는 경우, 스위칭 제어 신호 출력부(560)는, 과변조 제어를 수행한다.At this time, the vector of the inverter output voltage command value (

) Exceeds the voltage that can be output from the inverter 120, the switching control signal output unit 560 performs overmodulation control.

)와 인버터 출력 전압 지령치 벡터(

)에 기초한 인버터의 출력 전력과 동일하도록, 최종 출력 전압 지령치 벡터(

)를 산출한다. 그리고, 산출된 최종 출력 전압 지령치 벡터(

)에 기초하여, 스위칭 제어 신호(Sic)를 출력한다.That is, the switching control signal output unit 560 is a vector of the output current flowing to the motor (

) And inverter output voltage setpoint vector (

To the output power of the inverter based on

) Is calculated. Then, the calculated final output voltage command value vector (

), The switching control signal Sic is outputted.

)가, 출력 전압 제한 경계 영역에 위치하며, 인버터 출력 전압 지령치 벡터(

)에 가장 가까운, 벡터인 것을 예시한다. 9B shows the calculated final output voltage command value vector (

) Is located at the output voltage limit boundary region and the inverter output voltage setpoint vector (

Exemplifies a vector closest to

)를 고려한다. That is, when the switching control signal output unit 560 exceeds the output voltage limit boundary region of the space vector region 310, the switching control signal output unit 560 is a vector of the output current flowing to the motor (

Consider.

)에서, 전동기에 흐르는 출력 전류의 벡터(

)에 수직하는 법선(Lv2)을 그린 후, 법선(Lv2)과 출력 전압 제한 경계 영역과 교차하는 지점(Pa)을, 이용하여, 산출되는 최종 출력 전압 지령치 벡터(

)를, 유효 벡터(V_new)로서 설정하는 기법을 예시한다.Specifically, Fig. 9B is a vector of the inverter output voltage command value (

), The vector of output current flowing to the motor (

After drawing a normal line Lv2 perpendicular to), the final output voltage command value vector (calculated) is calculated using the point Pa that intersects the normal line Lv2 and the output voltage limiting boundary region.

) Is illustrated as a valid vector (V_new).

)를 보정하기 위한, 보정 전압 지령치 벡터(

)를 연산하고, 인버터 출력 전압 지령치 벡터(

)와 보정 전압 지령치 벡터(

)를 이용하여, 최종 출력 전압 지령치 벡터(

)를 유효 전압 지령치 벡터로 산출하는 것을 예시한다.9B is an inverter output voltage command value vector (

) To correct the correction voltage setpoint vector (

) And the inverter output voltage setpoint vector (

) And the compensation voltage setpoint vector (

), The final output voltage setpoint vector (

) Is calculated as an effective voltage command value vector.

In this case, unlike FIG. 9A, since the correction voltage command value is generated in consideration of the output current component flowing to the motor, output power control for the inverter 120 is performed. Accordingly, the output power output from the inverter 120 is not distorted.

)의 크기는, 과변조가 아닌, 인버터가 출력 가능한 전압이므로, 안정적인 인버터 제어가 가능하게 된다.In addition, since the correction voltage command value is generated in consideration of the output current component flowing in the motor, the final output voltage command value vector (

) Is a voltage that the inverter can output, not overmodulation, so that stable inverter control is possible.

In addition, distortion of the input current input to the inverter 120, that is, power factor deterioration can be minimized.

On the other hand, the above-mentioned allowable voltage value may be changed according to the variation of the dc terminal voltage. Nevertheless, as described above, the inverter can be stably controlled by calculating the effective voltage command value within the allowable voltage value.

On the other hand, the electric motor drive device 100 according to the embodiment of the present invention, various examples are possible, the application range, laundry appliances, vacuum cleaners, household appliances such as air conditioners and refrigerators, electric cars, hybrid cars, elevators It can be applied to need various power sources.

The motor driving apparatus according to the embodiment of the present invention is not limited to the configuration and method of the embodiments described as described above, the embodiments are all or part of each embodiment so that various modifications can be made May be optionally combined.

On the other hand, the motor drive device of the present invention can be implemented as a code that can be read by the processor on a processor-readable recording medium provided in the motor drive device. The processor-readable recording medium includes all kinds of recording devices that store data that can be read by the processor.

In addition, while the preferred embodiments of the present invention have been shown and described above, the present invention is not limited to the specific embodiments described above, but the technical field to which the invention belongs without departing from the spirit of the invention claimed in the claims. Of course, various modifications can be made by those skilled in the art, and these modifications should not be individually understood from the technical spirit or the prospect of the present invention.

Claims (16)

A rectifier for rectifying the input AC voltage;
An inverter configured to drive the electric motor by converting the rectified voltage;
A capacitor disposed between the rectifier and the inverter; And
A control unit for controlling the operation of the switching element in the inverter;
The control unit,
A torque command generation unit that generates a torque command value based on the speed command value;
A current command generation unit that generates a current command value based on the torque command value;
A voltage command generator that generates a first voltage command value based on the current command value;
Based on the torque command value, an input power instant value based on the input AC voltage is calculated, and the inverter output power command value is based on an dc terminal power instant value that is both ends of the capacitor and an input power instant value based on the input AC voltage. A power command generation unit that calculates a;
A power controller that generates a compensation voltage command value based on the inverter output power command value;
When the vector of the inverter output voltage command value obtained by adding the first voltage command value and the compensation voltage command value exceeds the output voltage limit boundary area output from the inverter, the vector is located at the output voltage limit boundary area and the inverter output voltage command value. And a switching control signal output unit configured to calculate a final output voltage command value vector from a vector closest to a vector of the output voltage, and output a switching control signal for driving the inverter based on the calculated final output voltage command value vector. Electric motor drive. delete The method of claim 1,
The final output voltage command value vector,
And a point perpendicular to a vector perpendicular to a vector of an output current flowing in the motor and a point crossing the output voltage limiting boundary region. The method of claim 1,
The voltage limit value according to the output voltage limit boundary region is varied according to the voltage across the capacitor. delete delete delete The method of claim 1,
The power controller,
The compensation voltage is a vector parallel to a vector of the output current flowing in the motor, and the final output voltage command value vector is the vector closest to the vector of the first voltage command value in consideration of the inverter output power command value. An electric motor drive apparatus characterized by calculating by a vector of a command value. The method of claim 1,
The power command generation unit,
Includes; phase detection unit for detecting the phase of the input AC voltage,
Calculating the input power instant value based on the detected phase of the input AC voltage, the torque command value, and the speed of the calculated motor;
Calculating the dc terminal power instant value based on a direct current power source stored across the capacitor,
And the inverter output power command value is calculated based on the dc terminal power instantaneous value which is both ends of the capacitor and the input power instantaneous value based on the input AC voltage. The method of claim 9,
The power command generation unit,
And an adder configured to add the dc stage power instantaneous values which are both ends of the capacitor, and the input power instantaneous value based on the input AC voltage. The method of claim 1,
An input voltage detector detecting the input AC voltage; And
And a dc end voltage detector detecting a dc end voltage at both ends of the capacitor. The method of claim 1,
And the voltage command generation unit generates the first voltage command value based on the current command value and the output current flowing to the motor. The method of claim 12,
The torque command generation unit,
And the torque command value is calculated at each zero crossing time point of the input voltage. The method of claim 12,
The torque command value is
And a constant time interval between a first zero crossing time point of the input voltage waveform and a second continuous zero crossing time point. The method of claim 12,
The control unit,
And a position estimator configured to estimate a rotor position of the motor based on an output current flowing through the motor, and calculate a speed of the motor based on the estimated rotor position.
And the torque command generation unit generates the torque command value based on the speed command value and the calculated speed. The method of claim 12,
The current command generation unit,
In the field weakening control, the motor drive device, characterized in that for generating the weak field current command value for controlling the field weakening.

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