JP6726839B2 - Inverter control device - Google Patents

Description

The present invention relates to an inverter control device that drives a motor at any rotation speed.

Conventionally, as an inverter control device that drives a brushless DC motor in an overmodulation region, a voltage filter that smoothes a change in the time-axis direction of an AC voltage command obtained by calculation is used to rapidly change the amplitude and phase of the AC voltage command. There is a technique of removing harmonic components of the motor current by avoiding the change and further using a current filter for smoothing the change of the motor current in the time axis direction by the current detector (for example, refer to Patent Document 1).

According to the method of Patent Document 1, it is possible to prevent a sudden change in the voltage applied to the motor at the time of switching from the sine wave PWM control to the overmodulation control, and to improve the control stability of the motor.

Japanese Patent No. 5133834

However, in the control device having the conventional configuration, since the filtering process is performed for both the voltage and the current, the control responsiveness of the motor with respect to the load change is lowered, and the control performance sufficient when the motor is driven in the overmodulation region. May not be obtained.

An object of the present invention is to solve the above-mentioned conventional problems, and to provide an inverter control device capable of improving control stability of a motor in an overmodulation region while ensuring control response of the motor. To do.

In order to solve the above-mentioned conventional problems, an inverter control device of the present invention detects a multi-phase motor, an inverter that converts DC power into AC power and supplies power to the motor, and a current value that drives the motor. In an inverter control device that controls the current value for driving the motor so that the output torque of the motor becomes a required torque, the current command value given to the motor and the current detection value from the current detection means are used. Current control means for generating a voltage command value using proportional-plus-integral control, and adjusts the motor current phase so that the modulation factor calculated based on the voltage command value and the DC bus voltage of the inverter matches its target value. The voltage is adjusted and a substantially trapezoidal wave voltage is applied to the motor during overmodulation in which the voltage command value exceeds the DC bus voltage.

As a result, it is possible to achieve stable motor drive in the overmodulation region with the minimum required current phase without excessively increasing the effect of field weakening control while ensuring the control response of the motor, thus improving the efficiency of the motor drive system. And the drive range of the motor can be expanded.

The inverter control device of the present invention can improve the control stability of the motor in the overmodulation region while ensuring the control response of the motor.

System configuration diagram of the inverter control device of the present invention The 1st block diagram of the current control part in the inverter control apparatus of this invention. The 2nd block diagram of the current control part in the inverter control apparatus of this invention. The figure showing the example of the cutoff frequency setting of the filter in the inverter control apparatus of this invention. 1st operation characteristic view of the inverter control device of the present invention Second operation characteristic diagram of the inverter control device of the present invention Third operation characteristic diagram of the inverter control device of the present invention Definition diagram of coordinate axes in the inverter control device of the present invention

A first invention includes a multi-phase motor, an inverter that converts DC power into AC power and supplies power to the motor, and current detection means that detects a current value for driving the motor. In an inverter control device that controls a current value for driving a motor so as to obtain a required torque, a voltage command value is generated using proportional-plus-integral control based on a current command value given to the motor and a current detection value from a current detection means. The current phase of the motor is adjusted, and the current phase of the motor is adjusted so that the modulation rate calculated based on the voltage command value and the DC bus voltage of the inverter matches the target value, and the voltage command value exceeds the DC bus voltage. A voltage with a substantially trapezoidal waveform is applied to the motor during overmodulation, and while ensuring the control response of the motor, the motor in the overmodulation region is controlled by the minimum necessary current phase without excessively increasing the field weakening control effect. Since stable driving can be realized, the efficiency of the motor drive system can be improved and the drive range of the motor can be expanded.

In a second aspect of the present invention, in particular, in the inverter control device of the first aspect of the invention, there is provided a filter for smoothing a change in the time axis direction to either the q-axis current detection value or the q-axis voltage command value of the rotating coordinate system during overmodulation. Since the cutoff frequency of the filter is changed according to the number of motor revolutions, the influence of the torque pulsation component that synchronizes with the number of motor revolutions can be eliminated, improving motor control stability in the overmodulation region. can do.

A third aspect of the invention is to set the cutoff frequency of the filter so as to remove a harmonic component including a frequency that is six times the electric frequency of the motor in the inverter control device of the second aspect of the invention. Since the influence of the torque ripple component due to the characteristic distortion can be appropriately eliminated, the control stability of the motor in the overmodulation region can be improved even in the motor having the magnetic characteristic distortion.

Embodiments of the present invention will be described below with reference to the drawings. The present invention is not limited to this embodiment.

(Embodiment 1)
FIG. 1 is a system configuration diagram of an inverter control device of the present invention. This inverter control device is supplied with electric power from an AC power supply 1 such as a commercial power supply, rectifies means 2 for rectifying the supplied power, smoothing means 3 for smoothing an output voltage from the rectifying means 2, and smoothing means 3. The orthogonal transformation means 4 for transforming the smoothed voltage into an AC voltage having a desired frequency and voltage value, and the inverter control means 6 for transmitting information for driving the motor 5 to the orthogonal transformation means 4.

The inverter control means 6 can be configured by a microcomputer, a system LSI, or the like, and includes a base driver 10, a PWM signal generation unit 11, a current control unit 12, a current command generation unit 13, a rotor position/speed estimation unit 14, and a current phase. Each functional block of the adjusting unit 15, the phase current converting unit 16, and the modulation factor calculating unit 17 is provided.

The phase current converter 16 observes the DC-side bus current of the orthogonal converter 4 flowing through the current detector 7 and converts the bus current into the phase current of the motor 5 (a specific method of the phase current converter 16). For details, refer to documents such as JP-A-2003-189670).

Needless to say, the phase current of the motor 5 may be directly detected using a current sensor or the like.

The rotor position/velocity estimation unit 14 uses the phase current of the motor 5 converted by the phase current conversion unit 16 and the voltage command value of the motor 5 calculated by the current control unit 12 to determine the rotor magnetic pole position of the motor 5. And the rotation speed is estimated (for the specific method of the rotor position/speed estimation unit 14, refer to, for example, Japanese Patent Laid-Open No. 2001-37281).

The modulation factor calculation unit 17 calculates the modulation factor σ based on the information on the DC bus voltage Vdc detected by the voltage detection unit 8 and the voltage command value of the motor 5 calculated by the current control unit 12.

σ=phase voltage command value amplitude/(Vdc/2) (1)
FIG. 5 shows a waveform example of one phase of the phase voltage command value (Vus, Vvs, Vws) output from the current control unit 12, and the amplitude A1 of the phase voltage command value Vs1 is less than (Vdc/2). Therefore, normal modulation (modulation rate σ≦1) is performed, and the amplitude A2 of the phase voltage command value Vs2 exceeds (Vdc/2), resulting in overmodulation (modulation rate σ>1). Further, since the voltage range that can be output by the orthogonal transformation means 4 is (-Vdc/2 to +Vdc/2), the phase voltage command value itself (sine wave) cannot be applied to the motor 5 during overmodulation. The voltage actually applied to the motor 5 is a substantially trapezoidal wave limited by ±Vdc/2 (in the case of the phase voltage command value Vs2, the waveform shown by the thick line in FIG. 5). Note that during overmodulation (in the case of the phase voltage command value Vs2), the voltage that can be applied to the motor 5 (corresponding to the shaded area in FIG. 5) may be increased compared to during normal modulation (in the case of the phase voltage command value Vs1). Therefore, the drive range of the motor 5 can be expanded.

The current phase adjustment unit 15 adjusts the current phase βm so that the modulation rate σ calculated by the modulation rate calculation unit 17 matches the preset target modulation rate σtg.

Specifically, when the modulation rate σ is smaller than the target modulation rate σtg, the current phase βm is in an excessive state. Therefore, by reducing the current phase βm at a predetermined time change rate Δβdn, the effect of the field weakening control is obtained. To increase the modulation rate σ. When the modulation rate σ is larger than the target modulation rate σtg, the current phase βm is insufficient. Therefore, the current phase βm is changed at a predetermined time change rate Δβup. By increasing it, the effect of field weakening control is enhanced, and the modulation factor σ is made smaller.

In the inverter control device of the present invention, the drive of the motor 5 in the overmodulation region is realized by making the target modulation rate σtg larger than 1.

In the current command generation unit 13, the rotation speed of the motor 5 matches the target speed based on the deviation information between the estimated speed value of the motor 5 estimated by the rotor position speed estimation unit 14 and the target speed given from the outside. The current command amplitude Is is derived by using a proportional-plus-integral calculation or the like. A current command value (Ids, Iqs) of the rotating coordinate system is derived from the current command amplitude Is and the current phase βm output from the current phase adjusting unit 15.

Ids=−Is×sin(βm) (2)
Iqs=Is×cos(βm) (3)
The definition of coordinate axes is shown in FIG. θ is a magnetic pole position estimated value of the motor 5 estimated by the rotor position/velocity estimation unit 14, and βm is a current phase. The three-phase (u, v, w)/2-phase (d, q) conversion and the two-phase (d, q)/3-phase (u, v, w) conversion are known and will not be described below.

FIG. 2 is a configuration diagram of the current control unit 12 according to the first embodiment. The current control unit 12 includes functional blocks of a voltage command calculation unit 12a, a 2-phase/3-phase conversion unit 12b, a 3-phase/2-phase conversion unit 12c, and a current filter processing unit 12d.

The current control unit 12 is configured to use proportional-plus-integral control so as to be applied to applications in which control response of the motor 5 is required, and the phase currents (Iu, Iv, Iw) of the motor 5 converted by the phase current conversion unit 16 are used. ) Is converted to a current detection value (Id, Iq) in the rotating coordinate system by the three-phase/two-phase conversion unit 12c, and the converted q-axis current detection value (Iq) is used by the current filter processing unit 12d using a low-pass filter. The q-axis current filter value (Iqf) is derived.

FIG. 4 is a diagram showing an example of the cutoff frequency setting of the low-pass filter in the current filter processing unit 12d, and the cutoff frequency is set between the fundamental wave and the sixth frequency of the electric frequency of the motor 5.

In FIG. 4, the cutoff frequency is set by a linear function with respect to the electric frequency of the motor 5, but the cutoff frequency is not particularly limited, and after considering the control response of the motor 5 in advance, a function such as a polynomial function is used. The cutoff frequency may be set with.

In the inverter control device of the present invention, the current filter processing unit 12d always uses a low-pass filter. However, the modulation rate σ calculated by the modulation rate calculation unit 17, the target speed given from the outside, and the rotor position speed estimation. The low-pass filter of the current filtering unit 12d may be passed only when at least one of the estimated speed values of the motor 5 estimated by the unit 14 is equal to or greater than a predetermined value (if the speed is less than the predetermined value, the low-pass filter is set to the low-pass filter). The q-axis current detection value (Iq) that is input without passing it is directly output as the q-axis current filter value (Iqf). In this case, the calculation amount of the microcomputer, system LSI, etc. can be minimized.

In the voltage command calculation unit 12a, the current command values (Ids, Iqs) of the rotating coordinate system derived from the current command generation unit 13, the d-axis current detection value (Id) and the q-axis current filter obtained as described above. The voltage command value (Vds, Vqs) of the rotating coordinate system is derived by using proportional-plus-integral control so that the value (Iqf) matches. The voltage command values (Vds, Vqs) in the rotating coordinate system are converted into phase voltage command values (Vus, Vvs, Vws) by the two-phase/three-phase conversion unit 12b.

Specifically, the voltage command value (Vds, Vqs) of the rotating coordinate system in the voltage command calculation unit 12a is derived by the following equation.

Vds=KPD×(Ids−Id)+KID×∫(Ids−Id)dt (4)
Vqs=KPQ×(Iqs−Iqf)+KIQ×∫(Iqs−Iqf)dt (5)
Here, KPD is a d-axis proportional gain, KID is a d-axis integral gain, KPQ is a q-axis proportional gain, and KIQ is a q-axis integral gain.

When the motor 5 is driven in the overmodulation region, it is difficult to apply a voltage according to the command to the motor 5, and the input deviation (d-axis and q-axis current deviation) of the proportional-plus-integral calculation increases or decreases. Therefore, the upper and lower limits are set to the integral term of the proportional integral calculation (the second term of the equations (4) and (5)) so that the calculated value of the proportional integral calculation does not increase or decrease excessively in the overmodulation region. It is configured to be provided.

The PWM signal generator 11 generates a PWM signal for driving the motor 5 from the phase voltage command values (Vus, Vvs, Vws) obtained as described above.

The PWM signal generator 11 generates a PWM signal for driving the motor 5 from the phase voltage command values (Vus, Vvs, Vws) obtained as described above.

The PWM signal obtained as described above is finally output to the base driver 10 to drive the switching element forming the orthogonal transform means 4.

Here, the difference in the operation waveform of the inverter control device depending on the presence or absence of the low-pass filter in the current filter processing unit 12d will be described with reference to FIGS. 6 and 7.

FIG. 6 is an example of operation waveforms when a motor having a magnetic characteristic distortion is driven in an overmodulation region. FIG. 6 shows a case where a low-pass filter is provided and FIG. 7 shows a case where no low-pass filter is provided.

When the low-pass filter is not provided as shown in FIG. 7, the phase voltage command value (the voltage actually applied to the motor 5 is a waveform limited to ±Vdc/2 as shown by the bold line) is used as the magnetic characteristic of the motor. Since a torque ripple component (six times the electrical frequency of the motor 5) due to the distortion is generated, when the motor 5 is driven in the overmodulation region, the duty of the PWM signal generated by the PWM signal generation unit 11 becomes asymmetrical, The control stability of the motor 5 may decrease.

Therefore, when there is a low-pass filter as shown in FIG. 6, the phase voltage command value (actually, the actual value) is removed by removing the torque ripple component (six times the electrical frequency of the motor 5) due to the distortion of the magnetic characteristics of the motor. The voltage applied to the motor 5 is shaped into a substantially trapezoidal wave with a waveform limited by ±Vdc/2 as shown by the thick line, and the duty of the PWM signal generated by the PWM signal generation unit 11 is made symmetrical. By doing so, it is possible to improve control stability when driving a motor having magnetic characteristic distortion in the overmodulation region (in the case of a motor having no magnetic characteristic distortion, the operation of FIG. Almost equal to the waveform).

In this way, the current phase βm of the motor 5 is adjusted so that the modulation rate σ calculated based on the voltage command value of the motor 5 and the DC bus voltage Vdc matches the target modulation rate σtg, and the target modulation rate σtg is set to 1 By making it larger, a voltage having a substantially trapezoidal waveform is applied to the motor 5 when the voltage command value of the motor 5 exceeds the DC bus voltage Vdc, and the voltage command value of the motor 5 is proportionally integrated. By generating the motor 5 while ensuring the control response of the motor 5, it is possible to realize the stable drive of the motor 5 in the overmodulation region by the necessary minimum current phase βm without excessively increasing the effect of the field weakening control. The efficiency of the system can be improved and the drive range of the motor 5 can be expanded.

Further, the q-axis current detection value (Iq) at the time of overmodulation is provided with a filter for smoothing the change in the time axis direction, and the cutoff frequency of the filter is changed according to the rotation speed of the motor 5, thereby rotating the motor 5. Since the influence of the torque pulsation component synchronized with the number can be eliminated, the control stability of the motor 5 in the overmodulation region can be improved.

Further, by setting the cutoff frequency of the filter so as to remove the harmonic component including the frequency 6 times the electric frequency of the motor 5, the influence of the torque ripple component due to the distortion of the magnetic characteristic of the motor can be appropriately eliminated. Therefore, it is possible to improve the control stability of the motor in the overmodulation region even in the case where the magnetic characteristic is distorted.

(Embodiment 2)
Hereinafter, the inverter control device according to the second embodiment of the present invention will be described. The main system configuration is the same as that of the inverter control device in the first embodiment, and the description thereof will be omitted to avoid duplication. Here, the content of the current control unit 12 having a different configuration will be described.

FIG. 3 is a configuration diagram of the current control unit 12 according to the second embodiment. The current control unit 12 includes functional blocks of a voltage command calculation unit 12a, a 2-phase/3-phase conversion unit 12b, a 3-phase/2-phase conversion unit 12c, and a voltage filter processing unit 12e.

The current control unit 12 is configured to use proportional-plus-integral control so as to be applied to applications in which control response of the motor 5 is required, and the phase currents (Iu, Iv, Iw) of the motor 5 converted by the phase current conversion unit 16 are used. ) Is converted into a current detection value (Id, Iq) in the rotating coordinate system by the three-phase/two-phase converting unit 12c, and the current command value (Ids, Iqs) in the rotating coordinate system derived from the current command generating unit 13, The voltage command value (Vds, Vqs) of the rotating coordinate system is derived by using proportional-plus-integral control so that the detected current value (Id, Iq) of the rotating coordinate system obtained as described above matches.

Specifically, the voltage command value (Vds, Vqs) of the rotating coordinate system in the voltage command calculation unit 12a is derived by the following equation.

Vds=KPD×(Ids−Id)+KID×∫(Ids−Id)dt (6)
Vqs=KPQ×(Iqs−Iq)+KIQ×∫(Iqs−Iq)dt (7)
Here, KPD is a d-axis proportional gain, KID is a d-axis integral gain, KPQ is a q-axis proportional gain, and KIQ is a q-axis integral gain.

When the motor 5 is driven in the overmodulation region, it is difficult to apply a voltage according to the command to the motor 5, and the input deviation (d-axis and q-axis current deviation) of the proportional-plus-integral calculation increases or decreases. Therefore, upper and lower limits are set on the integral terms of the proportional-plus-integral calculation (the second term of the formulas (6) and (7)) so that the calculated value of the proportional-plus-integral calculation does not increase or decrease excessively in the overmodulation region. It is configured to be provided.

The q-axis voltage command value (Vqs) derived by the voltage command calculation unit 12a is used by the voltage filter processing unit 12e to derive the q-axis voltage filter value (Vqf) using a low-pass filter.

Here, the cutoff frequency setting of the low-pass filter in the voltage filter processing unit 12e is similar to that of the low-pass filter of the current filter processing unit 12d in the first embodiment, as shown in FIG. Set the cutoff frequency between the wave and the 6th frequency.

Further, similarly to the low-pass filter of the current filter processing unit 12d in the first embodiment, in FIG. 4, the cutoff frequency is set by a linear function with respect to the electric frequency of the motor 5, but the cutoff frequency is not particularly limited. Alternatively, the cutoff frequency may be set by a function made of a polynomial or the like after considering the control response of the motor 5 in advance.

In the inverter control device of the present invention, the voltage filter processing unit 12e always uses the low-pass filter. However, the modulation rate σ calculated by the modulation rate calculation unit 17, the target speed given from the outside, and the rotor position speed estimation. The low-pass filter of the voltage filter processing unit 12e may be passed only when at least one of the estimated speed values of the motor 5 estimated by the unit 14 is equal to or higher than a predetermined value (if the speed is less than the predetermined value, the low-pass filter is used). The q-axis voltage command value (Vqs) that is input without passing it is directly output as the q-axis voltage filter value (Vqf). In this case, the calculation amount of the microcomputer, system LSI, etc. can be minimized.

The d-axis voltage command value (Vds) and the q-axis voltage filter value (Vqf) obtained as described above are converted into phase voltage command values (Vus, Vvs, Vws) by the 2-phase/3-phase conversion unit 12b.

Here, the difference between the operation waveforms of the inverter control device depending on the presence or absence of the low-pass filter in the voltage filter processing unit 12e will be described with reference to FIG. 6 and FIG. 7 (the operation waveforms of the current filter processing unit 12d in the first embodiment will be described. The waveform will be similar to the low-pass filter).

FIG. 6 is an example of operation waveforms when a motor having magnetic characteristic distortion is driven in an overmodulation region. FIG. 6 shows a case with a low-pass filter, and FIG. 7 shows a case without a low-pass filter.

When the low-pass filter is not provided as shown in FIG. 7, the phase voltage command value (the voltage actually applied to the motor 5 is a waveform limited to ±Vdc/2 as shown by the bold line) is used as the magnetic characteristic of the motor. Since a torque ripple component (six times the electrical frequency of the motor 5) due to the distortion is generated, when the motor 5 is driven in the overmodulation region, the duty of the PWM signal generated by the PWM signal generation unit 11 becomes asymmetrical, The control stability of the motor 5 may decrease.

Therefore, when there is a low-pass filter as shown in FIG. 6, the phase voltage command value (actually, the actual value) is removed by removing the torque ripple component (six times the electrical frequency of the motor 5) due to the distortion of the magnetic characteristics of the motor. The voltage applied to the motor 5 is shaped into a substantially trapezoidal wave with a waveform limited by ±Vdc/2 as shown by the thick line, and the duty of the PWM signal generated by the PWM signal generation unit 11 is made symmetrical. By doing so, it is possible to improve control stability when driving a motor having magnetic characteristic distortion in the overmodulation region (in the case of a motor having no magnetic characteristic distortion, the operation of FIG. Almost equal to the waveform).

In this way, the current phase βm of the motor 5 is adjusted so that the modulation rate σ calculated based on the voltage command value of the motor 5 and the DC bus voltage Vdc matches the target modulation rate σtg, and the target modulation rate σtg is set to 1 By making it larger, a voltage having a substantially trapezoidal waveform is applied to the motor 5 when the voltage command value of the motor 5 exceeds the DC bus voltage Vdc, and the voltage command value of the motor 5 is proportionally integrated. By generating the motor 5 while ensuring the control response of the motor 5, it is possible to realize the stable drive of the motor 5 in the overmodulation region by the necessary minimum current phase βm without excessively increasing the effect of the field weakening control. The efficiency of the system can be improved and the drive range of the motor 5 can be expanded.

Further, the q-axis voltage command value (Vq) is provided with a filter that smoothes changes in the time-axis direction during overmodulation, and the cutoff frequency of the filter is changed according to the number of rotations of the motor 5. Since the influence of the torque pulsation component synchronized with the number can be eliminated, the control stability of the motor 5 in the overmodulation region can be improved.

Further, by setting the cutoff frequency of the filter so as to remove the harmonic component including the frequency 6 times the electric frequency of the motor 5, the influence of the torque ripple component due to the distortion of the magnetic characteristic of the motor can be appropriately eliminated. Therefore, it is possible to improve the control stability of the motor in the overmodulation region even in the case where the magnetic characteristic has distortion.

As described above, the inverter control device according to the present invention can improve the control stability of the motor in the overmodulation region while ensuring the control responsiveness of the motor. Therefore, the air conditioner, the refrigerator, and the refrigerator. It can be applied to applications such as heat pump water heaters that drive compressor motors.

DESCRIPTION OF SYMBOLS 1 AC power supply 2 Rectification means 3 Smoothing means 3a Reactor 3b Capacitor 4 Orthogonal conversion means 5 Motor 6 Inverter control means 7 Current detection means 8 Voltage detection means 10 Base driver 11 PWM signal generation section 12 Current control section 12a Voltage command calculation section 12b 2 Phase/3 phase converter 12c 3 phase/2 phase converter 12d Current filter processor 12e Voltage filter processor 13 Current command generator 14 Rotor position/speed estimator 15 Current phase adjuster 16 Phase current converter 17 Modulation factor calculation Part 18 Speed/modulation rate setting part

Claims (2)

A motor having a plurality of phases, an inverter that converts DC power into AC power and supplies power to the motor, and a current detection unit that detects a current value for driving the motor, and the output torque of the motor is a required torque. In the inverter controller that controls the current value for driving the motor so that the voltage command value is generated using proportional-plus-integral control based on the current command value given to the motor and the current detection value from the current detection means. Current control means for adjusting the current phase of the motor so that the modulation rate calculated based on the voltage command value and the DC bus voltage of the inverter matches the target value, and the q-axis of the rotating coordinate system. A filter for smoothing changes in the time axis direction to either the current detection value or the q-axis voltage command value is provided, the cutoff frequency of the filter is changed according to the rotation speed of the motor, and the voltage command value is An inverter control device, wherein a substantially trapezoidal wave voltage is applied to the motor during overmodulation exceeding a DC bus voltage. The inverter control device according to claim 1 , wherein the cutoff frequency of the filter is set so as to remove a harmonic component including a frequency that is six times the electrical frequency of the motor.

Images