JP4804381B2 - Electric motor drive control device and electric motor - Google Patents

Description

  The present invention relates to an electric motor drive control device for driving an electric motor having a stator winding independently wound.

  Conventionally, an electric motor in which the stator winding is an independent winding expands the operating range of the motor or increases the output torque of the applied product by switching the Y winding and Δ connection of the motor winding. Also, it is used for the purpose of increasing the maximum rotational speed or increasing the capacity.

  As this type of control device, there is a control device that performs vector control of an electric motor by switching a Y-connection and a Δ-connection with a mechanical switching device (see, for example, Patent Document 1).

  In addition, there are two inverters that switch Y connection and Δ connection by operation of the inverter (see, for example, Patent Document 2).

  Also, there is a motor that drives a motor by referring to a voltage vector output from two inverters from a table from the primary magnetic flux and output torque of the motor (see, for example, Patent Document 3).

  In addition, the output of two inverters is combined to increase the capacity, and there is a low noise, small size, economical and high efficiency with excellent low speed torque characteristics (see, for example, Patent Document 4).

  In addition, there is a device that obtains a large output by positively using harmonic components that cannot be passed to a Y-connection or Δ-connection motor (for example, see Patent Document 5).

  In addition, there is a motor that has achieved high output of the motor by eliminating the voltage shortage without using a booster circuit (see, for example, Patent Document 6).

  Furthermore, there is also one that drives an electric motor with two inverters coupled to first and second energy sources (see, for example, Patent Document 7).

Japanese Patent Publication No. 7-99958 Japanese Patent Publication No. 7-99959 Japanese Patent Publication No. 6-101958 Japanese Patent No. 3352182 JP 2006-136144 A JP 2006-149145 A JP 2006-238686 A

  In the case of the technique described in Patent Document 1, when switching between Δ connection and Y connection, the motor needs to be stopped, and the motor needs to be a slip induction motor as shown in Patent Document 1. . Furthermore, at the time of connection switching, it is necessary to switch control constants in vector control, such as motor constants and control gains, according to the connection.

  In the case of the technique shown in Patent Document 2, the connection can be switched without a mechanical winding switching mechanism. However, as in Patent Document 1, control constants in vector control, such as motor constants and control gains, can be obtained at the time of connection switching. It is necessary to switch according to the connection.

  The technique shown in Patent Document 3 is a control method that eliminates the need to switch the control constant in vector control according to the connection when switching the connection, but according to the hysteresis comparator output corresponding to the magnitude of the primary magnetic flux and output torque of the motor. The voltage vector obtained from the switching table is output. While there are no control constants, the table is very large and the conditions for table selection are diverse, so the control is complicated.

  Patent Document 4 discloses a technique for increasing the output voltage by outputting two inverters with opposite polarities. Furthermore, the synthesis theory of the output voltage is shown (see page 6, FIG. 7). However, the relationship with the rotor position of the electric motor is not described at all, and the rotor position is obtained from a pulse speedometer (PG).

  Patent Documents 5 and 6 show a technique for increasing the output by superimposing harmonic components on the fundamental wave by two inverters or by driving the two units to drive in opposite phases. Yes. In addition, technologies for increasing the output of conventional motors based on Y-connection and conventional motors based on Δ-connection are shown, but both advantages of Y-connection and Δ-connection are used. There is no indication of control to improve performance by switching as much as possible. Furthermore, the rotor position uses a phase detector, and no position sensorless technique is shown.

  Patent Document 7 shows a conceptual diagram specialized in one phase considering the three-phase symmetry of two inverters (FIG. 2 of Patent Document 7). However, the control has dq axes for two inverters, which is synonymous with driving one electric motor with two controllers, and complicates the control. Furthermore, it is intended for a slippery electric motor such as an AC motor.

  The present invention has been made to solve the above-described problems. The first object is to provide the advantages of both the Y connection and the Δ connection, so that the intermediate state between the Y connection state and the Δ connection state can be obtained. By controlling the motor and overmodulation operation with the control of two inverter units as in Y connection, the motor efficiency is maximized while maintaining the linearity without changing the motor constant according to the connection state. An electric motor drive control device capable of always driving under conditions is obtained.

  The second object is to obtain a motor drive control device applicable to a non-slip synchronous motor, for example, a permanent magnet synchronous motor.

  A third object is to obtain an electric motor drive control device capable of realizing position sensorless driving for driving an electric motor without detecting the rotor position.

An electric motor drive control device according to the present invention includes an AC / DC power converter that converts AC power into DC power, and an AC voltage generated from the DC power of the AC / DC power converter, and outputs the AC voltage to the independent winding motor. And a second inverter section, and a control means for controlling the first and second inverter sections so that the independent winding type motor performs a Y-connection equivalent operation or a Δ-connection equivalent operation. Is shifted from the Y-connection equivalent operation state to the Δ-connection equivalent operation state, the phase of the vector of the magnetic flux generated in the armature winding of the independent winding type motor and the vector of the combined voltage by the first and second inverter units is The output voltages of the first and second inverter units are controlled so as to maintain the parallel relationship.

  In the present invention, it is possible to output a voltage without harmonic distortion up to twice the voltage with respect to the sine wave output in the configuration of one inverter, and thereby twice the case where one terminal is connected without forming an independent winding. The output voltage can be expanded, and high-speed rotation and high output can be realized by the increase of the output voltage.

  In addition, the motor characteristics can be changed smoothly with two inverters for a motor that has high efficiency at low speeds at low speeds and a motor that can rotate at high speeds at high speeds, which can greatly improve the efficiency at low speeds. it can.

  In addition, since the phase of the armature stator coil axis and the combined output voltage axis is controlled to be unique, the motor constant in the controller may be unchanged, and position sensorless control can also be realized without losing linearity, Stable control can be realized. Furthermore, there is an effect that the conventional control block can be easily applied.

  Furthermore, connection switching can be realized smoothly without stopping the electric motor, and the efficiency in the intermediate state of the connection state can also be improved.

FIG. 1 is a circuit diagram showing a configuration of an electric motor drive control device according to an embodiment of the present invention.
The motor drive control device according to the present embodiment is equivalent to converting the AC / DC power converter 2 that converts AC power supplied from the AC power source 1 into DC power and the connection (Y connection, Δ connection) of the independent winding motor 3. The two inverter units 4 and 5 that are driven and controlled, the current detector 6 that detects the current flowing in the coil of the independent winding type motor 3, and the DC shunt resistor inserted in the negative bus of the AC / DC power conversion unit 2 7 a to 7 c and control means (not shown) for controlling the two inverter units 4 and 5. The above-described independent winding type motor 3 is an electric motor in which a stator coil is not connected, and both ends of the stator coil are input ends.

  In the present embodiment, the coils of each phase of the independent winding type motor 3 will be described as a-phase, b-phase, and c-phase coils, and the currents flowing through the coils will be described as ia, ib, and ic. Further, as shown in FIG. 2, the output phase current from the inverter in the case of driving the Δ-connected and Y-connected motors with the conventional single inverter configuration will be described as iu, iv, iw.

FIG. 2A shows Δ connection, and as shown in the figure, the relationship between the output phase currents iu, iv, iw from the inverter and the coil currents ia, ib, ic is output phase current ≠ coil current. . FIG. 2B shows a Y connection. In this case, the relationship between the output phase current from the inverter and the coil current is output phase current = coil current. Therefore, in the case of Δ connection, it is common to drive the motor by setting the motor constant equivalently converted to Y connection. Incidentally, when L _Δ of the _Δ connection is equivalently converted to the Y connection, L _Y = L _Δ / 3 is expressed. Further, the back electromotive force by the magnet is 1 / √3 times.

  In the present embodiment, since the motor 3 has independent windings, the detectable currents are the coil currents ia, ib, and ic, which are considered as application development of the Y-connected motor. When the independent winding type motor 3 is operated in an equivalent manner to the Y connection by the two inverter units 4 and 5, the inverter unit 5 is operated at a modulation factor = 0 (output voltage = 0). In the case of an equivalent operation with Δ connection, the inverter unit 5 may be operated at the same modulation rate as the inverter unit 4 with a delay phase of 120 degrees with respect to the inverter unit 4. This control is described in detail on page 3 of Patent Document 2 cited as the prior art.

  First, a case where the operation is equivalent to Δ connection will be described with reference to FIG. When the inverter unit 4 and the inverter unit 5 shown in FIG. 1 are expressed as voltage sources, they become voltage sources of symmetrical three-phase Y connection as shown in the figure. Here, a1, b1, c1 and a2, b2, c2 of the voltage source have the same reference numerals as in FIG. In the case of Δ connection, as described in FIG. 2A, the relationship between the coil currents ia, ib, ic and the output phase currents iu, iv, iw from the inverter is the coil current ≠ output phase current. In the case of the configuration of two inverters, the coil current is equal to the output phase current as shown in FIG.

  Therefore, the amplitude and phase of the current output from the inverter differ between the case of Δ connection equivalent operation with two inverters and the case of Δ connection operation with one inverter. In the case of two inverters, the inverter For one unit configuration, the amplitude is 1 / √3 and the phase is advanced by 30 degrees.

  Therefore, in the conventional Δ connection equivalent operation, the detection current of the motor control is calculated as iu = ia−ic using the fact that the output phase current iu in the single inverter configuration matches the coil current ia−ic, It is controlled by the motor constant equivalently converted to Y connection.

  However, when Δu connection equivalent operation is performed by calculating iu = ia−ic, it means that the state during the transition from the Y connection state to the Δ connection state cannot be controlled. In the present embodiment, not only the state in the middle of the transition from the Y-connection state to the Δ-connection state, but also the independent winding type electric motor 3 that simulates all operations equivalent to the Y-connection including the Δ-connection equivalent operation is driven. This is an electric motor drive control device having two inverters.

  From the equivalent circuit of FIG. 3 (a), a voltage of the difference between the phase voltage a1 by the inverter unit 4 and the phase voltage a2 by the inverter unit 5 is applied to the a-phase coil. If this a1-a2 is a12, a12 is It can be regarded as a line voltage between a1 and a2. The same applies to the b-phase coil and the c-phase coil, and an equivalent circuit diagram in the case where the voltage sources of the inverter unit 4 and the inverter unit 5 in FIG. 3A are newly replaced with line voltages is shown in FIG. Shown in In this figure, assuming that the terminal side indicated by the black dots among the combined voltages a12, b12, and c12 of the inverter unit 4 and the inverter unit 5 is a virtual GND, it is equivalent to a conventional Y-connection voltage source and a Y-connection motor. Can be replaced.

  This is represented by a voltage vector diagram as shown in FIG. FIG. 4A shows a Y-connection equivalent operation state. The voltage vectors a2, b2, and c2 of the inverter unit 5 have the modulation rate = 0 as described above, and therefore, the voltage vector indicated by the white point is zero and exists at the origin. Since the combined voltage vectors a12, b12, and c12 in the same direction as the vector direction of each phase coil may be output, the voltage vectors a1, b1, and c1 output from the inverter unit 4 are on the vector axis of each phase coil. Exists. Note that the circle shown in FIG. 4A indicates the modulation rate = 1, and the voltage output from the inverter unit 4 indicates the state where the modulation rate = 1. The definition of this modulation factor will be described later.

  FIG. 4B is a voltage vector diagram showing the Δ connection equivalent operation state. The output voltage of the inverter unit 5 is output from the inverter unit 4 with a 120-degree delayed phase, and the combined voltage vectors a12, b12, and c12 of the inverter units 4 and 5 indicated by arrows in the figure are the vectors of the respective phase coils. When oriented in the axial direction, the voltage vectors a1, b1, and c1 of the inverter unit 4 are delayed by 30 degrees for each phase coil, and the voltage vectors a2, b2, and c2 of the inverter unit 5 are 150 degrees for each phase coil. Is output with a delay phase of.

  Although not shown in FIG. 4B, voltage vectors a1 and c2 are on the a-phase coil axis, voltage vectors b1 and a2 are on the b-phase coil axis, and voltage vectors c1 and b2 are on the c-phase coil axis. Then, it is clear that the vertex of the equilateral triangle by the combined voltage vectors a12, b12, and c12 is on the vector axis of each phase coil, and the sides of the triangle and the coil axis are not parallel.

  4 (a) and 4 (b), an operation state during the transition from the Y connection equivalent operation state to the Δ connection equivalent operation state will be considered. 4A is the initial state, and until the Δ connection equivalent operation state of FIG. 4B is reached, the output voltages of the inverter units 4 and 5 draw concentric circles. Furthermore, the combined voltage vectors a12, b12, and c12 of the inverter units 4 and 5 move in parallel with the vector direction of each phase coil.

  From these two matters, the voltage vectors a1, b1, and c1 of the inverter unit 4 move on the circumference of the modulation factor = 1 as shown in FIG. In addition, the voltage vectors a2, b2, and c2 of the inverter unit 5 move along concentric circle points (on the dotted line in the figure) that increase with a delay of 30 degrees from the vector axis of each phase coil. As a result, a voltage vector that is an intermediate state between the Y-connection equivalent operation state and the Δ-connection equivalent operation state is derived as a vector diagram.

  From FIG. 4 (c), the combined voltage vectors a12, b12, and c12 of the inverter units 4 and 5 maintain a predetermined relationship with the phase of the stator coil vector, in other words, the armature magnetic flux vector of the stator. In addition, by controlling the output voltages of the two inverter units 4 and 5, two inverters can be configured without changing the vector control portion of the motor control in the single inverter configuration.

  As described above, even when the two inverters are configured, if the phase relationship between the combined voltage axis of the voltages output from the two inverter units 4 and 5 and the magnetic flux axis of the stator coil is maintained at a predetermined value, the inverter 2 All states of the operation by the stand can be considered as an equivalent circuit with the Y connection, and it is possible to easily drive the motor 3 with independent winding without changing any conventional motor drive control unit.

  In the state transition from FIG. 4 (a) → (c) → (b), the large circle indicated by the dotted line is set to the maximum output, and the modulation rate is set to 1. Therefore, in the case of a single inverter configuration, the modulation is performed. When the ratio exceeds 1, harmonic distortion occurs in the output. However, in the case of two inverters, the voltage can be output without harmonic distortion even when the modulation ratio is increased to √3 (FIG. 4A to FIG. 4). (See (c)).

  Further, since the operation state of FIG. 4C is between the Y-connection equivalent operation state of FIG. 4A and the Δ-connection equivalent operation state of FIG. 4B, the connection switching is smoothly performed. It is possible to switch the connection while continuing the operation without stopping and restarting the independent winding type electric motor 3 once.

  In the present embodiment, the vector axis of each phase coil and the combined voltage vectors a12, b12, and c12 have been described as having the same direction. However, the armature magnetic flux of the stator is originally generated orthogonal to the current flowing through the coil. . Since the electric motor is an inductive load, the power factor does not always be 1. Therefore, even if the vector axis of each phase coil and the combined voltage vectors a12, b12, c12 are in the same direction, the direction of the voltage and current vector is the force. When the ratio ≠ 1, they do not match and the armature flux vector and voltage vector are not always orthogonal. However, since the reactive power increases, the operation is performed with a power factor of approximately 1, so if the combined voltage vectors a12, b12, c12 and the vector axis of each phase coil are controlled in the same direction, the magnetic flux vector and the combined voltage vector. a12, b12, and c12 can be controlled in a substantially orthogonal relationship, and a predetermined phase relationship can be maintained.

  Alternatively, the magnetic flux may be calculated by calculation, and the magnetic flux vector and the combined voltage vectors a12, b12, and c12 may be controlled in an orthogonal relationship. Furthermore, it is only necessary to maintain a predetermined phase relationship even if it is not an orthogonal relationship, and in that case, reactive power increases compared to maintaining the orthogonal relationship. Needless to say, if the magnetic flux can be measured rather than calculated, it has never been exceeded. Furthermore, a detection circuit for detecting the composite voltage vectors a12, b12, and c12 may be added. In this case, errors such as a short-circuit prevention time and semiconductor device on-voltage variations in the inverter units 4 and 5 can be reduced. It is possible to maintain a highly accurate vector phase relationship.

  When the independent winding type motor 3 is a permanent magnet synchronous motor, it is necessary to control the magnetic flux generated from the stator by energizing the stator according to the position of the rotor. Therefore, in the prior art, a rotor position detector is required. Further, in the case of an induction motor with slip, the steep fluctuation at the time of transition from the Y-connection equivalent operation state to the Δ-connection equivalent operation state can absorb the slip, and the operation of the motor can be continued.

  In the present embodiment, the motor control is performed so that the combined voltage of the voltages output from the inverter units 4 and 5 matches the stator coordinate axis, generally, the stationary coordinate axis for control (referred to as “αβ axis”). By doing this, the position sensorless control of the independent winding type motor 3 can be easily changed by changing only the voltage output part of the position sensorless control block of the same rotor as the conventional permanent magnet synchronous motor and one inverter configuration. Expansion to a two-unit configuration can be realized. This makes it possible to easily apply existing position sensorless control that guarantees reliable operation without complicating position sensorless control that requires complex calculations more than necessary. It can be applied to high-efficiency permanent magnet synchronous motors to achieve higher efficiency as a product.

  FIG. 5 shows an example of the position sensorless control unit 13 having a single-phase configuration of an electric motor 11 and an inverter unit 12 that are not three-phase connected, which is not a conventional independent winding type. In this embodiment, only the coordinate conversion unit 15 that converts the two-axis voltages (vd, vq) output from the output voltage calculation unit 14 in the position sensorless control unit 13 to the three-phase voltage (vuvw) is changed. Sensorless control can be realized. In addition, the coordinate conversion part 15 in FIG. 5 is represented by Formula (1) in the case of the structure of one unit of the conventional three-phase connection motor 11 and the inverter part 12. FIG.

  When driving the motor 3 having independent windings by the inverter units 4 and 5, the configuration of two inverters that can support position sensorless control only by changing the coordinate conversion unit 15 in FIG. 5 to the block shown in FIG. A drive control device is obtained.

  Here, the control means of this apparatus will be described with reference to the block diagram shown in FIG. The biaxial voltage (vd, vq) from the output voltage calculation unit 14 shown in FIG. 5 is distributed to the outputs from the two inverter units 4 and 5 to generate AC power to be supplied to the independent winding type motor 3. Is the output voltage generator 21. This output voltage generation unit 21 is a difference between the single inverter configuration and the dual inverter configuration. Next, the modulation rate distributor 22 distributes the biaxial voltages (vd, vq) to the inverter units 4 and 5. If the command modulation rate from the output voltage calculation unit 14 is set to vk, vk is represented by a norm value by vd and vq. Therefore, as shown in FIG. 7, the radius of the circle inscribed in the regular hexagon by the voltage vector is set as the modulation rate. Define vk = 1. The modulation factor distributor 22 calculates the command modulation factor vk from the input biaxial voltages (vd, vq), and the command voltage (vd1, vq1) to the inverter unit 4 and the command voltage (vd2, vq1, vq1) to the inverter unit 5. vq2) is calculated. When the modulation rate of the inverter unit 4 is vk1, and the modulation rate of the inverter unit 5 is vk2, vd1, vq1, vd2, and vq2 are given as shown in the equation (2).

  If the command modulation rate vk is 1 or less, it can be output in the Y-connection equivalent operation state as shown in FIG. In the case of a permanent magnet synchronous motor, the same torque can be output with less current in the Y-connection equivalent operation, so the Y-connection equivalent operation state is the most efficient operation method with the least loss. Therefore, up to vk ≦ 1, the modulation factor distributor 22 operates to drive in the Y-connection equivalent operation state which is the state of FIG.

  Therefore, the modulation rate vk1 for the inverter unit 4 is equal to the command modulation rate vk, and the modulation rate vk2 = 0 for the inverter unit 5. The phase compensated from the phase compensator 23 is θ1 = 0. Further, since the modulation rate vk2 = 0 for the inverter unit 5, the output becomes 0 regardless of the number of θ2. Therefore, the coordinate conversion units 24 a and 24 b for the inverter units 4 and 5 perform rotational coordinate conversion at the phase angle θ input to the output voltage generation unit 21. Similarly to the coordinate conversion unit 15, the coordinate conversion units 24a and 24b also perform coordinate conversion based on Expression (1).

  Next, when vk> 1, since output voltage distortion starts to occur in the Y-connection equivalent operation state, a transition is made from the Y-connection equivalent operation state to the intermediate operation from the Δ-connection equivalent operation state. This is the state shown in FIG. At this time, the modulation rate vk1 of the inverter unit 4 is kept at vk1 = 1, and the output vk2 to the inverter unit 5 is gradually increased from 0, thereby enabling an output with the command modulation rate vk exceeding 1. To do. At this time, vk2 is obtained from the geometrical shape of the triangle by vk and is given as shown in equation (3).

  Therefore, according to the modulation rate vk1 = 1 to the inverter unit 4 and the modulation rate vk2 to the inverter unit 5 according to the equation (3), the dq-axis voltage for the two inverters shown in the equation (2) is output from the modulation rate distributor 22 And input to the respective coordinate conversion units 24a and 24b.

  Now, with respect to the rotation angle, the present embodiment drives the motor by making the phase relationship between the combined voltage axis of the voltages output from the two inverters 4 and 5 and the magnetic flux axis of the stator coil constant. . Therefore, θ2 is 150 degrees delayed by adding a delay angle = 120 degrees due to the Δ connection equivalent operation state and a delay angle = 30 degrees for matching the phase relationship of the vector of the composite voltage axis with the vector of the stator magnetic flux axis. Coordinate conversion.

  Since the delay direction is defined as negative, θ2 = 150 degrees in FIG. 6, and θ1 becomes a constant value regardless of vk until 1 <vk ≦ √3. Therefore, it is added that if θ2 = 150 degrees even in the range up to vk ≦ 1, θ2 can be a constant value regardless of vk even in the range of 0 ≦ vk ≦ √3.

  Furthermore, θ1 becomes θ1 = 0 when vk = 1, in other words, in the Y-connection equivalent operation state (FIG. 4A), and in other words, when vk = √3, in other words, the Δ-connection equivalent operation state (FIG. 4 ( In the case of b)), θ1 = 30 degrees. The inverter unit 4 in the range of 1 <vk ≦ √3 operates on a circular locus of vk1 = 1. Therefore, θ1 can also be obtained geometrically of a triangle and is given by equation (4).

  Therefore, the coordinate conversion unit 24a of the inverter unit 4 performs rotational coordinate conversion at an angle θ−θ1 obtained by subtracting θ1 given by the equation (4) from the phase angle θ input to the output voltage generation unit 21, and the inverter The coordinate conversion unit 24b of the unit 5 performs rotational coordinate conversion at an angle θ−θ2 obtained by subtracting θ2 (= 150 degrees) from the phase angle θ.

  Next, the range of √3 <vk ≦ 2 will be described. In this range, there are two methods that can output the voltage applied to the independent winding type motor 3 without distortion. Therefore, two methods will be described in order.

  First, method 1 will be described. When vk = √3, the operation state is the same as the Δ-connection equivalent operation state. Therefore, when vk ≧ √3, each voltage source (a1 to c1 and a2 to c2) output from the inverter units 4 and 5 applies the third harmonic to the voltage, and the line voltage (a12 , B12, c12) no third harmonic appears.

  If this is a Δ connection, as can be seen from FIG. 2A, the neutral point of the three-phase coil is not connected to the inverter units 4 and 5 and is floating with respect to the inverter units 4 and 5. It has become. Therefore, the third harmonic can be superimposed. However, since the states of FIGS. 4A and 4C are not the Δ connection equivalent operation state, when the third harmonic is superimposed, the third harmonic is generated in the current flowing through the coil. Therefore, it is desirable not to superimpose the third harmonic.

  Therefore, the effective value of the output voltage can be increased by using a conventional method called third harmonic superposition. FIG. 8 shows the waveform of the third harmonic superimposed and the waveform of the fundamental wave. When the third harmonic is superimposed on the amplitude of the fundamental wave, the peak can be suppressed to 0.866 (= √3 / 2), the output voltage can be increased, and even if the modulation factor is 1 or more, output is possible without distortion of the voltage. .

  Therefore, it is possible to output up to the circumscribed circle arc circumscribing the apex of the regular hexagon in FIG. 7, and the maximum output circle becomes larger as shown in the vector diagram of FIG. In this embodiment, the radius of the inner circle in FIG. 4D is set to 1, and in this case, the radius of the outer circle is 2 / √3 (= 1.155). Therefore, the combined voltage vectors a12, b12, and c12 in the Δ connection equivalent operation state in FIG. 4D are equivalent to √3 when vk = √3 and 2 when vk = 2 and have linearity. Recognize.

  From the vector diagram of FIG. 4D, combined voltage vectors a12, b12, and c12 are generated so as to be parallel to the stator coil axes a, b, and c, and the combined voltage vector a12 is maintained while maintaining the Δ connection equivalent operation state. , B12, and c12 are expanded, the modulation rate of the inverters 4 and 5 is set to vk1 = vk2 = vk / √3, and the compensation phase θ1 = 30 degrees and θ2 = 150 degrees from the phase compensator 23 are kept constant. Thus, the independent winding type electric motor 3 can be driven sensorlessly. By using third-order harmonic superposition, even in the range of √3 <vk ≦ 2, it can operate without distorting the output voltage in Δ connection equivalent operation state, and the current can be reduced by increasing the output voltage. And can be driven with high efficiency.

  Next, method 2 will be described. Method 2 is a method of increasing the output voltage by controlling the voltage phase relationship between the inverter units 4 and 5, and will be described with reference to the vector diagram of FIG.

  When the command modulation rate vk = √3, the state is as shown in FIG. 4B. From this state, the output phases of the inverters 4 and 5 are changed so that the state shown in FIG. In FIG. 4E, the compensation phase θ1 of the inverter unit 4 is shifted in a direction that becomes 0 degrees, and the compensation phase θ2 of the inverter unit 5 is shifted in a direction that becomes a delayed phase of 180 degrees. Through the state shown in FIG. 4E, the locus of the output voltage of the inverters 4 and 5 is drawn until the state shown in FIG. Even when the modulation factor is increased to 2, a voltage can be output without harmonic distortion.

  The modulation factor vk1 of the inverter unit 4 continues to hold vk1 = 1 as in the case of 1 <vk ≦ √3. Since the modulation factor vk2 of the inverter unit 5 reaches vk2 = 1 when vk = √3, vk2 also keeps the modulation factor = 1 (vk2 = 1) in the range of √3 <vk ≦ 2. Therefore, the modulation factor distributor 22 holds the modulation factor Vk1 = 1 for the inverter unit 4 and the modulation factor vk2 = 1 for the inverter unit 5, and outputs a dq-axis voltage.

  Furthermore, the operation of the phase compensator 23 in Method 2 will be described. It can be seen from FIG. 4B that when vk = √3, θ1 = 30 degrees and θ2 = 150 degrees. 4 (e) and 4 (f), the transition from this state to the state from FIG. 4 (e) to FIG. 4 (f) indicates that θ1 is directed in the direction of 0 ° and θ2 is directed in the direction of 180 °. Recognize. Accordingly, θ1 and θ2 geometrically of the triangle are given as equations (5) and (6) using vk.

  In the method 2 of the present embodiment, the Y-connection equivalent operation state in FIG. 4A → FIG. 4C → the Δ connection equivalent operation state in FIG. The process shifts to the operation state 4 (f). However, Patent Document 4 shows that the state of FIG. 4A is shifted to the operation state of FIG. 4E through the operation state of FIG. 4G. As shown in Patent Document 4, when vk> 1 and the modulation rate exceeds 1, this can be realized by simply extending a2, b2, and c2 in the opposite directions. However, rather than simply increasing the output composite voltage in the reverse direction, it is possible to finely control the output voltage by enlarging in the reverse direction after passing through the Δ connection equivalent operation state. Then, it is suitable for position sensorless driving in controlling the output combined voltage. Furthermore, since the independent winding type electric motor 3 can be driven with an appropriate output voltage even at low speed to high speed and intermediate speed, it can be driven with high efficiency in a wide range from low speed to high speed.

  As described above, two methods have been described for the output voltage control in the configuration of two inverters in the range of √3 <vk ≦ 2. If this method is used, the independent winding type motor 3 can be controlled while uniquely maintaining the phase relationship between the stator coil axis and the combined voltage axis, and position sensorless driving can be easily realized with a permanent magnet synchronous motor. Further, it is possible to drive while maintaining high efficiency in a wide rotation range.

  Next, the range in which the command modulation factor vk> 2 and vk is large will be described. When the command modulation rate vk increases, the combined voltage that becomes the line-to-line output between the coils becomes a trapezoidal waveform shown by a solid line in which the voltage is distorted as shown in FIG. The fundamental wave component of this combined voltage is sinusoidal as shown by the dotted line in FIG. 9, and an output having an amplitude of 1 or more is possible. This state is called overmodulation output, and driving the motor with overmodulation output is called overmodulation operation. In this embodiment, up to vk ≦ 2, the output line voltage waveform can be output without distortion, but overmodulation operation is required to output a modulation factor vk higher than this.

  First, the overmodulation operation in the case where the command modulation rate vk> 2 is expanded in the above-described method 1 in the Δ connection equivalent operation state will be described first.

  At the time of overmodulation operation, the phase compensator 23 outputs so as to hold θ1 (= 30 degrees) and θ2 (= 150 degrees) immediately before overmodulation, and the modulation rate distributor 22 is connected to the inverters 4 and 5. The distribution is 1: 1 so that the modulation factor vk1 = vk2 = vk / √3. Furthermore, since the Δ-connection equivalent operation state is the output of the line voltage, the output voltage can be controlled without losing linearity even if the third harmonic is superimposed, and there is no problem. For this reason, even in an operation region where vk> 2, overmodulation operation with no difference from a single inverter can be realized even if overmodulation operation is performed by superimposing third harmonics.

  Next, the above-described method 2 will be described. During overmodulation operation with vk> 2, the phase compensator 23 continues to hold the phase angles θ1 = 0 degrees and θ2 = 180 degrees when vk = 2. The modulation rate distributor 22 distributes the modulation ratios of the inverter units 4 and 5 to 1: 1 so that vk1 = vk2 = vk / 2. Thereby, the independent winding type electric motor 3 can be controlled while keeping the phase relationship between the stator coil axis and the combined voltage axis unique.

  When the overmodulation operation is performed in the state of FIG. 4F, unlike the method 1, the third harmonic current flows in the current (ia, ib, ic) flowing in the coil. This is because the neutral point of the independent winding type motor 3 interferes with the inverter units 4 and 5 unlike the Δ connection equivalent state in the state of FIG. In the case of the Δ connection equivalent state, the neutral point of the independent winding type motor 3 does not affect the output voltage of the inverter units 4 and 5 as described above.

  However, in Method 2, when the third-order harmonic is superimposed in the state of FIG. 4F, an operation state in which overmodulation operation is not performed can be created even in the range of 2 <vk ≦ 2.309. However, as described above, when the third-order harmonic is superimposed in the state of FIG. 4F, the third-order component of the line-to-line output voltage is not canceled due to the interference of the inverter units 4 and 5, and the third-order harmonic is generated in the coil. A current including a current flows. Therefore, since there is a concern about an increase in torque ripple and electromagnetic noise generated from the independent winding type electric motor 3, the method 1 described above is more desirable for applications that use the operating range until overmodulation operation. The technique described in No. 4 is not suitable for the operation range up to the overmodulation operation, and the effect of the present embodiment can be understood when the overmodulation operation is considered.

  As described above, regardless of the methods 1 and 2, in the region where the command modulation rate vk> √3, the modulation rate is distributed to the inverter units 4 and 5 so that vk1: vk2 = 1: 1. However, it goes without saying that even if the modulation rate is distributed so that vk = vk1 + vk2, the same effect as described above is obtained. However, when the distribution is not 1: 1, the phase angle compensated by the phase compensator 23 is output so as to output the synthesized voltage while uniquely maintaining the phase relationship between the stator coil axis and the synthesized voltage axis. The calculation is slightly more complicated than the fixed value or the compensation phase angle in the equations (5) and (6).

  For example, when the distribution is 1: 1, vk1 = vk2, and the current values flowing in the inverter units 4 and 5 are the same. Therefore, a semiconductor element or a semiconductor module having the same current capacity can be used, and heat dissipation from the inverter main element. The design can be the same, and a highly reliable motor drive control device can be provided.

  Conversely, in the case where the current capacity of the inverter part on one side is kept low, such as a product with a large current, it may be distributed according to the ratio of the current capacity of the inverter parts 4 and 5, Needless to say, it has the same effect as described above.

  Furthermore, not only the current capacity of the inverter units 4 and 5 but also the distribution ratio may be changed in the middle of overmodulation operation.

  The above output voltage control can be summarized as shown in Table 1 below. In Equation 2, the dq axis voltage of the inverter units 4 and 5, the rotation angle as shown in the block diagram of FIG. The AC voltage output from the two inverter units 4 and 5 is obtained. In Table 1, vk> 2 of method 2 indicates that the third harmonic superposition is present, but since the third harmonic current flowing through the coil is increased, it has the same effect even if it is absent. Therefore, the third harmonic of the current flowing through the coil can be reduced as compared with the case where there is.

  From the above, the configuration of the independent winding type motor 3 using two inverters can output a voltage up to twice the voltage of the sine wave output in the configuration of one inverter without harmonic distortion. As a result, the output voltage can be increased to twice that when one terminal is connected without forming an independent winding, and high-speed rotation and high output can be realized by the increase in the output voltage. Furthermore, it can be rotated at a high speed by the increase in the output voltage, the current flowing through the independent winding type motor 3 can be reduced, and it can be driven with high efficiency in a wide rotation range.

  Next, the efficiency improvement effect when such Y connection and Δ connection, and an intermediate state thereof are created will be described. In a permanent magnet motor, the motor constant changes according to the specifications of the armature winding, but the motor constant changes depending on the connection shape. Table 2 shows motor constants obtained by equivalently converting Δ connections to Y connections with respect to conventional Y connection motor constants (number of turns = N turns), and motor constants for Y connection motors having the same winding specifications.

  From the motor constants in Table 2, the Δ-connection motor in which the N-turn specification Y-connection motor is increased to √3 times the number of turns, the motor constant is the same as the conventional product. Therefore, the operating speed range is the same, and the motor efficiency is the same. On the other hand, the Y-connection motor with the N-turn specification Y-connection motor increased to √3 times the number of windings has a back electromotive voltage of √3 times and resistance and inductance three times that of the conventional product. Therefore, in the case of a Y-connection motor with the number of turns increased to √3 times that of the conventional product, the back electromotive force by the magnet is large, so torque can be output with a small current and efficiency at low speed rotation is improved, but the back electromotive voltage is reduced. Since it is large, field-weakening operation is performed on the high-speed rotation side, and the rotational speed range where the loss is large is narrowed.

  The motor efficiency of the above permanent magnet electric motor is shown in FIG. Although the efficiency of the conventional Y-connection motor is not shown in FIG. 10, this is because the efficiency is the same as that of the Δ-connection motor. Therefore, when the number of turns is set to √3 and a Δ connection is made, it can be driven without changing the efficiency from the conventional product.

  Therefore, the disadvantage that the rotational speed range of the Y-connection motor becomes narrow when the number of turns increases is solved by switching the connection, and the advantage of the high efficiency of the Y-connection motor with the increased number of turns is used during low-speed rotation.

  In the present embodiment, during low-speed rotation, the operation can be performed in the Y-connection equivalent operation state, and the state transition can be made from the Y-connection equivalent operation state to the Δ-connection equivalent operation state from the region where the efficiency decreases due to the Y connection. Therefore, during medium-high speed rotation, it is possible to drive while maintaining the efficiency peak state during Δ connection from the efficiency peak during Y connection.

  Even if it is a method of simply switching between Y connection and Δ connection, it is possible to improve the efficiency at the time of low-speed rotation compared with the conventional product, but in this embodiment, the operation state between Y connection and Δ connection is intermediate. Since the efficiency peak state can be maintained and driven by the creation, the efficiency in the intermediate speed region can be further increased.

  Conventionally, since it is necessary to secure an operating range until high-speed operation, the motor specification that achieves efficiency in the Y connection shown in FIG. 10 does not rotate at high speed. Can be driven in the Δ connection equivalent operation state (method 1 described above) or the Y connection equivalent operation state in the opposite phase (method 2), so that the same rotation speed range as the conventional product can be maintained.

  Therefore, in the present embodiment, a motor with high efficiency at low speed rotation at low speed rotation, a motor capable of high speed rotation at high speed rotation, and two motors, without narrowing the rotation speed range of the independent winding type electric motor 3 conventionally employed. The characteristics can be smoothly changed between the two inverters without impairing the linearity, and the efficiency during low-speed to medium-speed rotation can be greatly improved.

  For example, when the product to be applied is an air conditioner, APF (year-round energy consumption efficiency) exists as an index for displaying the electricity bill performance of the air conditioner. In the case of an air conditioner, it rarely operates at the maximum number of revolutions, and when the room temperature approaches the set temperature, the motor is controlled so that the number of revolutions of the motor is reduced and the room temperature is stabilized. For this reason, the higher the efficiency of the APF at a low speed, the higher the value, and the higher the electricity bill performance. Good electricity bill performance is synonymous with being friendly to the global environment.

  Therefore, APF can be improved when the electric motor control drive device of this Embodiment is used for an air conditioner.

  Furthermore, there is a technique in which a large air conditioner is divided into two compressors, and when it is desired to reduce the air conditioning capacity, the number of compressors to be operated is reduced to one to improve efficiency. In the present embodiment, when the same performance as that of the conventional product is maintained, the counter electromotive voltage on the low speed side can be increased up to √3 times. Therefore, it becomes the same situation as 1 / √3 (= 0.577) unit operation, which is equivalent to stopping one of the two units and operating one unit.

  As described above, the same air conditioning performance as that of the two compressor configuration can be realized by the single compressor configuration. As a result, the structure and control of the refrigerant branch valve, the refrigerant oil return mechanism, and the like are easier than in the case of two units, and a small, inexpensive, and highly reliable air conditioner can be realized. Further, refrigerant leakage and the like can be reduced, and the compression efficiency is improved.

  On the contrary, in the case of the independent winding type motor 3 having the same winding specification as the conventional product, the back electromotive force is smaller than that of the conventional product in the Δ connection equivalent operation state, so the rotation speed range on the high speed rotation side is expanded. However, it is possible to provide an electric motor control drive device that can maintain the same efficiency as that of a conventional winding specification Y-connection in the low-speed rotation.

  Also, although the motor characteristics have changed, the motor constants in the controller can be unchanged and the position sensorless control system is linear because the phase of the stator coil axis and the combined output voltage axis are controlled to be unique. Can be realized without losing. Therefore, it is possible to easily apply a control block in which a conventional electric motor and inverter are configured at 1: 1.

  In the present embodiment, torque control that reduces vibration by controlling output torque following conventional load torque pulsation, on the contrary, constant torque that makes output torque constant without responding to load torque pulsation It has the advantage that motor control can be easily expanded over conventional control, such as control.

  In addition, by superimposing the third harmonic from the Δ connection equivalent operation state, it is possible to output a voltage without distortion up to the same modulation rate as that of the conventional Patent Document 4, and furthermore, it is possible to cope with overmodulation operation.

  Furthermore, in the case of a permanent magnet motor, in the case of the mechanical connection switching method, it was necessary to stop and restart the operation once, but with the configuration of the present embodiment, connection switching can be realized smoothly, and the motor The operation can be continued without stopping, and the efficiency in the intermediate state of the connection state can also be improved.

  In particular, in the case of a compressor used in a freezer for storing foods, food may be thawed or damaged while the motor is stopped. Can be solved.

  Furthermore, since the current flowing through the coil is detected, a coil current that cannot be detected at the time of Δ connection when switching between mechanical Y connection and Δ connection can be detected. Therefore, demagnetization due to overcurrent can be protected by the current flowing through the coil, and an overcurrent cutoff level for demagnetization protection can be set uniquely.

  Further, the AC / DC power conversion unit 2 shown in FIG. 1 is a single-phase AC / DC power conversion unit because the AC power source 1 has a single phase, but the power source is a three-phase AC power source 1b as shown in FIG. It goes without saying that even if the power conversion unit is the three-phase rectifier 2b, the same effects as described above can be obtained as long as the control in the configuration of the inverter units 4 and 5 and the two inverters is the same.

  Further, in FIG. 1, for the position sensorless control of the independent winding type motor 3, the current detector 6 is described as means for detecting the current of the motor 3, but only the current detector 6 is used for the present embodiment. For example, if the control in the two inverters configuration is the same even if it is implemented in any of the DC shunt resistors 7a, 7b, 7c similarly described in FIG. Needless to say, there is an effect.

  Further, as the current detector 6, a magnetic current detector (referred to as “DCCT”) that detects the amount of magnetic flux due to the flowing current, an electric current that flows through the primary winding, and an induction current with a turns ratio on the secondary winding side. There is no difference even if it is implemented by ACCT that detects current by flowing. Furthermore, in DCCT and ACCT, it is predicted that an offset is superimposed on the current detection, but even if this is configured so that the independent winding type motor 3 is corrected while stopped, this embodiment has no effect. Needless to say. Further, offset correction may be performed even when the independent winding type motor 3 is rotating, and in the case of ACCT, the independent winding type motor 3 is driven after passing a current so as to remove the magnetization of the winding core. There is no problem even if it is configured to do so.

This embodiment will be described with reference to the DC shunt resistors 7a, 7b and 7c instead of the current detector 6.
The coil current that can be detected by the DC shunt resistors 7a, 7b, and 7c is similar to the configuration of one inverter that drives a general three-phase motor, at both ends of the DC shunt resistor in relation to the coil current with respect to the voltage vector. Current can be detected. The relationship between the voltage vector and the coil current is as shown in Table 3 below. In Table 3, the relationship between the voltage vector of the shunt resistors 7a and 7b and the coil current is reversed between positive and negative because the sign of the coil current is positive in the arrow direction of the current shown in FIG.

  If only the DC shunt resistor 7a connected to the inverter unit 4 is used, as shown in Table 3, the current flowing in the coil is uniquely determined with respect to the voltage vector output from the inverter unit 4. Since there are at least two different voltage vectors other than the zero vector in one carrier of PWM, two phases of coil current can be detected. Therefore, if there is one DC shunt resistor 7a, current can be detected.

  Next, the DC shunt resistor 7b connected to the inverter unit 5 will be described. In the Y-connection equivalent operation state of the inverter unit 5, the voltage vectors are only v0 and v7, and only the zero vector is output. Therefore, no current can be detected from the shunt resistor 7b. The DC shunt resistor 7b cannot detect the Y-connection equivalent operation state, but has the same effect except for the operation state. Therefore, the current detector 6 cannot be replaced only with the DC shunt resistor 7b, and when the Y-connection equivalent operation state is not used, the current can be detected only with the DC shunt resistor 7b and the independent winding type motor 3 can be driven. High-efficiency driving at low rotation cannot be realized.

  The current detected by the DC shunt resistor 7c is detected as a combined current of the DC shunt resistor 7a and the DC shunt resistor 7b, as is apparent from the circuit of FIG. If the detection current of the DC shunt resistor 7a is Idc1 and the detection current of the DC shunt resistor 7b is Idc2, the detection current of the DC shunt resistor 7c is determined from the relationship between the voltage vector in Table 3 and the coil current detected by the DC shunt resistor. Idc is given by Idc = Idc1 + Idc2.

  For example, when the voltage vector output from the inverter unit 4 is V5 and the voltage vector output from the inverter unit 5 is V6, the detected current Idc of the DC shunt resistor 7c is as shown in Table 3. Idc = Idc1 + Idc2 = −Ib + Ia, which is detected as a combined current of the coil currents.

  Therefore, in the Y-connection equivalent operation state, the voltage vector output from the inverter unit 5 is only the zero vector, so Idc2 = 0 and Idc = Idc1. Therefore, there are two different voltage vectors that are not zero vectors in one carrier, and the current can be detected only by the DC shunt resistor 7c.

Next, the current that can be detected by the DC shunt resistor during the transition from the Y-connection equivalent operation state to the Δ-connection equivalent operation state will be described.
The modulation factor vk1 = 1 and θ1 of the inverter unit 4 operate according to the equation (4), and the modulation factor vk2 of the inverter unit 5 operates according to the equation (3) and θ2 = 150 degrees. Therefore, the duty ratios of the inverter units 4 and 5 are different, and the DC shunt resistor 7c of the other voltage vectors is used except for the case where both inverter units 4 and 5 are zero vectors in the half carrier as shown in FIG. Current detection is possible at both ends. In the drive signal of FIG. 12, Hi indicates the upper arm ON, and Lo indicates the lower arm ON.

  The change timings of the drive signals of the inverter units 4 and 5 do not coincide with each other, and a voltage vector capable of detecting current is generated five times, and all of these hold Idc = Idc1 + Idc2. Therefore, conventionally, it is possible to detect all three-phase coil currents without restoring the coil current from Idc by setting three-phase coil current = 0 from three-phase equilibrium. Therefore, not only can the current be detected only by the DC shunt resistor 7c, but also the zero-phase current can be detected by using the DC shunt resistor 7c.

  Zero-phase current does not flow if it is three-phase balanced, so the independent winding motor 3 is not three-phase balanced, there is variation between coils, or there is an output error in the two inverter units 4 and 5. It means that there is. In Patent Document 2, a configuration in which a switch is provided for the zero-phase current is employed. However, the zero-phase current is controlled to be zero by detecting the zero-phase current from the current flowing through the DC shunt resistor 7c. It becomes possible.

  Further, in the case of the Δ connection equivalent operation state in the configuration of only the DC shunt resistor 7c, the phase difference of the output voltage in the inverter units 4 and 5 is 120 degrees, so that the duty ratio matches, and the drive signals of the inverter units 4 and 5 The change timing of is simultaneous. Therefore, there are two voltage vectors in which current can be detected in the half carrier, which is the same as the conventional 1: 1 inverter and motor configuration.

  From the above, even with the configuration of only the DC shunt resistor 7c, it is possible to detect two phases of the coil current, and it can be said that the current can be detected if there is one DC shunt resistor 7c. Further, during the transition from the Y-connection equivalent operation state to the Δ-connection equivalent operation state, the zero-phase current can also be detected, and the zero-phase current can be controlled to be zero.

  Although FIG. 1 shows the current detector 6 and the DC shunt resistors 7a to 7c, even if only one of them is configured, it is sufficient if the coil current of the independent winding type motor 3 can be detected. Needless to say, the present embodiment can be realized even if the current detector 6 and the DC shunt resistor 7c are configured. Furthermore, it goes without saying that any method can be used as long as the coil current of the independent winding type motor 3 can be detected without using the current detector 6 or the DC shunt resistors 7a to 7c.

  In the case of the configuration with only the DC shunt resistor 7c, it is possible to detect the zero-phase current during the transition from the Y-connection equivalent operation state to the Δ-connection equivalent operation state, but the zero-phase current can be generated even in the Y-connection equivalent operation state. There is sex. In view of this, the DC shunt resistor 7c and the current detector 6 may be used in combination to detect the zero-phase current and control the zero-phase current to stop flowing.

  In this case, there is no problem even if the configuration is such that the zero-phase current is detected by the combination of the DC shunt resistor 7a and the current detector 6 instead of the DC shunt resistor 7c and the zero-phase current does not flow. Needless to say.

  In FIG. 1, two current detectors 6 are provided for the three-phase independent-winding motor 3. However, if the number of the current detectors 6 is the same as the number of phases, zero-phase current can be detected. Needless to say, any configuration may be used as long as the zero-phase current can be suppressed by controlling the output voltage regardless of the detection means for suppressing the zero-phase current.

  Further, the zero phase current is generated by circulating through the inverter units 4 and 5. In order to suppress the circulation, the inverter units 4 and 5 may be connected via diodes as shown in FIG. Since no reverse current flows through the diode, the circulating current is suppressed, and as a result, the zero-phase current can be suppressed. Although FIG. 13 shows a single-phase rectifier, there is no problem even if the three-phase rectifier 2b shown in FIG. 11 is used. Further, even if the independent winding type electric motor 3 is driven without using an electrolytic capacitor by using a small-capacity film capacitor as the capacitors 8a and 8b in FIG. 13, the configurations of the 1: 1 inverter and the motor are the same. The independent winding type motor 3 can be driven by the operation, and furthermore, even if the capacity of the capacitor 8 of the three-phase rectifier 2b shown in FIG.

  In addition, the present embodiment is a technique related to the drive control of the independent winding type electric motor 3, but the control can be easily applied as long as the control can be realized by the configuration of the 1: 1 inverter and the motor. For example, a compressor used in an air conditioner has a load torque pulsation during one rotation as shown in FIG. Torque pulsation suppression control for controlling an inverter so as to suppress this load torque pulsation has been put into practical use. Accordingly, since the present embodiment is configured to distribute the output voltage to the two inverters in the portion where the dq axis voltage is converted into the output voltage, if the torque pulsation suppression control has been put into practical use, the control is followed as it is. Thus, it can be applied to a configuration of two inverters.

  So far, the DC shunt current detection and torque pulsation suppression control have been described, and the technology related to the drive control of the independent winding type electric motor 3 has been described. The same applies to the beat phenomenon suppression control in which the power source frequency and the motor frequency interfere with each other and the electric current of the motor pulsates, and the chemi-conless control to reduce the capacity of the capacitor. Any control configuration that can be realized by the configuration is easy to apply to the present embodiment, and the effect is equivalent.

  Furthermore, when applied to the drive of a compressor mounted on an air conditioner, it has been explained that the APF improvement and the configuration of two compressors can be realized by one unit, but the torque pulsation during one rotation of the compressor is explained. If the compressor is of a small scroll type, the current fluctuation during one rotation becomes small, and the inductance change that fluctuates due to the current flowing is mechanically suppressed, so that the three-phase equilibrium state can be mechanically maintained. There is an advantage that the zero phase current can be reduced without the suppression control of the zero phase current.

  In addition, when applied to an air compressor that compresses air rather than a compressor that uses refrigerant as in an air conditioner, the amount of heat transferred by the compression process is less than that of the refrigerant compressor, and the characteristics of the magnet whose characteristics change with temperature This also stabilizes the three-phase equilibrium state mechanically. Therefore, it has the same effect as described above. Further, a compressor that compresses carbon dioxide instead of air has not only the same effect but also a natural refrigerant having an ozone depletion coefficient of 0, and therefore has an advantage that is friendly to the global environment.

  Furthermore, the permanent magnet used as a permanent magnet motor uses a ferrite magnet, but if this is changed to a rare earth magnet, the maximum energy product can be increased and the magnetic force can be increased, so the demagnetization level is improved. In addition, variations in magnets due to demagnetization can be suppressed. In addition, the back electromotive force constant φ, which is a motor constant, can be easily suppressed from being varied by a high magnetic force. As a result, the three-phase equilibrium state can be mechanically maintained, and the zero-phase current can be reduced without the suppression control of the zero-phase current. In addition, a permanent magnet motor produced by mixing another powder with a magnet has higher formability than a sintered magnet and can easily suppress variations between three phases, so that it can be said that a zero-phase current hardly flows.

  Furthermore, although the number of phases described above has been described as being three phases, the phase relationship between the magnetic flux axis vector and the composite voltage vector is not limited to three phases, and even if it is multiphase. Needless to say, this embodiment can be applied as long as it is kept constant. Furthermore, up to six phases, the phase difference angle of each phase axis is 60 degrees or more, and can be applied without paying attention to each winding. When the number of phases is more than that, the effect of the present embodiment is not applied unless the combined voltage is smaller than the voltage before the combination.

  From the above, the zero-phase current can be reduced by maintaining the three-phase equilibrium state by suppressing the circulating current or the variation between each phase by the circuit configuration, the structure of the motor, the structure of the machine equipped with the motor, etc. It can also be suppressed. In addition, the zero-phase current can be suppressed by detecting the zero-phase current itself and correcting the output according to the variation of each phase.

  In addition, since this embodiment has two inverter main circuit parts, it is considered that the cost is simply doubled. However, the current flowing through the coil is different between the Y connection and the Δ connection. √3: 1. For this reason, the inverter main circuit portion requires two main circuit portions having a current capacity of 1 / √3 of the current capacity used in the single-unit configuration. Therefore, products having a large current capacity, such as commercial air conditioners and showcases, can reduce the current capacity and can be used for inexpensive parts for consumer products. In this case, it is added that the cost difference between two inexpensive main circuit units and one expensive main circuit unit is the cost difference of the present embodiment, and there is a possibility of cost reduction.

  As an example of use of this embodiment, in addition to an air conditioner equipped with a compressor in which position sensorless is essential, there are a refrigerator, a dehumidifier, a heat pump type hot water heater, a heat pump type drying washing machine, a refrigerator, and a showcase. Can be mentioned. Furthermore, the present invention can be applied to products not equipped with a compressor such as a washing / drying machine, a washing machine, and a vacuum cleaner, and can also be applied to a fan motor or the like.

It is a circuit diagram which shows the structure of the electric motor drive control apparatus which concerns on embodiment of this invention. It is a connection circuit diagram for demonstrating this Embodiment. It is a block diagram which shows the equivalent circuit for demonstrating this Embodiment. It is a voltage vector figure which shows the operation state of this Embodiment. It is a control block diagram with one inverter structure explaining this Embodiment. It is a control block diagram changed with two inverter structure in this Embodiment. It is a voltage vector figure which defines the modulation factor in this Embodiment. It is a voltage waveform diagram for demonstrating the 3rd harmonic superimposition in this Embodiment. It is a voltage waveform diagram for demonstrating the time of the overmodulation in this Embodiment. It is an efficiency curve figure in this Embodiment. It is a circuit block diagram which shows the other AC / DC power converter in this Embodiment. It is a wave form diagram for demonstrating the shunt detection of the synthetic current in this Embodiment. It is a circuit block diagram which shows other embodiment in this Embodiment. It is a wave form diagram for demonstrating the torque pulsation in this Embodiment.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 AC power supply, 2 AC / DC power conversion part, 3 Independent winding type motor, 4 1st inverter part, 5 2nd inverter part, 21 Output voltage generation part, 22 Modulation rate divider | distributor, 23 Phase compensator, 24a 1st Coordinate conversion unit for the inverter unit, 24b Coordinate conversion unit for the second inverter unit.

Claims (16)

AC / DC power converter for converting AC power to DC power;
First and second inverter units that generate AC voltages from the DC power of the AC / DC power conversion unit and output the generated AC voltages to the independent winding type motor;
Control means for controlling the first and second inverter units so that the independent winding type motor performs Y-connection equivalent operation or Δ-connection equivalent operation,
The control means includes a vector of magnetic flux generated in the armature winding of the independent winding type motor and the first and second inverters when the independent winding type motor shifts from the Y-connection equivalent operation state to the Δ connection equivalent operation state. The motor drive control device, wherein the output voltage of the first and second inverter units is controlled so that the phase with the vector of the synthesized voltage by the unit maintains a parallel relationship. The control means according to claim 1, characterized in that to detect only current flowing in the armature winding of the independent wire-wound motor, for controlling the output voltage of the first and second inverter based on this serial mounting of the motor drive control device. Said control means, said independent winding without detecting the position of the rotor of the linear motor, the motor drive control apparatus according to claim 2, wherein the drive controlling the independent wire-wound motor. 4. The electric motor according to claim 2, further comprising a current detector that detects a magnetic flux generated by a current flowing through an armature winding of the independent winding type motor and detects an armature current based on the detected magnetic flux. Drive control device. Detecting an induced current generated by the current flowing in the armature winding of the independent wire-wound electric motor, according to claim 2 or 3, wherein further comprising a current detector for detecting the armature current based on this Electric motor drive control device. 4. The motor drive control device according to claim 2, further comprising a current detector that detects an armature current from a relationship between an output voltage of the first and second inverter units and a voltage across the resistor.   7. The motor drive control device according to claim 1, wherein the control means sets the output of the second inverter unit to zero voltage until the output voltage of the first inverter unit is saturated. .   The control means changes only the phase while maintaining the output voltage of the first inverter unit at a saturated value until the output voltage of the first inverter unit is saturated and the output voltage of the second inverter unit is saturated. The motor drive control device according to any one of claims 1 to 6, wherein only the amplitude is changed while the phase of the output voltage of the second inverter unit is maintained.   When the output voltage of the second inverter section is saturated, the control means superimposes the third harmonic of the output frequency while maintaining the phase and amplitude of the first and second inverter sections. The motor drive control device according to any one of claims 1 to 6, wherein a voltage is output.   When the output voltage of the second inverter section is saturated, the control means outputs a voltage by changing the phase while maintaining the amplitudes of the first and second inverter sections. The electric motor drive control device according to claim 1.   An independent-winding motor that is driven and controlled by the motor drive control device according to any one of claims 1 to 10, wherein a permanent magnet is used for the rotor.   When the armature winding is Y-connected by Y-connection equivalent conversion, it does not reach a predetermined rotational speed, but when it is Δ-connected by Δ-connection equivalent conversion, it can be rotated to a predetermined rotational speed. The electric motor according to claim 11.   The electric motor according to claim 11 or 12, wherein a rare earth magnet is used as the permanent magnet of the rotor.   The electric motor according to claim 11 or 12, wherein the permanent magnet of the rotor is formed by adding powder.   15. The electric motor according to claim 11, wherein the electric motor is mounted on a scroll type compressor.   The electric motor according to claim 11, wherein the electric motor is mounted on a compressor using a refrigerant having an ozone depletion coefficient of zero.

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