JP3277637B2 - Inverter controlled welding power supply - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply for arc welding of an inverter control type, and provides a stable arc welding machine which is stable and has a low no-load voltage with respect to arc welding accompanied by severe load fluctuation. What you want to do.

[0002]

2. Description of the Related Art Conventionally, a forward converter has been used as an inverter control method with stable operation in an arc welding machine having a severe load fluctuation. FIG. 1 shows an example of a two-stone type forward converter, and FIG. 2 shows an operation waveform thereof. In FIG. 1, 1 is a DC power supply, 2 is an inverter 1
The next circuit, 3 is an inverter transformer, 4 is an inverter secondary circuit, 5 is a torch, and 6 is a base material. 2A and 2B are diagrams for explaining the operation of the device shown in FIG. 1, wherein FIG. 2A shows the base currents of the transistors Tp1 and Tp2, and FIG.
1, the collector current of Tp2, (c) is the transistor Tp1,
Tp2 collector-emitter voltage, (d) current through rectifier diode Ds1, (e) flywheel diode D
(f) is the voltage of the flywheel diode Ds2, and (g) is the current of the DC reactor L, that is, the load current.

In FIGS. 1 and 2, when a base current flows as shown in FIG.
2 flows through each transistor, and the collector-emitter voltage becomes as shown in FIG. While the base current flows through the transistors Tp1 and Tp2 and is ON, a forward voltage is applied to the rectifier diode Ds1 to conduct, a reverse voltage is applied to the flywheel diode Ds2, and the secondary current is the rectifier diode Ds1 and the DC reactor L , A torch 5, an arc, a base material 6, and a rectifier diode Ds1 on the secondary side of the inverter transformer 3.

When the base currents of the transistors Tp1 and Tp2 are exhausted, these transistors are turned off, and a voltage opposite to that at the time of being turned on appears on the primary and secondary windings of the inverter transformer 3. On the primary side, this voltage causes the diode Dp
3, Dp4 becomes conductive, and the inverter transformer 3 returns the magnetic energy stored at the time of ON to the DC power supply 1. Next ON
By the time, the feedback of the magnetic energy is finished, and the voltage appearing in the winding of the inverter transformer disappears. On the other hand, during this time, the rectifier diode Ds1 is turned off and the flywheel diode Ds2 is turned on, and the magnetic energy stored in the DC reactor L supplies power to the load. At this time, the secondary current flows through the circuit of the DC reactor L, the torch 5, the arc, the base material 6, the flywheel diode Ds2, and the DC reactor L.

[0005]

Assuming that the output voltage of the DC power supply is Eo and the times when the transistors Tp1 and Tp2 are ON and OFF are Ton and Toff, respectively, the voltage of the flywheel diode Ds2 is as shown in FIG. And the current flows at the time of Toff. Assuming that the number of turns of each of the primary and secondary windings of the inverter transformer 3 is Np, Ns, the maximum value Vmax of the load voltage is

Vmax = Eo. (NS / NP)

The average value Vavr of the load voltage is

Vavr = Eo. (NS / NP). (Ton /
(Ton + Toff))

[0008] Therefore,

(Vmax / Vavr) = (Ton + Toff) / Ton

## EQU1 ## However, in the forward converter, it is necessary to provide a time for returning the magnetic energy at least for a time required for storing the magnetic energy, and the ON DUTY is limited to 50% or less. That is,

Ton / (Ton + Toff) <0.5

To be

(Vmax / Vavr) = (Ton + Toff) / Ton> 2

## EQU1 ## On the other hand, other inverter systems

Ton / (Ton + Toff) <100%

Therefore,

(Vmax / Vavr) = (Ton + Toff) / Ton> 1

## EQU1 ##

By the way, even if the welding worker wears safety shoes and wears insulating gloves, there is always a risk that a voltage is applied between the torch and the base material. When the inverter is operating and no arc is generated, a small-capacitance capacitor that exists as a stray capacitance between the base material and the torch or a small-capacity capacitor connected between the output terminals to prevent intrusion of noise from the outside. It is charged at the aforementioned Vmax, which appears as a no-load voltage. If the no-load voltage is high, it is dangerous for the welding operator. However, as described above, the forward converter has ON DUTY, that is, Ton / (Ton + Toff) of 50% or less, so that the ON DUTY is almost 100%. Approximately twice as high Vmax is required, resulting in about twice the no-load voltage, which has the disadvantage of being very dangerous.

[0019]

According to the present invention, a plurality of sets of forward converters are prepared in order to solve the above-mentioned problems of the conventional device, the secondary side is connected in parallel, and each forward converter is sequentially turned on and off. By controlling, the overall ON DUTY is improved, and the difference between the average voltage and the maximum voltage is reduced.

[0020]

BRIEF DESCRIPTION OF THE DRAWINGS FIG.

FIG. 3 shows a first embodiment of the present invention. The inverter primary circuits 2a and 2b correspond to the inverter primary circuit 2 of FIG.
The inverter transformers 3a and 3b have the same configuration as the inverter transformer 3 of FIG. 1, and the inverter secondary circuits 4a and 4b have the same configuration as the inverter secondary circuit 4 of FIG. FIG.
FIG. 4 is a diagram for explaining the operation of the embodiment in FIG. 3. In FIG. 4, (aa) is the base current of the transistors Tp1, Tp2 of the inverter primary circuit 2a, (ba) is the collector current of the transistors Tp1, Tp2 of the inverter primary circuit 2a, and (ca) is the current of the inverter primary circuit 2a. Collector-emitter voltages of transistors Tp1, Tp2, (ab) is the base current of transistors Tp1, Tp2 of inverter primary circuit 2b, (bb) is the collector current of transistors Tp1, Tp1, Tp2 of inverter primary circuit 2b, ( cb) is the collector-emitter voltage of the transistors Tp1, Tp2 of the inverter primary circuit 2b, and (da) is the inverter secondary circuit 4.
(a) is the current of the rectifier diode Ds1, (ea) is the current of the flywheel diode Ds2 of the inverter secondary circuit 4a,
(Fa) is the voltage of the flywheel diode Ds2 of the inverter secondary circuit 4a, and (db) is the inverter secondary circuit 4
b, the current of the rectifier diode Ds1, (eb) is the current of the flywheel diode Ds2 of the inverter secondary circuit 4b,
(Fb) is the voltage of the flywheel diode Ds2 of the inverter secondary circuit 4b, and (ga) is the inverter secondary circuit 4
a, the current of the DC reactor L, (gb)
The current of the DC reactor L of the next circuit 4b, and (h) is the load current.

(Aa) and (ab), (ba) and (bb), (ca) and (cb), (da) and (db) in FIG.
(Ea) and (eb), (fa) and (fb), and (ga) and (gb) have the same waveform, respectively, but the phases are shifted from each other by a half cycle. The load voltage is expressed as (ga) through the DC reactor L of the inverter secondary circuit 4a.
b) appears through the DC reactor L of the inverter secondary circuit 4b. Therefore, the maximum value of the load voltage is V1max, and the average value of the load voltage is 2 · V1max · (Ton / (Ton
+ Toff)). Since the average value of the load voltage is a required value for welding and may be the same as that of the circuit of FIG. 1, Vmax = 2 · V1max. From this, the turns ratio (secondary / 1) of the inverter transformers 3a and 3b in FIG.
Next, the half of the inverter transformer 3 in FIG. This indicates that the no-load voltage of the circuit of FIG. 3 is one half that of the circuit of FIG. 1, and that the circuit of FIG. 3 is safer for the operator than the circuit of FIG. Things.

The inverter transformers 3a, 3a,
The maximum value of each current flowing through the secondary winding of the inverter 3b is also one half of that of the device of FIG. 1, and taking the above-mentioned turn ratio into account, the maximum current flowing through the primary winding of each inverter is as shown in FIG. 2/4. Therefore, the total capacity of the two inverter transformers in FIG. 3 is one half of the capacity of one inverter transformer in FIG. Also, in the case of FIG. 3, the product of the rated current and the rated voltage required for the transistors constituting each inverter can be obtained by preparing two one-fourth of the case of FIG.

FIG. 5 shows an example of a control circuit used in the embodiment shown in FIG. In the figure, 7 is a clock pulse generator, 8 is a control pulse generator, and 9a and 9b are transistor drive circuits. FIGS. 6A and 6B are diagrams for explaining the operation of the control circuit of FIG. 5, in which FIG. 6A shows a clock pulse Vck, FIG. 6B shows a control pulse Vca, FIG. 6C shows a transistor drive pulse Ip1a, and FIG. d) is a transistor drive pulse Ip2a, (e) is a control pulse Vcb, (f) is a transistor drive pulse Ip1b, and (g) is a transistor drive pulse Ip2b.

The clock pulse generator 7 generates a clock pulse Vck at a constant cycle and inputs the clock pulse Vck to the control pulse generator 8. The control pulse generator 8 divides the frequency of the clock pulse to generate the transistor Tp of the inverter primary circuit 2a of FIG.
1, Tp2 and the control pulse Vca corresponding to the maximum ON DUTY of each of the transistors Tp1, Tp2 of the inverter primary circuit 2b.
And Vcb to generate the transistor drive circuits 9a, 9b
To enter. The transistor driving circuits 9a and 9b output pulses Ip1a and Ip2a and pulses Ip1b and Ip2b having a time width corresponding to another input, that is, the output control signal Vcnt, in synchronization with the rise of the control pulses Vca and Vcb, and output the inverter 1 shown in FIG. The transistors Tp1 and Tp2 of the secondary circuit 2a and the transistors Tp1 and Tp2 of the inverter primary circuit 2b are driven. The transistor drive pulses Ip1a, Ip1a
The circuits p2a and Ip1b and Ip2b are insulated from each other.

Here, a clock pulse Vck is input as a control pulse generator, and this input pulse is divided into two phases to generate two sets of pulse trains whose ON DUTY is less than 50% and whose phases are mutually shifted by 180 degrees. Any circuit may be used. For this purpose, for example, an integrated circuit for controlling a stepping motor (such as TD62803P manufactured by Toshiba Corporation) can be used. FIG. 7 shows an embodiment of the control pulse generator 8. In the figure, IC1 is an integrated circuit for controlling a stepping motor,
Rck1 and Rck2 are resistors, and SWP is a switch. In the figure, when a clock pulse Vck is input, control pulses Vca and Vcb whose phases are different from each other by 180 degrees are obtained.

FIG. 8 shows the control pulse Vca and the output control signal Vcn.
The transistor drive circuit 9a which receives t as an input and obtains transistor drive pulses Ip1a and Ip2a corresponding to the input signal.
It is a connection diagram which shows the example of (a). In the figure, OP1 to OP3 are operational amplifiers, E1 to E5 are control power supplies, PC
1 to PC3 are photocouplers, TR11 to TR14 are transistors, DR1 to DR3 are diodes, AN
D1 is an AND gate. In the figure, an operational amplifier OP1 and resistors R1 and R2 constitute an inverting amplifier. Further, a resistor R3 and a control power supply E1 are connected in series with the polarity shown in the drawing to the minus input terminal of the operational amplifier OP1. The output of the operational amplifier OP1 is negative when the control pulse Vca is positive, and positive when the control pulse Vca is zero. The output of the operational amplifier OP1 is input to an integrator composed of an operational amplifier OP2, a resistor R4 and a capacitor C1, and a reset circuit for the integrator composed of a resistor R5 and a photocoupler PC1.

FIG. 9 is a diagram for explaining the operation of the transistor drive circuit 9a of FIG.
(A) is the control pulse Vca, (b) is the output voltage of the operational amplifier OP1, (c) is the output voltage Vth of the operational amplifier OP2,
(D) is the output voltage of the operational amplifier OP3, (e) is AND
1 output voltage. 8 and 9, when the control pulse Vca is zero, the output of the operational amplifier OP1 becomes positive, and current flows through the resistor R5 and the light emitting diode of the photocoupler PC1. Therefore, the output transistor of the photocoupler PC1 is turned on to discharge the electric charge of the capacitor C1 and reset the integration circuit. When the control pulse Vca is positive, the output of the operational amplifier OP1 becomes negative, and no current flows through the resistor R5 and the light emitting diode of the photocoupler PC1. Therefore, the photocoupler PC
The output transistor No. 1 is turned off, and the integrating circuit performs an integrating operation. As a result, a triangular wave Vth shown in FIG. 8C is obtained as an output of the integrating circuit.

The triangular wave Vth is output from the output control signal Vcnt, the operational amplifier OP3, the resistors R6 and R7 and the diode D
The data is input to a comparator composed of R2 and compared. This comparator outputs plus when Vth <Vcnt, and minus when Vth> Vcnt. The output of this comparator is AND gate A
It is applied to one input terminal of ND1. A control pulse Vca is applied to another input terminal of the AND gate AND1. As a result, when the control pulse Vca is positive and the output of the comparator is positive, the AND gate AND
The output of 1 becomes positive, and a current flows to the light emitting diodes of the photocouplers PC2 and PC3 through the resistor R8.
Photocouplers PC2 and P having these light emitting diodes
The output transistors of C3 are control power supplies E2 and E, respectively.
3, resistors R9, R10, transistors TR11, TR12
In addition, control power sources E4, E5, resistors R11, R12, and transistors TR13, TR14 are used, and transistor drive pulses Ip1a, Ip2a are generated.

The transistor driving circuit 9b in FIG. 5 is the same as the transistor driving circuit 9a, except that the input control pulse Vcb has a phase difference of 1/2 cycle from Vca, so that the transistor driving pulses Ip1a and Ip2a are 180 degrees out of phase. The shifted transistor drive pulses Ip1b and Ip2b are applied to the transistor drive circuit 9b.
Output.

FIG. 10 shows a second embodiment of the present invention. In the embodiment shown in FIG. 3, the flywheel diodes Ds1 and Ds2 of the inverter secondary circuits 4a and 4b and the DC reactor L are taken out and one of them is taken out. Is used in common. As shown in FIG. 11, the waveforms (e) and (f) of the flywheel diode Ds2 and the waveform (g) of the DC reactor L are different from the diagram of FIG. 4 illustrating the operation of the embodiment of FIG. A detailed description is omitted because it is only different.

FIG. 12 shows a third embodiment of the present invention. The difference between the third embodiment and the first embodiment is a DC reactor. In the first embodiment, the DC reactor L of the inverter secondary circuit 4a and the DC reactor L of the inverter secondary circuit 4b are independent from each other on the inverter a side and the inverter b side, whereas in the third embodiment, they are magnetically coupled to each other. A DC reactor Lsc is used, and currents flowing through the respective windings of Lsc (1/2) on the inverter a side and Lsc (2/2) on the inverter b side are magnetically coupled so as to cancel out magnetic flux.

FIG. 13 is a diagram for explaining the operation of the embodiment of FIG. 12 and 13, a DC power supply 1, a torch 5 and a base material 6, an inverter primary circuit 2
a and 2b operate the same as in FIG. 3 of the first embodiment. The operation waveforms are shown in (aa) to (ca) and (a) of FIG.
b) to (cb). The operations of (aa) to (ca) and the operations of (ab) to (cb) in FIG. 13 are shifted from each other by a half cycle. The operation waveforms of the rectifier diode Ds1 and flywheel diode Ds2 of the inverter secondary circuit 4a and the rectifier diode Ds1 and flywheel diode Ds2 of the inverter secondary circuit 4b are shown in FIG.
a), (ea) and (db), (eb).

Transistor T of inverter primary circuit 2a
When p1 and Tp2 are ON and the transistors Tp1 and Tp2 of the inverter primary circuit 2b are OFF, the inverter secondary circuit 4
The current of the rectifier diode Ds1 in (a) is the same as the current flowing in the coupled DC reactor Lsc (1/2). The current flowing through the circuit of the coupled DC reactor Lsc (2/2), the torch 5, the arc, the base material 6, the flywheel diode Ds2 of the inverter secondary circuit 4b, and the coupled DC reactor Lsc (2/2) is represented by the coupled DC reactor Lsc. It is the same as the current flowing through (1/2).
The transistors Tp1 and Tp2 of the inverter primary circuit 2a are O
In the FF, the transistors Tp1,
When Tp2 is ON, the current of the rectifier diode Ds1 of the inverter secondary circuit 4b is the same as the current flowing through the coupled DC reactor Lsc (2/2). At this time, the coupled DC reactor Lsc
(1/2), the torch 5, the arc, the base metal 6, the flywheel diode Ds2 of the inverter secondary circuit 4a, and the current flowing through the coupled DC reactor Lsc (1/2) are coupled DC reactor Lsc (2/2 ).

Transistor T of inverter primary circuit 2a
When p1 and Tp2 are OFF and the transistors Tp1 and Tp2 of the inverter primary circuit 2b are also OFF, the coupled DC reactor Lsc (1/2), torch 5, arc, base material 6, and inverter 2
The current flowing through the circuit of the flywheel diode Ds2 and the coupled DC reactor Lsc (1/2) of the next circuit 4a, the current flowing through the coupled DC reactor Lsc (2/2), the torch 5, the arc, the base metal 6, and the inverter secondary circuit 4b Flywheel diode Ds
2. The current flowing through the circuit of the coupled DC reactor Lsc (1/2) is the same. Voltage waveforms of the flywheel diode Ds2 of the inverter secondary circuit 4a and DRS2 of the inverter secondary circuit 4b are shown in (fa) and (fb). The voltage of the flywheel diode Ds2 of the inverter secondary circuit 4a becomes V3max when the transistors Tp1 and Tp2 of the inverter primary circuit 2a are ON. The voltage of the flywheel diode Ds2 of the inverter secondary circuit 4b is equal to the inverter primary circuit 2
It becomes V3max when the transistors Tp1 and Tp2 of b are ON.

Therefore, the maximum value of the load voltage is V3max, and the average value of the load voltage is 2 · V3max · (Ton / (Ton /
+ Toff)). Since the average value of this load voltage may be the same as that of the circuit of FIG. 1, Vmax = 2 · V
3max. This indicates that the no-load voltage of the circuit of FIG. 12 is half that of the circuit of FIG. 1, as in the circuit of FIG. 3, and the circuit of FIG. 12 is safer than the circuit of FIG. It shows that.

[0037]

[Other Embodiments] The above-mentioned Embodiments 1 to 3
Although the description has been made with reference to the two-stone type forward converter, the same can be said for the one-stone type forward converter. Also,
The input of each inverter primary circuit may be connected to a separate DC power supply.

The output of the DC power supply may be divided by a capacitor, and the input of each inverter primary circuit may be connected to each capacitor. FIG. 14 is a connection diagram showing an example of such a case. In the figure, C1 and C2 are capacitors connected to the output of the DC power supply 11 and divide the output of the DC power supply 11 into two. The terminal voltage of each capacitor is supplied to each of the inverter primary circuits 2a and 2b. The other parts of the figure are the same as those of the embodiment shown in FIG.

FIG. 15 shows that, in addition to the embodiment of FIG.
Divide the input terminals of each inverter primary circuit in parallel and connect them directly to the output terminals of the DC power supply, or
Switch S for connecting the input terminal of the next circuit to each capacitor that divides the output voltage of the DC power supply into two
WP is provided. In the case of FIG.
A common device can be used by selecting the changeover switch Swp for two types of power supply voltages, such as 0 V or 400 V, whose input voltages have a one-to-two relationship with each other.

Although the above description has been made with reference to two sets of inverter circuits, a structure in which three or more sets of inverter circuits are used and the secondary sides are connected in parallel may be used. In this case, the control pulse generators each have a phase that is one-third of a cycle (12
It is only necessary to prepare three sets of transistor driving circuits that obtain three control pulses shifted by 0 degrees and generate pulses having a time width corresponding to the output control signal in synchronization with the control pulses.

[0041]

According to the present invention, as described above, the no-load voltage is reduced to about one half compared with the conventional inverter-controlled arc welding machine using the forward converter.
The safety can be improved and the total capacity of the inverter transformer can be reduced to almost half.

According to the second aspect of the present invention, it is possible to provide an inverter-controlled arc welding machine using a forward converter capable of coping with a high input voltage with a transistor having a lower voltage rating.

The third aspect of the present invention further provides, for example,
It is possible to provide a forward converter inverter-controlled arc welding machine that can be shared with two types of input voltages having a substantially one-to-two relationship such as 00V.

In each of the fourth and fifth aspects of the present invention, the flywheel diode and the DC reactor or the DC reactor of the secondary circuit of the inverter can be integrated into a single component, so that the number of components can be reduced.

[Brief description of the drawings]

FIG. 1 is a connection diagram showing a circuit of a conventional two-stone forward converter,

FIG. 2 is a diagram showing operation waveforms of the forward converter of FIG. 1;

FIG. 3 is a connection diagram showing a first embodiment of the present invention;

FIG. 4 is a diagram for explaining the operation of the apparatus of the embodiment of FIG. 3,

FIG. 5 is a connection diagram illustrating an example of a control circuit used in the embodiment of FIG. 3;

6 is a diagram for clarifying the operation of the control circuit of FIG. 5,

7 is a connection diagram showing an example of a control pulse generator 8 used in the control circuit of FIG. 5,

8 is a connection diagram illustrating an example of a transistor drive circuit 9a used in the control circuit in FIG. 5;

FIG. 9 is a diagram for explaining the operation of the transistor drive circuit 9a in FIG. 8;

FIG. 10 is a connection diagram showing a second embodiment of the present invention,

11 is a diagram for explaining the operation of the apparatus of the embodiment of FIG. 10,

FIG. 12 is a connection diagram showing a third embodiment of the present invention;

FIG. 13 is a diagram for explaining the operation of the apparatus of the embodiment in FIG. 12,

FIG. 14 is a connection diagram showing a fourth embodiment of the present invention;

FIG. 15 is a connection diagram showing a fifth embodiment of the present invention.

[Explanation of symbols]

 1,11 DC power supply D1 to D6 Rectifier C, C1, C2 Smoothing capacitor Tm1 to Tm3 AC input terminal 2,2a, 2b Inverter primary circuit Tp1, Tp2 transistor TR11 to TR14 transistor Dp1, Dp2, Dp3, Dp4 Diode 3 Inverter Transformers 3a, 3b Inverter transformer 4 Inverter secondary circuit 4a, 4b Inverter secondary circuit 41a, 41b Inverter secondary circuit 42a, 42b Inverter secondary circuit Ds1 Rectifier diode Ds2 Flywheel diode L DC reactor Lsc Coupling DC reactor 5 Torch 6 mother Material 7 Clock pulse generator 8 Control pulse generator IC1 Stepping motor control IC Rck1, Rck2 Resistor Swp switch 9a Transistor drive circuit 9b Transistor drive circuit OP1 to OP3 Operational amplification It E5 control power PC1 to no E control power E1 to PC3 photocoupler AND1 AND gates DR1 DR3 diodes R1 to R12 resistor

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) B23K 9/073 H02M 7/48 H02M 9/00

Claims (4)

(57) [Claims] 1. Two sets of forward converters, which are supplied with power from a DC power supply and have output terminals connected in parallel, and each set of forward converters are sequentially turned on and off.
The waveforms to be F-controlled are the same, but the phases are
An inverter-controlled welding power supply having a control circuit that operates with a shift of one-half cycle . 2. Two sets of capacitors for dividing an output voltage of a DC power supply, and two sets of forward converters having terminal voltages of the capacitors as inputs and output terminals connected in parallel,
ON / OFF control of the forward converter of each set in turn
The controlled waveforms are the same, but the phases are different from each other.
An inverter-controlled welding power source including a control circuit that operates at a half cycle shift . 3. The inverter-controlled welding power source according to claim 1, wherein the output terminals of the forward converters of each set are connected in parallel, and then one DC reactor is connected. 4. The inverter-controlled welding power source according to claim 1, wherein the forward converter is provided in two sets, a DC reactor is provided at an output terminal of each set, and a winding of each DC reactor has a common winding. Inverter-controlled welding power source composed of windings wound around an iron core and having a polarity in which the magnetic fields generated by the current flowing through each winding cancel each other.