JP3210173B2 - Power converter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a voltage type power converter for converting AC power supplied from a single-phase AC power supply into DC power, and in particular, absorbs power fluctuations of the single-phase AC power supply on the DC side. The present invention relates to a power conversion device that performs compensation control so as to perform the compensation.

[0002]

2. Description of the Related Art FIG. 12 shows a configuration diagram of a conventional voltage type power converter. In the figure, SUP is a single-phase AC power supply, Ls is an AC reactor, and CNV is a pulse width modulation control converter (hereinafter, referred to as
Cdo is a DC smoothing capacitor, and INV is a pulse width modulation control inverter (hereinafter referred to as a PWM converter).
IM) is a three-phase induction motor.

As a control circuit, a DC voltage detector ISO
, Input current detector CTS, motor current detector CTU to CTW
, DC voltage control circuit AVR, input current control circuit ACRS, converter pulse width modulation control circuit PWMC1, speed detector P
G, a speed control circuit SPC, a load current control circuit ACRL, and an inverter pulse width modulation control circuit PWMC2 are provided.

[0004] The PWM converter CNV has a voltage applied to the DC smoothing capacitor Cdo so that Vd is substantially constant.
The input current Is supplied from the AC power supply SUP is controlled.
At this time, by controlling the input current Is to a sine wave having the same phase as the power supply voltage Vs, power conversion with an input power factor = 1 and less harmonics can be performed.

On the other hand, the PWM inverter INV uses the DC smoothing capacitor Cdo as a DC power supply to convert it into three-phase AC power of a variable voltage and variable frequency, and outputs an AC motor (induction motor) IM
Drive. A system that receives power from such a single-phase AC power supply, converts the power into a DC voltage, and further converts the power into AC power to drive an electric motor is generally known.

[0006]

The conventional voltage-type power converter has the following problems. That is, when power is supplied from the single-phase AC power supply SUP, the power fluctuates at twice the power supply frequency, and as a result, the DC smoothing capacitor C
The voltage Vd applied to do also fluctuates at twice the frequency of the power supply. The magnitude of the voltage fluctuation is proportional to the magnitude of the active power output from the inverter, and is inversely proportional to the capacitance of the DC smoothing capacitor Cdo.

Therefore, if the capacity of the DC smoothing capacitor Cdo is increased, the fluctuation of the DC voltage Vd converted by the single-phase PWM converter CNV can be reduced, but the weight of the device increases and the cost increases. .
In particular, in a train drive system, it is desirable to reduce the weight of the device as much as possible, and a certain degree of DC voltage fluctuation must be allowed.

[0008] However, the fluctuation of the DC voltage Vd affects the inverter side and causes the output current of the inverter to fluctuate. For example, if the frequency of the single-phase AC power supply SUP is 50 Hz, the fluctuation frequency of the DC voltage Vd is 100 Hz. At this time, as the output frequency of the inverter approaches 100 Hz, the fluctuation of the output current increases, causing a beat phenomenon. As a result, the torque generated by the motor IM pulsates, causing vibration and noise.

Further, the maximum value of the voltage applied to the semiconductor elements constituting the inverter and the converter is increased by the fluctuation of the DC voltage Vd, and a semiconductor element having a higher withstand voltage must be used. This increases the size and size and increases the cost.

SUMMARY OF THE INVENTION The present invention has been made to eliminate the above-mentioned problem, and the power fluctuation of a single-phase AC power supply is absorbed and controlled by a DC active filter installed on the DC side of an AC / DC converter, so that the DC voltage can be reduced. It is an object of the present invention to provide a power converter capable of eliminating fluctuations and reducing the capacity of a DC smoothing capacitor.

[0011]

According to one aspect of the present invention, there is provided a single-phase AC power supply, and a main PWM converter for converting AC power of the single-phase AC power supply into DC power. A main smoothing capacitor connected between the DC terminals of the main PWM converter, a load device using the main smoothing capacitor as a DC power supply, and a frequency twice as high as the frequency of the single-phase AC power supply connected in parallel to the main smoothing capacitor. An LC series resonance circuit whose resonance frequency is adjusted near the frequency, and a DC active filter provided in a DC circuit of the main PWM converter, wherein the DC active filter includes a DC constant voltage source; A voltage-type PWM inverter for converting a voltage into a variable AC voltage;
A single-phase transformer connected to the AC output terminal of the inverter,
A DC capacitor connected between the DC terminals of the main PWM converter via a secondary winding of the single-phase transformer; and a main PWM that fluctuates due to twice the frequency of the single-phase AC power supply. A converter is provided with means for suppressing fluctuations in the DC voltage of the converter.

According to a second aspect of the present invention, in the DC active filter, a rectifier for converting alternating current supplied through a transformer to direct current, and converting a direct current voltage of the rectifier to a variable alternating current voltage. A voltage-type PWM inverter, a reactor and a DC capacitor connected in series between the AC terminals of the voltage-type PWM inverter via a DC circuit of the main PWM converter, and a frequency twice as high as the frequency of the single-phase AC power supply. And a means for suppressing the fluctuation of the DC voltage of the main PWM converter.

Further, according to a third aspect of the present invention, a DC active filter includes a DC constant current source, and a current source PWM for converting a DC current of the DC constant current source into a variable AC voltage.
An inverter, a high-frequency capacitor connected to an AC output terminal of the current source PWM inverter, and one end connected to the main PWM.
A DC capacitor connected to one end of a DC terminal of the M converter, and the other end connected to the other end of the DC terminal of the main PWM converter via an AC output circuit of the current source PWM inverter; The present invention is characterized in that it comprises means for suppressing fluctuation of the DC voltage of the main PWM converter which fluctuates due to twice the frequency.

According to a fourth aspect of the present invention, there is provided a DC active filter comprising: a DC constant voltage source; and a voltage source PWM for converting a DC voltage of the DC constant voltage source into a variable AC voltage.
An inverter, a single-phase transformer having a primary winding connected to an AC output terminal of the voltage-type PWM inverter, and a secondary winding connected in parallel to a reactor constituting the LC series resonance circuit; And a means for suppressing fluctuation of the DC voltage of the main PWM converter, which fluctuates due to a frequency twice as high as the frequency of the main PWM converter.

Further, the invention described in claim 5 is the first invention.
And means for calculating the instantaneous AC power Pc of the main PWM converter, and means for calculating average instantaneous power Pav supplied from a single-phase AC power supply. Means for calculating the compensation current command value _IF ^* from the AC instantaneous power Pc, the average active power Pav, and the voltage detection value Vd of the main smoothing capacitor, and the compensation current command value _IF ^*.
The LC and means for the current I _FO flowing through the series resonant circuit by subtracting obtain an output current command value I _A ^* of the DC active filter, the output current command value I _A ^* said DC active filter of the output current in response to the it is characterized in that is constituted by means for controlling the I _a.

[0016]

According to the first aspect of the present invention, the DC voltage of the main PWM converter, which fluctuates due to twice the frequency of the single-phase AC power supply, is reduced.
It is suppressed by the LC series resonance circuit connected between the DC terminals of the M converter, and the portion that cannot be completely suppressed by the LC series resonance circuit is suppressed by controlling the output current of the voltage-type PWM inverter constituting the DC active filter. It is like that.

Further, according to the present invention, the main PW which fluctuates due to twice the frequency of the single-phase AC power supply is provided.
The fluctuation of the DC voltage of the M converter is suppressed by the LC series resonance circuit connected between the DC terminals of the main PWM converter, and the part that cannot be completely suppressed by the LC series resonance circuit is
The output current of the voltage-type PWM inverter constituting the DC active filter is controlled by controlling the output current. The difference from the first embodiment is that the connection is made to the AC output terminal of the voltage-type PWM inverter. The single-phase transformer is omitted, and the DC voltage of the voltage-source PWM inverter is obtained by converting AC supplied from another transformer into DC by a rectifier.

Further, according to the present invention, the main PW which fluctuates due to twice the frequency of the single-phase AC power supply is provided.
The fluctuation of the DC voltage of the M converter is suppressed by the LC series resonance circuit connected between the DC terminals of the main PWM converter, and the part that cannot be completely suppressed by the LC series resonance circuit is
The output current of the voltage-type PWM inverter constituting the DC active filter is suppressed by controlling the voltage-type PWM inverter. However, the point different from the invention according to claim 1 is that the voltage-type PWM inverter constituting the DC active filter is provided. Instead of using a current source PWM inverter.

Further, according to the present invention, the main PW which fluctuates due to twice the frequency of the single-phase AC power supply is provided.
The fluctuation of the DC voltage of the M converter is suppressed by the LC series resonance circuit connected between the DC terminals of the main PWM converter, and the part that cannot be completely suppressed by the LC series resonance circuit is
Although the output current of the voltage type PWM inverter constituting the DC active filter is suppressed by controlling it, the difference from the invention of claim 1 is that the DC capacitor of the DC active filter is omitted, That is, the capacitor of the LC series resonance circuit also has the function.

Further, according to the present invention, the main PW which fluctuates due to twice the frequency of the single-phase AC power supply is provided.
Preferable examples for suppressing the variation of M converter of the DC voltage, the command value I _A ^* of the output current I _A of DC active filter is supplied from single-phase AC power from the AC side instantaneous power Pc of the main PWM converter The average value Pav of the active power is subtracted to obtain the fluctuation power ΔPc = Pc−Pav,
Compensation current command value of the total combined with LC series resonance circuit by dividing the DC voltage Vd I _F ^* = calculated the Delta] Pc / Vd, the resonance current I flowing from the compensation current command value I _F ^* to the LC series resonant circuit command value sought I _a ^* of the current I _a DC active filter supplies by subtracting _FO,
By flowing the DC active filter current I _A = I _A ^* , the DC voltage fluctuation due to the fluctuation of the single-phase power of the main smoothing capacitor can be suppressed with high accuracy. Therefore, the main smoothing capacitor is a PWM converter and a three-phase PWM
It suffices if there is a capacity enough to absorb the harmonic components on the DC side of the M inverter, and a very large one is not required.

[0021]

FIG. 1 is a block diagram of a main circuit and a control circuit showing an embodiment of a power converter according to the present invention. In the figure, SUP is a single-phase AC power supply, Ls is an AC reactor, CNV is a main PWM converter, and INV is a main PWM.
Inverter, IM is three-phase induction motor, Cd is main smoothing Kon'ndensa, L _F, C _F is a reactor and a capacitor, DC-AF of the LC series resonance circuit is a DC active filter.

[0022] DC active filter DC-AF, a current detector CTA, a DC voltage source E _A, single-phase voltage source PWM inverter VSI, a single-phase transformer TR, is composed of a reactor L _A and capacitor C _A.

As a control device, an input current detector CTS
, DC voltage detector PTD, current detector CTF, load current detector CTL, speed detector PG, DC voltage control circuit AVR, input current control circuit ACRS, converter PWM control circuit PWMC1
, Speed control circuit SPC, load current control circuit ACRL, three-phase inverter PWM control circuit PWMC2, compensation current calculation circuit CA
L, a compensation current control circuit ACRA, and a PWM control circuit PWMC3 for a single-phase voltage type PWM inverter VSI.

The main PWM inverter INV uses the main smoothing capacitor Cd as a DC power supply, converts DC to AC,
The three-phase AC power having a variable voltage and a variable frequency is supplied to the induction motor IM.

That is, the rotational speed ωr of the electric motor is used as a speed detector.
Detected by PG and input to speed control circuit SPC. Speed control circuit SPC compares a speed command .omega.r ^* and speed detection value .omega.r, making three-phase load current command value I _L ^* corresponding to the deviation εr = ωr ^* -ωr.

The load current control circuit ACRL is a three-phase load current IL (I _U , I _V , I _W ) detected by the current detector CTL.
And three-phase load current command values _IL ^* ( _IU ^* , _IV ^* , _IW
^* ) And the three-phase voltage command value e _L according to the deviation.
^* (E _U ^* , e _V ^* , e _W ^* ) are converted to PWM control circuit PWMC2
Give to. As a result, three-phase PWM inverter 3-phase voltage command value e _L ^* 3-phase voltage proportional to _{V L (V U, V V} , V
_W) generates and controls the three-phase load current I _L.

It is known that the output characteristics of a DC motor can be obtained by performing vector control of the induction motor IM, but this is not the main point of the present invention, and therefore the description thereof is omitted here.

When viewed from the DC voltage source (main smoothing capacitor Cd), the three-phase output main PWM inverter INV and the induction motor IM are a kind of constant current that takes the DC current Id2 of FIG. Can be considered a source.

The main PWM converter CNV controls the input current Is so that the voltage Vd applied to the main smoothing capacitor Cd becomes substantially constant. At this time, the input current Is is changed to the power supply voltage V
By controlling the sine wave in the same phase (or opposite phase) as s, the operation with the input power factor = 1 can be performed.

That is, the voltage Vd of the main smoothing capacitor Cd is detected by the DC voltage detector PTD, input to the voltage control circuit AVR, compared with the voltage command value Vd ^*, and the deviation εv = V
The peak value command Ism ^* of the input current is obtained by amplifying d ^* -Vd. Further, the voltage Vs of the single-phase AC power supply SUP is detected, a unit sine wave sinωt synchronized with the voltage Vs is obtained, and multiplied by a current peak value command Ism ^* to obtain an input current command value Is ^* . That is,

[0031]

## EQU1 ## Is ^* = Ism ^* .sin ωt. The input current control circuit ACRS compares the input current Is detected by the current detector CTS with the current command value Is ^* , amplifies the deviation ε1 = Is ^* −Is (by a factor of −k1), and increases the PWM of the main converter CNV. The input signal ec ^* is set to the control circuit PWMC1. The main PWM converter CNV receives the input signal e
A voltage Vc proportional to c ^* is generated on the AC side, and the input current I
Control s. The power supply voltage Vs is connected to the AC reactor Ls.
Difference voltage V _LS = Vs -Vc of the converter voltage Vc is applied with.

For example, when Is ^* > Is, the deviation ε1 has a positive value, and the input signal ec ^* of the PWM control circuit has a negative value. Thus, the voltage V _LS to be applied to AC inductor Ls is increased, increasing the input current Is. Conversely, Is ^* <
In the case of Is, the deviation .epsilon.1 has a negative value, and the input signal ec ^* of the PWM control circuit has a positive value. Therefore, the voltage V _LS applied to the AC reactor Ls decreases, and the input current Is decreases. Therefore, the input current Is is controlled to match the current command value Is ^* . In this case, the current command value Is ^*
Is given by a sine wave in phase with the power supply voltage Vs, and the input current I
s is also controlled accordingly, achieving operation with less input harmonics at input power factor = 1.

The DC voltage Vd is controlled as follows. For example, when Vd ^* > Vd, the deviation εv becomes a positive value, and the current peak value command Ism ^* is increased by a positive value. As a result, power Ps = Vs supplied from the end-phase AC power supply SUP
・ Is becomes a positive value and the power Ps becomes the main smoothing capacitor C
d to increase the DC voltage Vd. Conversely, Vd
^{* If} <Vd, the deviation εv is a negative value, and the current peak value command Ism ^* is a negative value. As a result, the energy stored in the main smoothing capacitor Cd is regenerated to the single-phase AC power supply SUP, and the DC voltage Vd is reduced. In this way, the DC voltage Vd is controlled to match the command value Vd ^* .

FIG. 2 shows the voltage on the AC power supply side of the apparatus shown in FIG.
FIG. 4 shows an example of a current vector diagram. In the figure, (a) shows a vector diagram during the power running operation, in which the input current Is is controlled to be in phase with the power supply voltage Vs. When the input current Is flows, the voltage V _LS = jω · L is applied to the AC reactor Ls.
s · Is is applied, and the generated voltage Vc of the PWM converter CNV is as shown in the figure. The magnitude V of the voltage Vc at this time
The cm and the phase angle θ (delay) are represented by the following equations.

[0035]

(Equation 2) _{^{θ = tan -1 (V LSm /}} Vsm) , however, and Vs = Vsm · sinωt Is = Ism · sinωt V LSm = ω · Ls · Ism.

FIG. 4B is a vector diagram of the regenerative operation, in which the input current Is is controlled in the opposite phase to the power supply voltage Vs. Voltage V applied to AC reactor Ls
_Since the phase of _LS is inverted, the phase angle θ of the generated voltage Vc of the PWM converter CNV is advanced.

FIG. 3 shows the main PWM converter CN of the device of FIG.
5 shows a voltage-current waveform at the time of power running operation of V. The input current Is leads the generated voltage Vc of the main PWM converter CNV by the phase angle θ. The instantaneous power Ps supplied from the single-phase AC power supply SUP is

[0038]

## EQU3 ## Ps = Vs.Is = Vsm.sin.omega.t.times.Ism.sin.omega.t = (Vsm.Ism / 2). (1-cos2.omega.t), and fluctuates at twice the frequency of the single-phase AC power supply SUP. Also, the instantaneous power Pc of the main PWM converter CNV
Is the product of the converter generated voltage vc and is, and the fluctuation is increased by the voltage drop of the AC reactor Ls. That is,

[0039]

Pc = Vc · Is = Vcm · sin (ωt−θ) × Ism · sinωt = (Vcm · Ism / 2) · ｛cosθ−cos (2ωt−
θ)｝. Since Vcm = Vsm / cos θ holds, the average value of the active power is Pav = Vsm · Ism / 2. In the steady state, this active power Pav is equal to the load (main PW
Match the power P _L of M inverters + induction motor) is consumed by the power fluctuation ΔPc is out of the main smoothing Kon'ndensa Cd. That is,

[0040]

Equation 5] _{ΔPc = Pc -P L = - (} Vcm · Ism / 2) · cos (2ωt-θ) , and the average value of the current idc is a DC voltage through the main smoothing capacitor Cd as Vdo, as follows Can be approximated.

[0041]

Equation 6 idc = ΔPc / Vdo = − (Vcm · Ism / 2Vdo) · cos (2ωt−θ) = − {Pav / (Vdo · cosθ)} · cos (2ωt−
θ) Therefore, the variation ΔVd of the DC voltage is given by the following equation.

[0042]

(Equation 7) = − {Vcm · Ism / (4 · Vdo · ω · Cd)} · sin
(2ωt−θ) = − {Pav / (2 · cos θ · Vdo · ω · Cd)} · si
n (2ωt−θ) = − ΔVdm · sin (2ωt−θ) That is, the magnitude of the DC voltage fluctuation ΔVd is proportional to the active power P _L = Pav = Vsm · Ism / 2 taken by the load, and the value of the main smoothing capacitor Cd is It is inversely proportional to capacity.

For example, P _L = 3,000 [kw], Vdo
= 2,000 [v], f = ω / (2π) = 50 [Hz],
When Cd = 0.02 [F] and cos θ = 0.9,
The magnitude of ΔVd is ΔVdm = 132.6 [v].
FIG. 4 shows the voltage and current waveforms of the respective parts during the regenerative operation.
Therefore,

[0044]

Pc = Vc · Is = Vcm · sin (ωt + θ) × Ism · sin (ωt +
π) = − (Vcm · Ism / 2) · ｛cosθ−cos (2ωt
+ Θ)｝. Since Vcm = Vsm / cos θ holds, the average value Pav of the active power is Pav = Vsm · Ism / 2. The active power Pav is at steady state coincide with the power P _L that is regenerated from the load (the main PWM inverter + induction motor), only the power fluctuation ΔPc is out of the main smoothing capacitor Cd. That is,

[0045]

Equation 9] _{ΔPc = Pc -P L = (Vcm} · Ism / 2) · cos (2ωt + θ) , and the current flowing in main smoothing capacitor Cd idc as Vdo the average value of the DC voltage can be approximated by the following equation .

[0046]

## EQU10 ## idc = ΔPc / Vdo = (Vcm · Ism / 2Vdo) · cos (2ωt + θ) = {Pav / (Vdo · cosθ)} · cos (2ωt +
θ) Therefore, the variation ΔVd of the DC voltage is given by the following equation.

[0047]

[Equation 11] = {Vcm · Ism / (4 · Vdo · ω · Cd)} · sin (2
ωt + θ) = {Pav / (2 · cos θ · Vdo · ω · Cd)} · sin
(2ωt + θ) = ΔVdm · sin (2ωt + θ).

Next, the control operation of the DC active filter DC-AF of the apparatus shown in FIG. 1 will be described. First, the compensation current calculator
CAL by calculating a command value I _A ^* of the current I _A supplied from DC active filter DC-AF, providing a current control circuit ACRA.

Meanwhile, detects compensation current I _A flowing into DC active filter DC-AF by current detector CTA, is input to the current control circuit ACRA. The current control circuit ACRA, the compensation current command value I _A ^* with compensation current detection value I is compared with _A, the deviation ε _A = I _A ^* -I _A and inversely amplifies the voltage command value to the PWM control circuit PWMC3 e Give _A ^* . Voltage source PWM inverter VSI generates a voltage V _A which is proportional to the voltage command value e _A ^*, to control the compensation current I _A.

[0050] That is, when it becomes the I _{A ^*>} I A, the deviation epsilon _A is a positive value and rounding the voltage command value e _A ^* is a negative value the output voltage V _A becomes a negative value. As a result, increased compensation current I _A is controlled such that I _A = I _A ^*.

Conversely, if I _A ^* <I _A , the deviation ε
_A is ^* a negative value and becomes the voltage command value e _A is the output voltage V _A becomes a positive value to a positive value. As a result, compensation current I _A decreases again, it is controlled such that I _A = I _A ^*.

FIG. 5 shows an equivalent circuit of the apparatus shown in FIG. 1, in which a main PWM converter CNV and a main PWM inverter IN
V can be represented as a current source. That is, PWM
When ignoring the harmonic current accompanying the control, the main PWM
The input current Id2 of the inverter INV includes only the direct current component Ido, and the output current Id1 of the main PWM converter CNV includes the direct current component Ido and the alternating current component idc that changes at twice the frequency of the power supply frequency.
It is included. Also, the resonance frequency f of the LC series resonance circuit
R is set close to twice the power supply frequency fs, and the resonance current _IFO flows. Here, DC active filter DC-AF
There by taking the compensation current I _A = idc-I _FO, current Id3 flowing in main smoothing capacitor Cd becomes zero. Precisely because the compensation current I _A does not contain harmonic current,
Harmonic components flow into the main smoothing capacitor Cd. However, since the frequency is high, the main smoothing capacitor Cd
The capacitance of each stage becomes smaller and voltage fluctuations are almost eliminated.

FIG. 6 shows the DC active filter DC-A of FIG.
4A is a configuration diagram showing a specific embodiment of F. FIG. 4A is a configuration diagram of a main circuit, and FIG. 4B is a configuration diagram of a compensation current calculation circuit CAL that suppresses fluctuations in the DC voltage of the main PWM converter CNV.

[0054] In FIG. 6 (a), P, N are DC positive and negative terminals of the main circuit, the E _A DC voltage source, VSI is single-phase voltage source PWM inverter, TR is single-phase transformer, L _F Is a reactor, and _CF is a DC smoothing capacitor. Single-phase voltage type P
The WM inverter VSI includes switching elements S1 to S4 and freewheeling diodes D1 to D4. The reactor LA is connected to the primary side of the single-phase transformer TR. In addition, for convenience of explanation, the primary /
The next turn ratio is 1: 1.

As shown in FIG. 1, the control device includes a current detector CTA, a compensation current calculation circuit CAL, and a compensation current control circuit A.
CRA, PWM control circuit PWM of single phase voltage type PWM inverter
Equipped with C3.

As shown in FIG. 6B, the compensation current calculation circuit CAL includes multipliers ML1 and ML2, a proportional calculator OA, and an adder AD1,
It consists of an AD2 divider DIV. The compensation current control circuit ACRF includes a comparator C1 and a control compensation circuit G _A (s).

First, the multiplier ML2 finds the product of the AC side voltage Vc of the main PWM converter CNV and the input current Is. At this time, since the detected voltage value Vc contains many harmonic components, the PWM control input signal (voltage command value) ec ^* of the converter CNV may be used instead. Similarly, the current command value Is ^* may be used instead of the input current detection value Is. The output of the multiplier ML2 is the instantaneous power P of the converter CNV.
becomes c.

The power supply voltage peak value V is calculated by the multiplier ML1.
Calculates the product of sm and the input current peak value Ism, and calculates a proportional calculator OA
And multiply by (1/2). This result is the average value Pav of the active power supplied from the power supply. Note that the command value Ism ^{* is used} instead of Ism ^.
May be used.

Pc−Pav is calculated by the adder AD1,
The fluctuating power ΔPc is obtained and input to the divider DIV. Divider D
The power fluctuation ΔPc in IV is divided by the DC voltage detection value Vd and the results, and the compensation current command value of the entire including current I _FO flowing through the LC series resonant circuit I _F ^*. The overall compensation current command value I _F ^* is equal to fluctuation idc of the converter CNV of the DC side current based on the power fluctuation ΔPc of the single-phase AC power source SUP.

On the other hand, the current I _FO flowing in the LC series resonance circuit
Was detected by a current detector CTF, taking the difference between the adder compensation current command value of the entire input to AD2 I _F ^*, the command value I _A of the current I _A supplied from DC active filter DC-AF
^{Ask *} .

[0061] In other words, the _{^{I A * = I F * -I}} FO. Is said current command value I _A ^* is input to comparator C1 of the next current control circuit ACRF, is compared with compensation current I _A which is detected by the current detector CTA. The deviation ε _A = I _A ^* −I _A is the control compensation circuit G
_A (s) is input and inverted and amplified (−k _A times) to obtain a voltage command value e _A ^{* of} the single-phase voltage-type PWM inverter VSI. Since the control of the compensation current so that I _A = I _A ^* has been described above, the description thereof is omitted.

As described above, the DC active filter DC-A
F takes the compensation current ^{_{I A = I A * = idc}} -I FO, by LC series resonant circuit takes the I _FO, prevents variation in the single-phase power flows through the main smoothing capacitor Cd, without a voltage variation can do.

[0063] In the above description, determined by calculating the command value I _A ^* of compensation current I _A of DC active filter DC-AF, an example has been described for controlling the DC active filter DC-AF, for example, a DC circuit The DC component can be removed from the DC current to directly detect the current compensated by the DC active filter DC-AF, and control can be performed so as to follow this detected current. Occurs.

FIG. 7 is a block diagram showing an embodiment of a DC active filter used in the power converter according to the second aspect of the present invention. The LC series resonance circuit (L _F , C _F ) and the main smoothing capacitor Cd are described for convenience.

In the figure, P and N are DC positive and negative terminals of the main circuit, AC-SUP is an AC power supply, AC-TR is a transformer, REC is a rectifier circuit, and _CE is the DC of the active filter DC-AF. Power supply smoothing capacitor, VSI is single phase voltage type PWM inverter, L
_A is a reactor, and C _A is a DC capacitor. The single-phase voltage type PWM inverter VSI includes switching elements S1 to
S4 and free-wheeling diodes D1 to D4.

The control device includes a current detector CTA, a compensation current calculation circuit CAL, a limiter circuit LIM, a comparator C1, a current control compensation circuit G _A (s), and a PWM control circuit PWMC3.

The compensation current calculation circuit CAL has the same configuration as that described with reference to FIG. Figure 6 differs from the DC voltage source E _A insulated by AC transformer AC-TR, is that omitting the output transformer of the inverter VSI. Power supplied from DC active filter DC-AF is basically the reactive power may be supplied only loss from the DC voltage source E _A. Therefore, the capacity of the AC transformer AC-TR and the rectifier circuit REC is small, and there is an advantage that the size and weight can be reduced as compared with the apparatus of FIG. 6 in which the transformer TR is provided on the output side of the inverter VSI.

[0068] limiter circuit LIM of the control circuit is inserted to limit the maximum value of the supplied current I _A from DC active filter DC-AF. FIG. 8 is a block diagram showing an embodiment of a DC active filter used in the power converter according to the third aspect of the present invention. LC series resonance circuit (L _F ,
C _F ) and the main smoothing capacitor Cd are shown for convenience.

In the figure, P and N are DC positive and negative terminals of the main circuit, CHO is a switching element for DC chopper, D _CH is a free-wheeling diode for chopper, Lo is a DC reactor, and CSI is a single-phase current type. PWM inverter, C _H is high frequency capacitor, TR is transformer, the C _a is the DC capacitor. The single-phase current-source PWM inverter CSI includes switching elements S11 to S14.

Further, the control device includes current detectors CTO and CTA.
, CTF, comparator C2, current control compensation circuit Ho (s), compensation current calculation circuit CAL, divider DIV, PWM control circuit PWMC4,
Equipped with PWMC5.

First, the operation of the DC chopper will be described. The DC current Io is detected by the current detector CTO, input to the comparator C2, and compared with the current command value Io ^* . The deviation εo =
Io ^* -Io is amplified by the current control compensation Ho (s) to produce a voltage command value eo ^*, which is input to the PWM control circuit PWMC5. The PWM control circuit PWMC5 supplies a gate signal to the switching element CHO such that the average value of the voltage applied to the DC reactor Lo is proportional to the voltage command value eo ^* .

That is, when the voltage command value eo ^* is small,
The ratio of the on period t _ON for the switching period T of CHO reduced, and increase the ratio of the ON period t _ON of the device CHO accordance eo ^* becomes larger.

When Io ^* > Io, the deviation ε becomes a positive value, the voltage command value eo ^* increases, the ratio of the _ON period t _ON of the element CHO increases, and the DC current Io increases. Conversely, if Io ^* <Io, the deviation ε
Becomes a negative value, the voltage command value eo ^* decreases, and the element CHO
, The ratio of the ON period t _ON decreases, and the DC current Io decreases. In this way, the DC current Io is
o Controlled to match ^* .

Next, the operation of the DC active filter DC-AF will be described. The compensation current calculation circuit CAL of FIG.
As described in (b), the power supply voltage peak value Vsm, the input current peak value command Ism ^* , the voltage command ec of the PWM converter CNV ^{*, the} input current command Is ^* , and the DC voltage detection value Vd
When obtains the command value I _A of the compensating currents is DC active filter DC-AF by calculating the like current detection value I _FO flowing through the LC series resonant circuit ^*. A compensation current command value I _A ^* is input to divider DIV, divided by the DC current detection value Io (or DC current command value Io ^*), P of current source inverter CSI
It is assumed that the input signal k _A ^* for the WM control.

FIG. 9 shows the current source PWM inverter CSI of FIG.
3 is a time chart for explaining the PWM control operation of FIG. In the figure, X and Y are carrier signals of PWM control, and +
A triangular wave that changes between 1 and −1 is often used. The phase of the triangular wave Y (broken line) is shifted by 180 ° from the phase of the triangular wave X (solid line).

The triangular wave X is compared with the modulation factor k _A ^* to generate a gate signal g1 for the elements S11 and S12. The triangular wave Y is compared with the modulation factor k _A ^* to obtain the gate signal g for the elements S13 and S14.
Make 2 That is, when the k _A ^*> X, with g1 = 1, S11: On (S12: OFF) when k _A ^* ≦ X, with g1 = 0, S11: Off (S12: ON) of k _A ^*> Y when, in g2 = 1, S14: on (S13: oFF) when k _a ^* ≦ Y, with g2 = 0, S14: a: (oN S13) off. In the current source inverter CSI, a lap period is provided so that the ON periods of the elements S11 and S12 slightly overlap to secure a path for the current Io to flow. Similarly, a lap period is provided between the elements S13 and S14.

Output current (compensation current) I of inverter CSI
_A is as follows depending on whether the elements S11 to S14 are on or off. When S11 and S14 is turned on, when I _A = + Io S11 and S13 is turned on, when I _A = + 0 S12 and S14 is turned on, when I _A = + 0 S12 and S13 is ON, I _A = -Io view at the bottom of the 9 shows the waveform of the output current I _a. The average value I _{A (av)} is equal to k _A ^* · Io, which coincides with compensation current command value I _A ^*. In FIG. 8, the high-frequency capacitor C
_H is provided to absorb the harmonic components of compensation current I _A. Thus, the compensation current I _A = I _A ^* is supplied from the current source PWM inverter CSI.

The compensation current I _A = I _A ^* is added to the current flowing through the LC series resonance circuit (L _F , C _F ), and
Cancels the fluctuating current idc on the DC side of the WM converter CNV. As a result, the fluctuation current idc does not flow through the main smoothing capacitor Cd shown in FIG. 1, and the fluctuation of the DC voltage Vd can be eliminated.

[0079] According to the present invention apparatus using a DC active filter of FIG. 8, there is an advantage that it is possible to control the compensation current I _A directly. FIG. 10 is a block diagram showing another embodiment of the DC active filter used in the power converter according to the third aspect of the present invention. LC series resonance circuit (L _F ,
C _F ) and the main smoothing capacitor Cd are shown for convenience.

In the figure, P and N are DC positive and negative terminals of the main circuit, Eo is a DC voltage source, CHO is a switching element of a DC chopper circuit, D _CH is a wheeling diode, Lo is a DC reactor, and CSI is a DC reactor. Single phase current type PWM inverter,
C _H is a high-frequency capacitor and C _A is a DC capacitor. The single-phase current-source PWM inverter CSI includes switching elements S11 to S14.

The difference from FIG. 8 is that the output transformer of the single-phase current source PWM inverter CSI is omitted. Since the capacity of the DC voltage source Eo is small and the output transformer is omitted, there is an advantage that the size and weight of the DC active filter DC-AF can be reduced. Other operations are the same as in FIG.

FIG. 11 is a block diagram showing an embodiment of a DC active filter used in the power converter according to the fourth aspect of the present invention. In the figure, P, N are DC positive and negative terminals of the main circuit, E _A DC Rabbit voltage source, VSI is single-phase voltage source PWM
Inverter, TR is single-phase transformer, L _F is the reactor, C _F
Is a DC smoothing capacitor. The single-phase voltage-type PWM inverter VSI is composed of switching elements S1 to S4 and wheeling diodes D1 to D4. Also, the reactor L _F is connected to the primary side of the single-phase transformer TR.
For convenience of explanation, the primary / secondary turns ratio of the single-phase transformer TR is 1: 1.

As a control device, a current detector CTA,
A compensation current calculation circuit CAL, a compensation current control circuit ACRF, and a PWM control circuit PWMC3 for a single-phase voltage-type PWM inverter are provided. The compensation current calculation circuit CAL is the same as that described with reference to FIG.

[0084] In the embodiment of FIG. 11, DC active filter DC-AF of the DC capacitor C _A is omitted, also serves as the effect by the capacitor C _F of the LC series resonant circuit. That is, the capacitor C _F flows the sum of the compensation current I _A from the DC active filter DC-AF and the resonance current I _FO,
The control is performed so as to cancel the fluctuation current idc based on the power fluctuation of the single-phase power supply. As a result, the DC capacitor C _A can be omitted, and the device can be reduced in size and weight and cost can be reduced.

As described above, according to the power converter of the present invention, the power fluctuation based on the single-phase power supply is absorbed by the LC series resonance circuit (L _F , C _F ) and the DC active filter DC-AF, It is possible to eliminate the fluctuation of the DC voltage Vd. Only the harmonic current based on the PWM control flows through the main smoothing capacitor Cd, and the capacity of the main smoothing capacitor Cd can be greatly reduced.

Further, in the device of the present invention, since most of the compensation current based on the variation of the single-phase power is supplied from the LC series resonance circuit, there is an advantage that the capacity of the DC active filter DC-AF can be extremely small. is there.

The capacity of the PWM control converter (VSI or CSI) used in the DC active filter DC-AF is as follows, assuming that the primary / secondary turns ratio of the output transformer is 1: 1. .

[0088] That is, in order to cancel the power fluctuation of single-phase power supply, it is necessary to flow the entire compensation current I _F = idc, to flow the resonance current I _FO to LC series resonance circuit, DC active filter DC- from the _{AF, I a = idc-I} FO
Just let it flow. This value is the current capacity of the PWM control converter (VSI or CSI). Current I _A = I _Am · sin
When (2ωt) flows into the DC capacitor C _A , a voltage V _CA represented by the following equation is applied to the capacitor C _A.

[0089]

V _CA = V d + I _Am / (2ωC _A ) · cos (2ωt) As a result, in the DC active filter DC-AF, _VA = V _CA −V d = I _Am / (2ωC _A ) · cos ( 2ω
t) is applied, and the voltage capacity of the PWM control converter (VSI or CSI) needs to be V _Am = I _Am / (2ωC _A ).

For example, load P _L = 3,000 [kw], DC voltage Vd = 2,000 [v], power supply frequency f = ω / 2π = 50 [Hz], c
When operated at osθ = 0.9, the peak value of the entire compensation current is I _Fm = V _Cm · I _Sm / (2 · Vdo) = 1,666 [A]. Assuming that 90% of the current flows through the LC series resonance circuit, the maximum value of the current of the DC active filter DC-AF is I _Am = 16
6.6 [A]. When C _A = 1000 [μF], the voltage capacity is V _Am = 265 [v]. Therefore, the capacity of the PWM control converter (VSI or CSI) is

[0091]

[Expression 13] I _Am · V _Am /2=166.6 [A] · 265 [v] / 2 = 22 [kVA] This is 0.7 [%] for 3,000 [KVA] main transformer.
It means that the capacity is enough.

As described above, by providing the DC active filter DC-AF having a small capacity, DC voltage fluctuation can be eliminated, and a stable constant voltage can be supplied to a load device using the DC active filter as a voltage source. Become like

[0093]

As described above, according to the first aspect of the present invention, the fluctuation of the DC voltage based on the fluctuation of the power of the single-phase AC power supply can be eliminated, and the utilization rate of the power converter is correspondingly reduced. Is improved. Further, the beat phenomenon of the inverter output current, which has been a problem in the induction motor driven by the PWM inverter, is eliminated, and the vibration and noise of the motor can be greatly reduced.
Further, the capacity of the DC smoothing capacitor can be significantly reduced, and the shape and weight of the entire power converter can be reduced.

According to the second aspect of the present invention, in addition to the above-described effects, the input side of the DC active filter is insulated by the transformer, and the transformer on the output side is omitted. There is an advantage that the size and weight can be reduced as compared with the invention described in claim 1 in which a transformer is provided on the output side.

Further, according to the third aspect of the present invention, the converter for converting direct current into alternating current constituting the direct current active filter is constituted by a current type PWM converter to achieve the above-mentioned effect. .

According to the fourth aspect of the present invention, in addition to the above-described effects, the DC capacitor of the DC active filter is omitted and the function of the LC series resonance circuit is also used. Small size, light weight, and cost reduction can be achieved. Furthermore, according to the fifth aspect of the invention, it is possible to suppress the DC voltage fluctuation based on the power fluctuation of the single-phase AC power supply by achieving high-precision control.

[Brief description of the drawings]

FIG. 1 is a configuration diagram showing one embodiment of a power conversion device according to the present invention.

FIG. 2 is a voltage and current vector diagram for explaining the operation of the power converter of FIG. 1;

FIG. 3 is a waveform chart of voltages, currents, and powers of respective parts for describing an operation of the power converter of FIG. 1 during a power running operation.

FIG. 4 is a waveform diagram of voltages, currents, and powers of respective parts for describing an operation of the power converter of FIG. 1 during a regenerative operation.

FIG. 5 is an equivalent circuit diagram for explaining the operation of the power converter of FIG. 1;

FIG. 6 is a configuration diagram showing a specific example of the DC active filter in FIG. 1; FIG. 6A is a configuration diagram on a main circuit side;
(B) is a configuration diagram on the control circuit side.

FIG. 7 is a configuration diagram showing an embodiment of a DC active filter used in the power converter according to the invention of claim 2;

FIG. 8 is a configuration diagram showing one embodiment of a DC active filter used in the power converter according to the third aspect of the present invention.

FIG. 9 is a time chart for explaining a PWM control operation of the DC active filter in FIG. 8;

FIG. 10 is a configuration diagram showing another embodiment of the DC active filter used in the power converter according to the third aspect of the present invention.

FIG. 11 is a configuration diagram showing an embodiment of a DC active filter used in the power converter according to the invention of claim 4.

FIG. 12 is a configuration diagram of a conventional power converter.

[Explanation of symbols]

SUP… Single-phase AC power supply L _S
... AC reactor CNV ... Main PWM converter Cd
… Main smoothing capacitor L _F … Inductance C _F
… Capacitor DC-AF… DC active filter INV
… Main PWM inverter IM… Induction motor

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H02M 7/155 H02M 1/14 H02M 7/48

Claims (2)

(57) [Claims] 1. A single-phase AC power supply; a main PWM converter for converting AC power of the single-phase AC power supply to DC power;
A main smoothing capacitor connected between the DC terminals of the WM converter, a load device using the main smoothing capacitor as a DC power supply, and a power supply connected in parallel to the main smoothing capacitor and having a frequency around twice the frequency of the single-phase AC power supply; An LC series resonance circuit having a matched resonance frequency, and a DC active filter provided in a DC circuit of the main PWM converter, wherein the DC active filter varies a DC constant voltage source and a DC voltage of the DC constant voltage source. A voltage-type PWM inverter for converting a voltage into an AC voltage, a single-phase transformer connected to an AC output terminal of the voltage-type PWM inverter, and two of the single-phase transformer.
A DC capacitor connected between the DC terminals of the main PWM converter via a secondary winding; and a compensation current for compensating a difference between an AC component included in an output current of the main PWM converter and a resonance current flowing in the LC series resonance circuit. Means for controlling the voltage-type PWM inverter as a command value to suppress fluctuations in the DC voltage of the main PWM converter that fluctuate due to twice the frequency of the single-phase AC power supply. Power converter. 2. A single-phase AC power supply; a main PWM converter for converting AC power of the single-phase AC power supply to DC power;
A main smoothing capacitor connected between the DC terminals of the WM converter, a load device using the main smoothing capacitor as a DC power supply, and a power supply connected in parallel to the main smoothing capacitor and having a frequency around twice the frequency of the single-phase AC power supply; An LC series resonance circuit having a matched resonance frequency, and a DC active filter provided in a DC circuit of the main PWM converter, wherein the DC active filter includes a rectifier that converts an AC supplied through a transformer into a DC. A voltage-type PWM inverter for converting the DC voltage of the rectifier into a variable voltage AC voltage, and a reactor connected between one end of an AC terminal of the voltage-type PWM inverter and one end of a DC terminal of the main PWM converter. A direct connection between the other end of the AC terminal of the voltage type PWM inverter and the other end of the DC terminal of the main PWM converter. Controlling the voltage-type PWM inverter by using a capacitor and a difference between an AC component included in an output current of the main PWM converter and a resonance current flowing through the LC series resonance circuit as a compensation current command value to thereby control the single-phase AC power supply. A power converter, comprising: means for suppressing fluctuation of a DC voltage of the main PWM converter, which fluctuates due to a frequency twice as high as the frequency.