JP2889303B2 - Inverter device - Google Patents

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter device for converting a commercial AC power supply into a DC voltage, switching the DC voltage by turning on / off a switching element, and supplying the switching to a load. It is.

[Prior Art] Conventional Example 1 FIG. 7 shows a conventional general inverter device (Japanese Patent Application Laid-Open No. 60-1985).
FIG. 134 is a circuit diagram. AC voltage of the AC power source Vs is totally wave rectified by the diode bridge circuit via an AC filter F consisting of the diode D ₃ to D _6, is smoothed by the capacitor C ₀ for smoothing, a DC voltage. A series circuit of transistors Q ₁ and Q ₂ is connected in parallel to both ends of the capacitor C ₀ . Diodes D ₁ and D ₂ are connected in anti-parallel to the transistors Q ₁ and Q ₂ , respectively. Transistor Q
The _first ends, the discharge lamp la is connected through a capacitor C ₁ and the inductor L _2. The non-power side of the discharge lamp la, the capacitor C ₃ is connected in parallel. Transistors Q ₁ and Q ₂
Are driven to alternately turn on and off at a high speed. First, when the transistor Q ₁ is a transistor Q ₂ is turned off in the on state, the transistor Q ₁ from the capacitor C ₁
A current flows in one direction to the discharge lamp la through the discharge lamp la. Then, when the transistor Q ₁ is the transistor Q ₂ is turned on in the OFF state, the capacitor C ₁ from the capacitor C _0, the discharge lamp l
a, inductor L _2, a current flows in the reverse direction through the transistor Q _2. Therefore, high-frequency power is supplied to the discharge lamp la. Thus, a half-bridge type inverter circuit is configured.

In this circuit, the transistor Q ₂ also serves as a switching device of the chopper circuit. First, the transistor
When Q ₂ is turned on, thereby shorting the DC output ends of the diode bridge in the inductor L _1. Thus, the current flowing through the inductor L ₁ is increased in inclination in proportion to the magnitude of the DC output voltage of the diode bridge, an inductor L ₁
Energy is stored. Next, the transistor Q ₂
There Once off, the energy of the inductor L ₁ is released via the diode D ₁ to charge the capacitor C _0. At this time, the capacitor C _0, the voltage obtained by adding the voltage generated across the inductor L ₁ into a DC output voltage of the diode bridge is charged, a DC voltage higher than the peak value of the AC power source Vs to capacitor C ₀ Can be obtained. Also,
Compared to the conventional example 1, since a long period in which the charging current flows into the capacitor C _0, the voltage of the capacitor C ₀ is smoother.

Conventional Example 2 FIG. 8 shows another conventional inverter device (Japanese Patent Application No. 1-6446).
FIG. 5 is a circuit diagram. Hereinafter, the circuit configuration will be described. The transistors Q ₁ and Q ₂ are bipolar transistors. The emitter of the transistor Q ₁ is connected to the collector of the transistor Q _2. Transistor
The cathodes and anodes of diodes D ₁ and D ₂ are connected to the collectors and emitters of Q ₁ and Q ₂ , respectively. The collector of the transistor Q ₁ is connected the cathode of a diode D _3, the anode of the diode D ₃ is connected to the cathode of the diode D _4, the anode of the diode D ₄ is the transistor Q ₂
Connected to the emitter. The collector of the transistor Q ₁ is connected to one end of the capacitor C _01, the capacitor C
The other end ₀₁ is connected to one end of the capacitor C _02, the other end of the capacitor C ₀₂ is connected to the emitter of the transistor Q _2. Connection point between transistors Q ₁ and Q ₂ and capacitors C ₀₁ and C ₀₂
A load circuit is connected between the connection points. The load circuit, the capacitor C ₃ connected in parallel to the discharge lamp la, the discharge lamp lighting circuit inductor L ₂ connected in series are connected. A connection point between the transistors Q ₁ and Q ₂ is connected to one end of an AC power supply Vs via an AC filter F. diode
The connection point between D ₃ and D ₄ is connected to the other end of the AC power supply Vs via the inductor L ₁ and the AC filter F.

 FIG. 9 is an operation waveform diagram of the above circuit.

Hereinafter, the operation will be described. When the AC power source Vs is in a positive half cycle, when transistor Q ₁ is turned on, inductor L _1, diode D _3, a current flows from the AC power source Vs through a path passing through the transistor Q ₁ to the inductor L _1, inductor L ₁ currents continue to increase with a slope proportional to the instantaneous value of the input AC voltage Vin, energy is stored in inductor L _1. When the transistor Q ₁ is turned off, the energy of the inductor L ₁ is a diode D _3, capacitor C _01,
C ₀₂ , released in the path through diode D ₂ and capacitor C
₀₁ and C ₀₂ are charged. Then, during the positive half cycle of the AC power source Vs, by repeating the above process, the envelope of the current flowing through the inductor L ₁ for positive period may be a sine wave.

Next, in the negative half cycle of the AC power source Vs, when the transistor Q ₂ is turned on, the transistor Q _2, the diode D _4, a current flows from the AC power source Vs in a path through the inductor L ₁ to the inductor L _1. Current flowing through the inductor L ₁ is a slope proportional to the instantaneous value of the input AC voltage Vin, the case of the positive half cycle continue to increase in the opposite direction, energy is stored in inductor L _1. When transistor Q ₂ is turned off, the energy of the inductor L ₁ is a diode D _1, the capacitor C
_01, C _02, is released through a path passing through the diode D _4, a capacitor C _01, C ₀₂ is charged. By repeating the above process during the negative half cycle of the AC power supply Vs, the inductor
It may be sinusoidal even for negative period the envelope of the current flowing in L _1. In addition, the transistors Q ₁ and Q ₂ are alternately turned on and off, so that the high frequency voltage V
Is applied.

In the above circuit, the transistors Q ₁ and Q ₂ are alternately turned on and off at a high speed, so that the chopper operation can be performed in an AC manner in synchronization with the positive and negative half cycles of the AC power supply Vs. Then, by inserting the AC filter F in the preceding stage, the input current can be made continuous, and the distortion rate of the input current can be reduced. Further, the input current at this time can be made into a sine wave shape having substantially the same phase as the input voltage, and the input power factor becomes substantially 1.

Conventional Example 3 The circuit shown in FIG. 8 can be operated as shown in the operation waveform diagram of FIG. In the tenth operation example shown in FIG, when the AC power source Vs is a positive half cycle, the transistor Q ₁ is a high-frequency manner on and off the drive, transistor
Q ₂ is turned off. Further, when the AC power source Vs is a negative half cycle, the transistor Q ₂ is a high-frequency manner on and off the drive, the transistor Q ₁ is turned off. Hereinafter, the operation of the above circuit will be described in detail.

First, when the AC power source Vs is a positive half cycle, when transistor Q ₁ is turned on, inductor L _1, diode D _3,
Inductor from the AC power source Vs through a path passing through the transistor Q ₁
Current flows through L _1, the current value increases with a gradient proportional to the instantaneous value of the input AC voltage Vin. At this time, a current flows from the capacitor C ₀₁ to the load circuit via the transistor Q _1. Next, when the transistor Q ₁ is turned off, the inductor
L ₁ , diode D ₃ , capacitor C ₀₁ , load circuit, path through AC power supply Vs, inductor L ₁ , diode
In a path passing through D ₃ , capacitors C ₀₁ and C ₀₂ , diode D ₂ , and AC power supply Vs, energy of inductor L ₂ is released, and capacitors C ₀₁ and C ₀₂ are charged.

Thus, the AC power source Vs is positive half cycle, which transistor Q ₁ is also serves as a switching element for the load current supplied to the switching element of the chopper, the transistor Q ₂ is at rest.

Then, when the AC power source Vs is a negative half cycle, the transistor Q ₂ is turned on, the AC power source Vs, the transistor Q _2,
The path through diode D ₄ and inductor L ₁
Current flows through L _1, the current value is gradually increased with a slope proportional to the instantaneous value of the input AC voltage Vin. In this case, the load circuit from the capacitor C _02, a current flowing through the load circuit path through the transistor Q _2. Next, when the transistor Q ₂ is turned off, AC power source Vs, a load circuit, a capacitor C _02, a diode D _4, path through the inductor L _1, and an AC power source Vs,
The energy passing through the diode D ₁ , the capacitors C ₀₁ and C ₀₂ , the diode D ₄ , and the inductor L ₁ releases the energy of the inductor L ₁ and charges the capacitors C ₀₁ and C ₀₂ .

Thus, the AC power source Vs is negative half cycle, which transistor Q ₂ serves also as the function of the switching element for the load current supplied to the switching element of the chopper, the transistor Q ₁ is at rest.

[SUMMARY OF THE INVENTION Incidentally, in the conventional example 1, the transistor Q ₂ is also used as a switching element for the chopper for the inverter, there is a problem that stress is applied only to the transistor Q _2. Further, in the conventional example 2 or 3, but equally stress to the transistor Q _1, Q ₂ is distributed, the chopper current flows through the transistors Q _1, Q _2, the current capacity of the transistors Q _1, Q ₂ is increased There is a problem.

The present invention has been made in view of such a point, and an object of the present invention is to control an input current and a load current without increasing a load on a switching element, to provide a high input power factor, and to reduce an input current distortion. It is an object of the present invention to provide an inverter device capable of reducing the size.

[Means for Solving the Problems] In the inverter device according to the present invention, as shown in FIG. 8 (or FIG. 3), in order to solve the above-described problems, the first device and the first device are connected in series. A second switching element and first and second capacitors C ₀₁ and C ₀₂ connected in series;
Are connected in parallel, an AC power supply Vs is connected to at least one end of the first and second switching elements via an inductance element and a rectifying element, and a connection point of the first and second switching elements is First and second capacitors C ₀₁ , C
_{In the} inverter device configured by connecting a load circuit between the connection points of ₀₂ , the other switching element is switched a plurality of times during a period in which no current flows in a closed circuit between the AC power supply, the inductance element, and the one switching element. An on-off period is provided to output a rectangular wave voltage to the load circuit.

Here, the first and second switching elements are configured by, for example, connecting the diodes D ₁ and D ₂ in antiparallel to the transistors Q ₁ and Q ₂ and do not block the reverse current.

[Operation] In the present invention, the switching element on the side that operates for chopper current control is stopped, and the switching on the side stopped by the feedback current of the load generated by the on / off operation of the other switching element. A reverse current is passed through the element, and the apparently stopped switching element is turned on. Thereby, the chopper operation can be performed by flowing the input current through the load circuit, and the input power factor can be increased and the input current distortion can be reduced. Therefore, since the chopper current flows through the load, it is not necessary to supply a current to the switching element, and the loss of the switching element is significantly reduced.

Embodiment 1 FIG. 1 is an operation waveform diagram of an embodiment of the present invention. In this embodiment, the circuit shown in FIG. 8 is used, but the control method is different. That is, the AC power source Vs is the positive half cycle (Vin> 0), the transistor Q ₁ is turned off, only the transistor Q ₂ is turned on and off. In the negative half cycle (Vin <0), the transistor Q ₁
Only it is turned on and off, transistor Q ₂ is turned off.
This is because the period during which the transistors Q ₁ and Q ₂ are operated is reversed as compared with the operation waveform diagram of the conventional example 3 (FIG. 10). Therefore, in this embodiment, both the transistors Q ₁ and Q ₂ do not control the chopper on the input side but control only the output on the load side, and the load current I la as shown in FIG. Flows. The input current is
Starts to flow when the feedback to the capacitor C ₀₁ or C ₀₂ of the load current I la.

First, when the positive half cycle of the AC power source Vs (Vin> 0), when the transistor Q ₂ is turned on, the capacitor C ₀₂
The load circuit, current flows into the transistor Q ₂ from. At this time, the input current stops. Further, the transistor Q ₂ is turned off, the diode D ₁ from the load circuit, the capacitor C ₀₁
The feedback current flows to. Further, AC filters F from the AC power source Vs, an inductor L _1, diode D _3, capacitor C _01,
A current also flows from the input to the AC power supply Vs via the load circuit and the AC filter F via the load (see the period Ta in FIG. 2A). When the load current becomes 0, the AC power
A current flows from Vs to the AC power supply Vs through the AC filter F, the inductor L ₁ , the diode D ₃ , the capacitors C ₀₁ and C ₀₂ , the diode D ₂ , and the AC filter F, and the capacitors C ₀₁ and C ₀₂
(See period Tb in FIG. 2 (b)).

When the transistor Q ₂ is turned off, the load current is a feedback current to the capacitor C ₀₁ through the diode D _1.
At this time, the voltage between both ends of the load is substantially equal to the voltage of the capacitor _C01 . Accordingly, both ends of the transistor Q ₁ is, only the ON voltage of the diode D _1, if you look at the voltage only, the same state as the transistor Q ₁ is turned on. However, since the transistor Q ₁ is not turned on, no current flows through the transistor Q _1, a capacitor
C _01, the current in inductor L ₁ from the collector of the transistor Q ₁ to the emitter side to flow through the load. Become the feedback current is zero load, the diode D ₁ is turned off, the capacitor C ₀₁ with the energy stored in the inductor L _1, C
Charge ₀₂ .

Then, when the AC power source Vs is negative half cycle (Vin <0), when the transistor Q ₁ is turned on, the capacitor C ₀₁
From the current flows to the load through the transistor Q _1. At this time, the input current has stopped. Also, the transistor Q ₁
Turns off, the capacitor C ₀₂ , diode D ₂
A feedback current flows through. Also, from the AC power supply Vs,
Filter F, load, the capacitor C _02, a diode D _4, inductor L _1, AC filter F, a current flows through the AC power source Vs, a period of a current also flows from the input through the load (FIG. 2 (b) Tc). When the load current becomes zero, the AC from the power supply Vs, AC filter F, the diode D _1, the capacitor C _01, C _02, a diode D _4, inductor L _1, AC filter F, a current flows to the AC power source Vs , Capacitor C
₀₁ and C ₀₂ are charged (see period Td in FIG. 2 (b)).

The reason why the input current flows is the same as the case of Vin> 0, a voltage approximately equal to the voltage of the capacitor C ₀₂ during feedback of the load current is generated across the load, the diode
D ₂ is turned on, the transistor Q ₂ is the same state as is on if you look at only the voltage. However, since the transistor Q ₂ is not on, the input current flows through the load and the capacitor C _02. When the load feedback current becomes 0,
Diode D ₂ turns off, diodes D ₁ and D ₄ turn on,
Capacitor C ₀₁ by the energy stored in the inductor L _1,
To charge the C _02.

In this embodiment, as described above, only the load current flows through the switching element, so that the loss in the switching element is significantly smaller than in the conventional examples 1 to 3.
Therefore, the size of the switching element can be reduced. In addition, since a chopper effect is performed via a load and an input current flows, an input power factor is high and an input current distortion is small.

Embodiment 2 FIG. 3 is a circuit diagram of another embodiment of the present invention. This embodiment is an example in which the present invention is applied to Conventional Example 1. However, the capacitors C ₀₁ and C ₀₂ are large-capacity smoothing capacitors, and the series circuit thereof is a smoothing capacitor C ₀ shown in FIG.
Also serves as a work. Each of the capacitors C ₀₁ and C ₀₂ has a function as a power source for a load, similarly to the DC cut capacitor C ₁ shown in FIG.

FIG. 4 is an operation waveform diagram of the present embodiment. First, period T ₁
In, it is turned on and off only the transistor Q _1. The current I la flows through the load. Also transistors
By the feedback of the load current at the time to Q ₁ off, just like the input current flows as in Example 1. During this time, since only flows only load currents in transistors Q _1, loss of the transistor Q ₁ is small. During this period, the capacitor _C01 performs charging and discharging, and the capacitor _C02 performs only charging.

Next, in a period T _2, are turned on and off only the transistor Q _2. Load current flows in opposite direction to the period T ₁ in. In the period T _2, the transistor Q ₂ is at the same time as controlling the load current, also controls the input chopper current. In the period T _2, the capacitor C ₀₁ is charged only, capacitor C ₀₂ performs charging and discharging.

In this embodiment, in order to keep the voltages of the capacitors C ₀₁ and C ₀₂ at substantially the same voltage, it is appropriate to set the capacitance to be large. The periods T ₁ and T ₂ are appropriately switched so as to maintain the voltage balance between the capacitors C ₀₁ and C ₀₂ . It is preferable that the switching timing is synchronized with the polarity reversal of the AC power supply Vs in order to always minimize the distortion of the input current.

In the present embodiment, in the period T _1, the input current flows only on-off switching element for current control of the load. Accordingly, a rectangular wave voltage can be applied to the discharge lamp la by the circuit shown in FIG. 3, and a rectangular wave lighting circuit can be easily realized.

In this embodiment, the same period the transistor Q ₂ in the transistor Q _1, the period T ₂ in the period T _1, when the switching at the same on-time, the magnitude of the input current does not become the same, when the period T ₂ The larger the input current is.
This is because better period T ₂ is longer the time that the current flows from the AC power source Vs to the inductor L _1, it is because a larger current. If it is desired to equalize the input current in a period T ₁ and T _2, in the period T _2, it may moderately shorten the ON time or period. In this way, the periods T ₁ and T ₂
Does not need to be synchronized with the power supply polarity.

Note that, as shown in FIG. 5, the same effect even when connected to the inductor L ₁ on the side of the AC power source Vs than the diode bridge D ₃ to D ₆ is obtained of course.

Embodiment 3 FIG. 6 is an operation waveform diagram of still another embodiment of the present invention. In the control method shown in FIG. 1 described above, since the magnitude of the input current is smaller than that in the conventional example, the capacitors C ₀₁ and C
₀₂ output voltage decreases. The voltage drop of the capacitors C ₀₁ and C ₀₂ causes a decrease in the lamp current, that is, the output. Therefore, in order to prevent the output from lowering, the capacitor C ₀₁ , C ₀₁
It is preferable to prevent the voltage drop of ₀₂ .

In Figure 6, it uses a method of controlling the period T ₃ in the present invention uses a method of controlling the period T ₄ Conventional Example 2.
The lengths of these periods T ₃ and T ₄ are not particularly limited. For example, a method of controlling the voltages of the capacitors C ₀₁ and C ₀₂ to be within a certain range, or a method of controlling the lamp current value to a certain range It is possible to consider a method of controlling within.

In this example, the period T ₃ and switching-life of T ₄ is, because it does not match the timing of the reversal of the supply polarity, becomes discontinuous to the current waveform of a half cycle that contains a switching time point, the input distortion is increased.
However, since this point is only a half cycle of the switching, the influence on the whole is small.

In this case, similarly to the operation example of FIG. 4, if the period T ₃ and switching period of T ₄ is consistent with the reversal timing of the power supply polarity, input current distortion can be minimized.

According to the present invention, the first and second switching elements connected in series and the first and second capacitors connected in series are connected in parallel, and the first and second switching elements are connected. An AC power supply is connected to at least one of both ends of the element via an inductance element and a rectifying element, and a load circuit is provided between a connection point of the first and second switching elements and a connection point of the first and second capacitors. In the inverter device configured to be connected, a period in which no current flows in a closed circuit between the AC power supply, the inductance element, and the one switching element is provided with a period in which the other switching element is turned on and off a plurality of times, and the load circuit has a rectangular shape. It is configured to output a wave voltage, so that the feedback current of the load circuit when the switching element is turned off is used to allow the load circuit to pass through. Thus, the chopper operation can be performed by flowing the input current to increase the input power factor and reduce the input current distortion. Further, since the chopper current flows through the load circuit, it is not necessary to supply a current to the switching element, and there is an effect that the loss of the switching element is significantly reduced.

[Brief description of the drawings]

FIG. 1 is an operation waveform diagram of one embodiment of the present invention, FIGS. 2 (a) and 2 (b) are main portion waveform diagrams of the above embodiment, FIG. 3 is a circuit diagram of another embodiment of the present invention, and FIG. The figure shows the operation waveform diagram of the above,
FIG. 6 is a circuit diagram of a modification of the above, FIG. 6 is an operation waveform diagram of still another embodiment of the present invention, FIG. 7 is a circuit diagram of a conventional example, FIG. 8 is a circuit diagram of another conventional example, FIG. 9 is an operation waveform diagram of the above, and FIG. 10 is an operation waveform diagram of still another conventional example. S ₁ and S ₂ are control signals.

Claims (1)

(57) [Claims] A first and a second switching element connected in series and a first and a second capacitor connected in series are connected in parallel, and both ends of at least one of the first and second switching elements are connected. And a load circuit connected between a connection point of the first and second switching elements and a connection point of the first and second capacitors. In the inverter device, a period in which a current does not flow in a closed circuit of the AC power supply, the inductance element, and the one switching element is provided with a period in which the other switching element is turned on and off a plurality of times, and a rectangular wave voltage is output to a load circuit. An inverter device characterized by comprising: