JP2015139312A - Switching power supply arrangement and electric power converter unit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an electric power converter unit having an active switch at the secondary side capable of reducing a circuit size.SOLUTION: A bridge circuit converts a DC voltage into an AC voltage. A transformer T includes: a primary winding connected to a bridge circuit; and a secondary winding N2 electromagnetically coupled with the primary winding. A secondary side rectifying circuit 11 rectifies an AC voltage which is input from the secondary winding N2. A smoothing circuit 12 smooths the voltage rectified by the secondary side rectifying circuit 11. A fifth switching element S5 is positioned in a current path between the secondary side rectifying circuit 11 and the smoothing circuit 12. In a ninth diode D9, the low potential side input terminal of the smoothing circuit 12 is connected to an anode terminal; and a high potential side input terminal of the smoothing circuit 12 is connected to a cathode terminal.

Description

  The present invention relates to an insulating switching power supply device and a power conversion device including the switching power supply device.

  In general, an insulation type switching power supply apparatus is configured such that an inverter is provided on the primary side and a rectifying and smoothing circuit is provided on the secondary side. In this configuration, there are a control method in which the output is adjusted by the switching element (active switch) constituting the primary side inverter, and a control method in which the switching element is provided in the secondary side rectifying and smoothing circuit and the output is adjusted by this switching element (For example, refer to Patent Document 1).

JP 2002-238257 A

  As described above, in the power conversion device including the active switch on the secondary side, the number of drive circuits is increased as compared with the power conversion device including no active switch on the secondary side. Since the primary side and the secondary side have different reference potentials, it is difficult to share the primary side active switch drive circuit and the secondary side active switch drive circuit. As the number of drive circuits increases, the circuit scale increases accordingly.

  This invention is made | formed in view of such a condition, The objective is to provide the technique of reducing the circuit scale of the power converter device provided with an active switch in a secondary side.

  In order to solve the above problems, a switching power supply device according to an aspect of the present invention includes a bridge circuit that converts a DC voltage into an AC voltage, a primary winding connected to the bridge circuit, and an electromagnetic coupling with the primary winding. A transformer including a secondary winding, a rectifier circuit that rectifies an AC voltage input from the secondary winding, a smoothing circuit that smoothes the voltage rectified by the rectifier circuit, and a current between the rectifier circuit and the smoothing circuit A switching element inserted in the path; and a diode having an anode terminal connected to the low potential side input terminal of the smoothing circuit and a cathode terminal connected to the high potential side input terminal of the smoothing circuit.

  Another aspect of the present invention is a power converter. This device includes the above-described switching power supply device and a driving device that drives the switching element.

  It should be noted that any combination of the above-described constituent elements and a conversion of the expression of the present invention among methods, circuits, devices, systems, etc. are also effective as an aspect of the present invention.

  ADVANTAGE OF THE INVENTION According to this invention, the circuit scale of a power converter device provided with an active switch on the secondary side can be made small.

It is a figure which shows the structure of the power converter device which concerns on the comparative example 1. FIG. It is a figure which shows an example of the control method of the switching power supply device of FIG. It is a figure which shows the structure of the power converter device which concerns on the comparative example 2. It is a figure which shows the detail of the secondary side circuit of a switching power supply device, and a control apparatus of the power converter device which concerns on the comparative example 1. FIG. It is a figure which shows the detail of the secondary side circuit of a switching power supply device, and a control apparatus of the power converter device which concerns on Embodiment 1 of this invention. It is a figure which shows the detail of the secondary side circuit of a switching power supply device, and a control apparatus of the power converter device which concerns on Embodiment 2 of this invention. It is a figure which shows the detail of the secondary side circuit of a switching power supply device, and a control apparatus of the power converter device which concerns on Embodiment 3 of this invention. It is a figure which shows the detail of the secondary side circuit of a switching power supply device, and a control apparatus of the power converter device which concerns on Embodiment 4 of this invention. It is a figure which shows an example of the control method of the switching power supply device of FIG. It is a figure which shows the detail of the secondary side circuit of a switching power supply device, and a control apparatus of the power converter device which concerns on Embodiment 5 of this invention. It is a figure which shows the detail of the secondary side circuit and control apparatus of a switching power supply device of the power converter device which concerns on Embodiment 6 of this invention. It is a figure which shows the structure of the electrical storage system with which the power converter device which concerns on Embodiment 1-6 is used. It is a figure which shows the structure of the vehicle in which the power converter device which concerns on Embodiment 1-6 is used. It is a figure which shows the structure of the charger in which the power converter device which concerns on Embodiment 1-6 is used.

  FIG. 1 is a diagram illustrating a configuration of a power conversion apparatus 100 according to the first comparative example. The power conversion device 100 according to the comparative example 1 includes a switching power supply device 10, an output voltage detection circuit 15, and a control device 20. The switching power supply device 10 is a secondary side phase shift type isolated DC-DC converter, and is a full bridge circuit, a first coil L1, a second coil L2, a sixth capacitor C6, a seventh capacitor C7, a transformer T, a first 3 coil L3, 4th coil L4, a secondary side rectifier circuit and a smoothing circuit are included.

  The full bridge circuit converts a DC voltage supplied from the DC power source E into an AC voltage. The full bridge circuit includes a first switching element S1, a second switching element S2, a third switching element S3, and a fourth switching element S4 that are connected in a full bridge. Specifically, a first arm including a first switching element S1 on the upper side and a second switching element S2 on the lower side, and a second arm including a third switching element S3 on the upper side and a fourth switching element S4 on the lower side. The first arm and the second arm are connected in parallel.

  A first diode D1 and a first capacitor C1 are connected in parallel with the first switching element S1. Similarly, the second switching element S2 to the fourth switching element S4 are connected in parallel with the second diode D2 to the fourth diode D4 and the second capacitor C2 to the fourth capacitor C4, respectively. The first diode D1 to the fourth diode D4 are reverse-biased to the first switching element S1 to the fourth switching element S4, respectively. The first capacitor C1 to the fourth capacitor C4 are snubber capacitors. For the first switching element S1 to the fourth switching element S4, for example, semiconductor switching elements such as MOSFETs and IGBTs are used. FIG. 1 illustrates an example using an n-channel IGBT. A p-channel type semiconductor switching element may be used for the first switching element S1 and the third switching element S3.

  In FIG. 1, the primary side is constituted by a partial resonance type full bridge circuit. The partial resonance type full bridge circuit performs commutation by utilizing the resonance operation only at the time of switching, and operates in a non-resonance mode in other modes. In FIG. 1, a resonance pole type configuration that can ensure zero voltage commutation even at light load is adopted. The sixth capacitor C6 and the seventh capacitor C7 are connected in series, and the sixth capacitor C6 and the seventh capacitor C7 connected in series are connected in parallel to the DC power source E. The first coil L1 is connected between a node between the sixth capacitor C6 and the seventh capacitor C7 and a node between the first switching element S1 and the second switching element S2. The second coil L2 is connected between a node between the sixth capacitor C6 and the seventh capacitor C7 and a node between the third switching element S3 and the fourth switching element S4. The resonant pole circuit of FIG. 1 is configured such that the first coil L1 and the second coil L2 share the sixth capacitor C6 and the seventh capacitor C7, and the capacitor for the first coil L1 and the capacitor for the second coil L2 are used. A capacitor may be provided separately.

  The first coil L1 and the second coil L2 are auxiliary inductors for resonance, and current is supplied from the sixth capacitor C6 and the seventh capacitor C7. When switching the first switching element S1 to the fourth switching element S4, zero voltage switching is realized by utilizing partial resonance generated by a resonance auxiliary inductor and a capacitor connected in parallel to the switching element. Specifically, when the first switching element S1 to the fourth switching element S4 are switched off, the rising of the switching element voltage is delayed by the capacitors C1 to C4 connected in parallel, so that switching that occurs due to a transient crossing of voltage and current is generated. Power loss can be reduced. On the other hand, when the first switching element S1 to the fourth switching element S4 are switched on, the first diode D1 to the first diode D1 arranged in parallel to the first switching element S1 to the fourth switching element S4 from the first coil L1 and the second coil L2. The switching loss of the first switching element S1 to the fourth switching element S4 can be reduced by turning on when the current flows through the four diodes D4 (that is, the state where the voltage applied to the switching elements is almost zero). .

  The transformer T is a high-frequency transformer including a primary winding N1 and a secondary winding N2. The primary winding N1 and the secondary winding N2 are coupled by electromagnetic induction. The transformer T insulates the primary side and the secondary side, and transforms according to the winding ratio of the primary winding N1 and the secondary winding N2. Both ends of the primary winding N1 are connected to both output ends of the full bridge circuit. A third coil L3 and a fourth coil L4 are provided at both ends of the secondary winding N2, respectively. The third coil L3 and the fourth coil L4 may be a leakage inductance of the secondary winding N2, or may be an inductor element.

  The secondary side rectifier circuit rectifies the AC voltage input from the secondary winding N2. The secondary side rectifier circuit includes a fifth diode D5, a sixth diode D6, a seventh diode D7, an eighth diode D8, a fifth switching element S5, and a sixth switching element S6. The fifth diode D5 to the eighth diode D8 are connected by a full bridge to form a rectifier circuit. The fifth switching element S5 and the sixth switching element S6 are inserted into the current path of the rectifier circuit, and adjust the amount of power extracted from the transformer T. For the fifth switching element S5 and the sixth switching element S6, semiconductor switching elements such as MOSFET and IGBT are used, for example.

  The smoothing circuit smoothes the voltage rectified by the secondary side rectifier circuit. The smoothing circuit in FIG. 1 includes an LC filter including a fifth coil L5 and a fifth capacitor C5. The configuration of the smoothing circuit in FIG. 1 is an example, and other configurations may be used. For example, a configuration in which the fifth coil L5 is omitted is possible.

  The output voltage detection circuit 15 detects the output voltage of the switching power supply device 10 supplied to the load 30. In FIG. 1, the voltage across the fifth capacitor C5 is detected as the output voltage. The output voltage detection circuit 15 can be constituted by an error amplifier circuit, for example. The error amplifier circuit is composed of a combination of an operational amplifier and a passive element. The output voltage detection circuit 15 outputs the detected voltage to the control device 20.

  The control device 20 drives the switching power supply device 10 by controlling on / off of the first switching element S1 to the sixth switching element S6. The control device 20 adaptively changes the phases of the fifth switching element S5 and the sixth switching element S6 according to the output voltage of the switching power supply device 10 supplied from the output voltage detection circuit 15. Thereby, the output voltage of the switching power supply device 10 is stabilized. Details of the configuration of the control device 20 will be described later.

  FIG. 2 is a diagram illustrating an example of a control method of the switching power supply device 10 of FIG. Transitions of the voltages Vc1 and Vc4 of the first capacitor C1 and the fourth capacitor C4 are drawn by bold lines, and changes of the voltages Vc2 and Vc3 of the second capacitor C2 and the third capacitor C3 are drawn by thin lines. The transition of the current id7 flowing through the seventh diode D7 on the secondary side is drawn with a thick line, and the transition of the current is5 (id5) flowing through the fifth switching element S5 and the fifth diode D5 on the secondary side is drawn with a thin line. Yes. Similarly, the transition of the current id8 flowing through the secondary-side eighth diode D8 is drawn with a bold line, and the transition of the current is6 (id6) flowing through the secondary-side sixth switching element S6 and the sixth diode D6 is drawn with a thin line. ing.

  The control device 20 supplies the primary side drive signal to the control terminals of the first switching element S1 to the fourth switching element S4 (the gate terminal in the case of FET and IGBT, and the base terminal in the case of the bipolar transistor), and the fifth switching element S5, a secondary drive signal is supplied to the control terminal of the sixth switching element S6.

  A forward current flows through the transformer T when the first switching element S1 and the fourth switching element S4 are in the on state and the second switching element S2 and the third switching element S3 are in the off state. On the other hand, a reverse current flows through the transformer T when the second switching element S2 and the third switching element S3 are in the on state and the first switching element S1 and the fourth switching element S4 are in the off state. The first switching element S1 and the fourth switching element S4 function as a primary-side forward switching element that is inserted into a path that supplies a forward current to the primary winding N1. On the other hand, the second switching element S2 and the third switching element S3 function as a primary-side reverse switching element inserted in a path for supplying a reverse current to the primary winding N1.

  In addition, a dead time is provided when switching between a period in which a forward current flows and a period in which a reverse current flows. During the dead time, the first switching element S1 to the fourth switching element S4 are all turned off. The capacitance component and the inductance component resonate during the dead time, and the capacitor connected in parallel to the switching element that is turned on next is discharged.

  The control device 20 drives the primary-side forward switching element and the primary-side reverse switching element with a fixed duty ratio and a fixed phase. Specifically, the primary side forward switching element and the primary side reverse switching element are driven complementarily with a duty of 50%, excluding the dead time. In the switching power supply device 10 of FIG. 1, the output voltage is adjusted on the secondary side, so the duty ratio and phase of the primary forward switching element and the primary reverse switching element are fixed.

  The forward current also flows through the secondary side rectifier circuit during at least a part of the period during which the forward current flows through the transformer T. Specifically, the forward current flows in the secondary rectifier circuit during the period in which the sixth switching element S6 is controlled to be in the ON state during the period in which the forward current flows in the transformer T. During the period in which the forward current flows, the sixth diode D6 and the seventh diode D7 are turned on, the fifth diode D5 and the eighth diode D8 are turned off, the sixth switching element S6 is turned on, and the fifth switching element S5 is turned off. It becomes a state.

  On the other hand, the reverse current also flows in the secondary rectifier circuit during at least a part of the period in which the reverse current flows in the transformer T. Specifically, the reverse current flows in the secondary rectifier circuit during the period in which the fifth switching element S5 is controlled to be in the ON state during the period in which the reverse current flows in the transformer T. During the period in which the reverse current flows, the fifth diode D5 and the eighth diode D8 are turned on, the sixth diode D6 and the seventh diode D7 are turned off, the fifth switching element S5 is turned on, and the sixth switching element S6 is turned off. It becomes a state.

  That is, the sixth diode D6 and the seventh diode D7 conduct the forward current from the secondary winding N2, and block the reverse current from the secondary winding N2. On the other hand, the fifth diode D5 and the eighth diode D8 block the forward current from the secondary winding N2 and conduct the reverse current from the secondary winding N2. Thereby, the AC voltage supplied from the secondary winding N2 is full-wave rectified.

  The fifth switching element S5 functions as a secondary-side forward switching element inserted in a forward current path including the fifth diode D5 and the eighth diode D8. On the other hand, the sixth switching element S6 functions as a secondary-side reverse switching element inserted in a reverse current path including the sixth diode D6 and the seventh diode D7.

  The control device 20 drives the secondary-side forward switching element and the secondary-side backward switching element with a fixed duty ratio and a varying phase. Specifically, the secondary-side forward switching element and the secondary-side reverse switching element are adaptively phase-shifted while being complementarily driven with a duty of 50%. In this embodiment, since the phase shift method is adopted instead of the PWM method, the control device 20 controls the phase rather than the duty ratio of the secondary forward switching element and the secondary reverse switching element. That is, the phases of the secondary forward switching element and the secondary reverse switching element are adaptively changed according to the output voltage of the switching power supply device 10.

  More specific description will be given below. The control device 20 delays the phase of the secondary forward switching element and the secondary backward switching element relative to the phase of the primary forward switching element and the primary reverse switching element (phase difference α in FIG. 2). And the output voltage of the switching power supply 10 is stabilized. When the output voltage of the switching power supply device 10 increases, the control device 20 delays the phases of the secondary-side forward switching element and the secondary-side reverse switching element to increase the phase difference α, and the amount of power extracted from the transformer T is increased. Reduce. On the other hand, when the output voltage of the switching power supply device 10 becomes low, the control device 20 advances the phases of the secondary forward switching element and the secondary reverse switching element to reduce the phase difference α and take it out from the transformer T. Increase the amount of power. A state in which the phase difference α is zero is a state in which the largest amount of power is extracted from the transformer T, and the amount of power to be extracted decreases as the phase difference α increases.

  FIG. 3 is a diagram illustrating a configuration of the power conversion device 100 according to the second comparative example. The power conversion device 100 according to the comparative example 2 is obtained by replacing the secondary side rectifier circuit of the power conversion device 100 according to the comparative example 1 from a full bridge type to a center tap type. The control method of the switching power supply device 10 according to Comparative Example 2 is the same as the control method of the switching power supply device 10 according to Comparative Example 1 shown in FIG.

  In Comparative Example 2, the output terminal of the center tap type rectifier circuit is connected to the high potential side input terminal of the smoothing circuit constituted by the fifth coil L5 and the fifth capacitor C5, and the center terminal of the secondary winding N2 is connected. The tap is connected to the low potential side input terminal of the smoothing circuit. The anode terminal of the fifth diode D5 is connected to one end of the secondary winding N2 via the fifth switching element S5 and the fourth coil L4, and the anode terminal of the sixth diode D6 is connected to the sixth switching element S6 and the third coil The other end of the secondary winding N2 is connected via the coil L3.

  The fifth diode D5 conducts the forward current from the secondary winding N2, and interrupts the reverse current from the secondary winding N2. On the other hand, the sixth diode D6 conducts the reverse current from the secondary winding N2, and cuts off the forward current from the secondary winding N2. The fifth switching element S5 acts as a secondary forward switching element that switches forward current, and the sixth switching element S6 acts as a secondary backward switching element that switches reverse current. The anode terminal of the ninth diode D9 is connected to a node between the center tap of the secondary winding N2 and the low potential side input terminal of the smoothing circuit, and the cathode terminal of the ninth diode D9 is connected to the smoothing terminal. It is connected to the high potential side input terminal of the circuit. The ninth diode D9 is a diode for forming a current return path.

  Comparing the center tap type rectifier circuit according to the comparative example 2 and the full bridge type rectifier circuit according to the comparative example 1, the center tap type can reduce the number of diodes. It is necessary to increase the breakdown voltage of the 6 diode D6.

  FIG. 4 is a diagram illustrating details of the secondary circuit of the switching power supply device 10 and the control device 20 of the power conversion device 100 according to the first comparative example. Hereinafter, in this specification, attention is paid to drive control of the fifth switching element S5 and the sixth switching element S6 included in the secondary rectifier circuit. Hereinafter, in order to simplify the drawing, the configuration relating to the drive control of the primary circuit in the switching power supply device 10 and the primary circuit in the control device 20 is omitted in the drawing.

  In FIG. 4, the reference terminal (source terminal in MOSFET, emitter terminal in IGBT, bipolar transistor) of the fifth switching element S5 is connected to the input terminal of the fifth coil L5. The conduction terminal of the fifth switching element S5 (drain terminal for MOSFET, collector terminal for IGBT, bipolar transistor) is connected to the cathode terminal of the fifth diode D5. The reference terminal of the sixth switching element S6 is connected to one terminal of the secondary winding N2, and the conduction terminal of the sixth switching element S6 is connected to the cathode terminal of the sixth diode D6.

  The reference potential applied to the reference terminal of the fifth switching element S5 is different from the reference potential applied to the reference terminal of the sixth switching element S6. Therefore, the control device 20 generates a drive signal corresponding to the reference potential of the fifth switching element S5 and a drive signal corresponding to the reference potential of the sixth switching element S6 by separate drive systems. Hereinafter, a configuration example of the control device 20 will be specifically described.

  The control device 20 includes a CPU 21 and a drive device 22. The driving device 22 includes a first gate buffer 23a, a first high-side DC-DC converter 24a, a first low-side DC-DC converter 25a, a first control logic circuit 26a, a first photocoupler 27a, a second gate buffer 23b, a second A high-side DC-DC converter 24b, a second low-side DC-DC converter 25b, a second control logic circuit 26b, and a second photocoupler 27b are provided.

  An external power supply is supplied to the CPU 21 and the driving device 22. The external power source is generated based on a commercial power source (not shown) or a power source supplied from a storage battery. The CPU 21 generates a control signal (digital signal) for the fifth switching element S5 and a control signal for the sixth switching element S6 according to the signal supplied from the output voltage detection circuit 15, and outputs the control signal to the control device 20.

  The first control logic circuit 26 a generates a drive voltage to be supplied to the first gate buffer 23 a in response to a control signal supplied from the CPU 21. The drive voltage is insulated by the first photocoupler 27a and supplied to the first gate buffer 23a. Similarly, the second control logic circuit 26b generates a drive voltage to be supplied to the second gate buffer 23b in accordance with a control signal supplied from the CPU 21. The drive voltage is insulated by the second photocoupler 27b and supplied to the second gate buffer 23b.

  The first gate buffer 23a drives the fifth switching element S5 according to the drive voltage supplied from the first control logic circuit 26a. The output terminal of the first gate buffer 23a is connected to the control terminal of the fifth switching element S5 via a current limiting element such as a gate resistor (not shown). The first gate buffer 23a can be constituted by, for example, an inverter in which a p-channel MOSFET and an n-channel MOSFET are connected in series.

  The fifth switching element S5 is on / off controlled according to the potential difference between the reference potential (source potential or emitter potential) applied to the reference terminal and the control potential (gate potential) applied to the control terminal. Therefore, the first gate buffer 23a needs to generate a control potential to be applied to the control terminal of the fifth switching element S5 with reference to the reference potential of the fifth switching element S5.

  The first high side DC-DC converter 24a generates the high side power supply potential of the first gate buffer 23a from the above external power supply. The first low side DC-DC converter 25a generates the low side power supply potential of the first gate buffer 23a from the external power supply. The first high-side DC-DC converter 24a and the first low-side DC-DC converter 25a can be composed of, for example, a step-down chopper. When the first gate buffer 23a has the above-described inverter configuration, the high-side power supply potential is applied to the source terminal of the p-channel MOSFET, and the low-side power supply potential is applied to the source terminal of the n-channel MOSFET.

  For example, the first high side DC-DC converter 24a generates a potential of +15 V with respect to the reference potential of the fifth switching element S5. For example, the first low-side DC-DC converter 25a generates a potential of −5V with respect to the reference potential of the fifth switching element S5. In this example, the first gate buffer 23a is controlled by a power supply voltage of 20V. 20V is an example and varies depending on the type and specification of the fifth switching element S5.

  The second gate buffer 23b needs to generate a control potential to be applied to the control terminal of the sixth switching element S6 with reference to the reference potential of the sixth switching element S6. The second high side DC-DC converter 24b generates the high side power supply potential of the second gate buffer 23b from the above external power supply. The second low side DC-DC converter 25b generates the low side power supply potential of the second gate buffer 23b from the above external power supply.

  For example, the second high-side DC-DC converter 24b generates a potential of + 15V with respect to the reference potential of the sixth switching element S6. For example, the second low-side DC-DC converter 25b generates a potential of −5V with respect to the reference potential of the sixth switching element S6. As described above, even if the power supply voltage of the first gate buffer 23a and the second gate buffer 23b is the same 20V, the reference potentials of the fifth switching element S5 and the sixth switching element S6 are different. The output potential of the two-gate buffer 23b is different.

  Therefore, the first high-side DC-DC converter 24a and the second high-side DC-DC converter 24b need to be provided separately, and the first low-side DC-DC converter 25a and the second low-side DC-DC converter 25b must also be provided separately. There is. For this reason, the number of components increases, which causes an increase in cost when an active switch is provided on the secondary side. Further, it is a factor that hinders downsizing of the power conversion apparatus 100.

  FIG. 5 is a diagram showing details of the secondary circuit of the switching power supply device 10 and the control device 20 of the power conversion device 100 according to Embodiment 1 of the present invention. The primary side circuit of the power conversion device 100 according to Embodiment 1 is the same as the primary side circuit of the power conversion device 100 according to Comparative Examples 1 and 2 shown in FIGS.

  As described above, the sixth diode D6 and the seventh diode D7 are diodes that conduct the forward current from the secondary winding N2. The fifth diode D5 and the eighth diode D8 are diodes that conduct reverse current from the secondary winding N2. The fifth switching element S5 is inserted in the forward current path between the fifth diode D5 and the first output node Nc of the secondary side rectifier circuit 11, and adjusts the forward power extracted from the transformer T. The sixth switching element S6 is inserted in the reverse current path between the seventh diode D7 and the first output node Nc, and adjusts the reverse power extracted from the transformer T. The reference terminals of the fifth switching element S5 and the sixth switching element S6 are commonly connected to the first output node Nc.

  More specific description will be given below. The anode terminal of the seventh diode D7 is connected to the first input node Na of the secondary side rectifier circuit 11 into which forward current flows from the secondary winding N2, and the cathode terminal of the seventh diode D7 is connected to the sixth switching element S6. Connected to the conduction terminal. The anode terminal of the fifth diode D5 is connected to the second input node Nb of the secondary side rectifier circuit 11 into which reverse current flows from the secondary winding N2, and the cathode terminal of the fifth diode D5 is connected to the fifth switching element S5. Connected to the conduction terminal.

  The cathode terminal of the eighth diode D8 is connected to the first input node Na, and the anode terminal of the eighth diode D8 is connected to the second output node Nd of the secondary side rectifier circuit 11. The first output node Nc is a high potential side output node of the secondary side rectifier circuit 11, and the second output node Nd is a low potential side output node of the secondary side rectifier circuit 11. The cathode terminal of the sixth diode D6 is connected to the second input node Nb, and the anode terminal of the sixth diode D6 is connected to the second output node Nd. The fifth switching element S5 is connected between the cathode terminal of the fifth diode D5 and the first output node Nc, and the sixth switching element S6 is between the cathode terminal of the seventh diode D7 and the first output node Nc. Connected to.

  The driving device 22 according to the first embodiment has the second high-side DC-DC converter 24b and the second low-side DC-DC converter 25b omitted as compared with the driving device 22 according to Comparative Example 1 shown in FIG. It is a configuration. In the first embodiment, the reference potential of the fifth switching element S5 and the reference potential of the sixth switching element S6 are common. Therefore, the high side power supply potential and the low side power supply potential of the first gate buffer 23a and the second gate buffer 23b can be shared. That is, the first high-side DC-DC converter 24a generates a common high-side power supply potential for the first gate buffer 23a and the second gate buffer 23b and supplies it to both in parallel. Similarly, the first low-side DC-DC converter 25a generates a common low-side power supply potential for the first gate buffer 23a and the second gate buffer 23b and supplies them to both in parallel.

  As described above, according to the first embodiment, the power supply voltages of the first gate buffer 23a and the second gate buffer 23b are generated by sharing the reference potentials of the fifth switching element S5 and the sixth switching element S6. Common power supply circuit. Therefore, the number of parts of the driving device 22 can be reduced, and the cost can be reduced. Further, the circuit scale of the entire power conversion apparatus 100 can be reduced.

  The fifth switching element S5 is connected between the anode terminal of the sixth diode D6 and the second output node Nd, and the sixth switching element S6 is connected between the anode terminal of the eighth diode D8 and the second output node Nd. It may be configured to connect between them. With this configuration, the reference potential of the fifth switching element S5 and the reference potential of the sixth switching element S6 can be shared. However, a switching element in which a parasitic diode that is forward biased in the direction from the source terminal to the drain terminal, such as a MOSFET, is basically not applicable. Any switching element that can cut off the current flowing from the source terminal to the drain terminal (or collector terminal) in the off state can be used.

  FIG. 6 is a diagram showing details of the secondary circuit of the switching power supply device 10 and the control device 20 of the power conversion device 100 according to Embodiment 2 of the present invention. The power conversion device 100 according to the second embodiment has a configuration in which a ninth diode D9 is added to the configuration of the power conversion device 100 according to the first embodiment. The anode terminal of the ninth diode D9 is connected to the low potential side input terminal of the smoothing circuit 12, and the cathode terminal of the ninth diode D9 is connected to the high potential side input terminal of the smoothing circuit 12. The ninth diode D <b> 9 is a diode for forming a current return path that bypasses the secondary side rectifier circuit 11.

  When the control method shown in FIG. 2 is adopted in the power conversion device 100 according to the comparative example 1 shown in FIG. 4, the seventh diode D7 and the eighth diode D8 are turned on, thereby causing the fifth coil from the load 30. There is a single reflux period in which current flows to L5. Due to the dead time of the first switching element S1 to the fourth switching element S4 on the primary side, a period in which the potential difference between both ends of the secondary winding N2 is eliminated becomes a single reflux period.

  Next, consider the case where the control method shown in FIG. 2 is adopted in power conversion apparatus 100 according to Embodiment 1 shown in FIG. In the control method shown in FIG. 2, the driving device 22 drives the fifth switching element S5 and the sixth switching element S6 in a complementary manner. Therefore, one of the fifth switching element S5 and the sixth switching element S6 is always on. Therefore, one of the current path formed by the fifth switching element S5, the fifth diode D5 and the sixth diode D6 or the current path formed by the sixth switching element S6, the seventh diode D7 and the eighth diode D8 is always effective. It is a state. Therefore, the power conversion device 100 according to Embodiment 1 can also secure a current path in a single return period.

  However, when connected to a load 30 that requires a large current, the loss of the fifth switching element S5 or the sixth switching element S6 due to the current generated during a single return period increases. For example, the forward drop voltage Vf of the fifth diode D5 to the eighth diode D8 is 0.5V, the forward drop voltage Vf of the fifth switching element S5 and the sixth switching element S6 is 1.0V, and the return current is 1.0A. Consider the example. The loss when the current flows through the path formed by the seventh diode D7 and the eighth diode D8 as in Comparative Example 1 is 1.0 W. The loss when the current flows back through the path formed by two diodes and one switching element as in the first embodiment is 2.0 W. Thus, the loss increases as the current increases.

  On the other hand, for example, if the ninth diode D9 having a forward drop voltage Vf of 0.4V is provided, the loss when the current flows through the path formed by the ninth diode D9 is 0.4W. Loss can be greatly reduced compared to the above example. When a plurality of current paths are formed in parallel, the current flows through a path having a low forward drop voltage Vf, and thus the return current passes through the ninth diode D9.

  As described above, according to the second embodiment, by adding the ninth diode D9, it is possible to reduce conduction loss during a single return period. In the first embodiment, when the reference terminals of the fifth switching element S5 and the sixth switching element S6 are commonly connected, the current needs to pass through the fifth switching element S5 or the sixth switching element S6 during a single return period. Therefore, compared with Comparative Example 1, the conduction loss in the single reflux period increases. In the second embodiment, by adding the ninth diode D9, it is possible to reduce the conduction loss in the single return period as compared with the comparative example 1 that passes through the two diodes. Although the circuit scale is increased by adding the ninth diode D9, the circuit scale reduction effect by deleting the second high-side DC-DC converter 24b and the second low-side DC-DC converter 25b is larger.

  As shown in FIG. 2, when the control method including the dead time is employed without completely controlling the fifth switching element S5 and the sixth switching element S6 in the opposite phase, a current path in a single return period is secured. Therefore, the ninth diode D9 is essential.

  FIG. 7 is a diagram showing details of the secondary circuit of the switching power supply device 10 and the control device 20 of the power conversion device 100 according to Embodiment 3 of the present invention. In the power conversion device 100 according to Embodiment 3, the arrangement of the fifth switching element S5 and the sixth switching element S6 in the power conversion device 100 according to Comparative Example 2 shown in FIG. 3 is changed.

  The anode terminal of the sixth diode D6 is connected to the first input terminal of the secondary side rectifier circuit 11 into which forward current flows from the secondary winding N2, and the anode terminal of the fifth diode D5 is connected to the secondary winding N2. To the second input terminal of the secondary side rectifier circuit 11 into which reverse current flows. The sixth switching element S6 is connected between the cathode terminal of the sixth diode D6 and the output node Nc of the secondary side rectifier circuit 11, and the fifth switching element S5 is connected to the cathode terminal of the fifth diode D5 and the output node Nc. Connected between.

  The anode terminal of the ninth diode D9 is connected to a node Ne between the center tap of the secondary winding N2 and the low potential side input terminal of the smoothing circuit 12, and the cathode terminal of the ninth diode D9 is connected to the smoothing circuit 12. To the high potential side input terminal.

  As described above, the reference potentials of the fifth switching element S5 and the sixth switching element S6 can be made common even if the center tap type is adopted for the secondary side rectifier circuit 11 as in the third embodiment. Therefore, the power supply circuit for generating the power supply voltage for the first gate buffer 23a and the second gate buffer 23b can be shared. Therefore, the power conversion device 100 according to the third embodiment also has the same effect as the power conversion device 100 according to the first and second embodiments.

  FIG. 8 is a diagram showing details of the secondary circuit of the switching power supply device 10 and the control device 20 of the power conversion device 100 according to Embodiment 4 of the present invention. In the power conversion device 100 according to Embodiment 4, the fifth switching element S5 and the sixth switching element S6 of the power conversion device 100 according to Embodiment 1-3 are combined into one.

  The secondary side rectifier circuit 11 is configured by a normal diode bridge circuit that does not include an active switch. The fifth switching element S5 is inserted in the current path between the secondary side rectifier circuit 11 and the smoothing circuit 12. The anode terminal of the ninth diode D9 is connected to the low potential side input terminal of the smoothing circuit 12, and the cathode terminal of the ninth diode D9 is connected to the high potential side input terminal of the smoothing circuit 12. The ninth diode D9 is a diode for bypassing the secondary rectifier circuit 11 and forming a current return path. In the fourth embodiment, the ninth diode D9 is indispensable in order to secure a current return path when the fifth switching element S5 is in the OFF state.

  In the fourth embodiment, the fifth switching element S5 includes a node Nf between the high potential side input terminal of the smoothing circuit 12 and the cathode terminal of the ninth diode D9, and a high potential side output terminal of the secondary side rectifier circuit 11. Connected between. The fifth switching element S5 adjusts the electric power extracted from the transformer T. The longer the ON period of the fifth switching element S5, the more electric power can be extracted from the transformer T.

  The driving device 22 according to the fourth embodiment is further omitted from the second gate buffer 23b, the second photocoupler 27b, and the second control logic circuit 26b as compared with the driving device 22 according to the first embodiment shown in FIG. It is the structure which was made.

  FIG. 9 is a diagram illustrating an example of a control method of the switching power supply device 10 of FIG. The control method on the primary side is the same as the control method shown in FIG. The transformer current i1 (i2) is full-wave rectified by the secondary side rectifier circuit 11. The control device 20 adaptively changes the duty ratio of the fifth switching element S5 according to the output voltage of the switching power supply device 10. In FIG. 9, the state in which the phase of the second switching element S2 and the third switching element S3 (or the first switching element S1 and the fourth switching element S4) and the phase of the fifth switching element S5 are the same. 5 The duty ratio of the switching element S5 is 100%. As the phase difference α between the rising phase of the second switching element S2 and the third switching element S3 and the rising phase of the fifth switching element S5 increases, the duty ratio of the fifth switching element S5 decreases. When the output voltage of the switching power supply device 10 increases, the control device 20 decreases the duty ratio of the fifth switching element S5 and decreases the amount of power extracted from the transformer T. Conversely, when the output voltage of the switching power supply device 10 decreases, the control device 20 increases the duty ratio of the fifth switching element S5 and increases the amount of power extracted from the transformer T.

  As described above, according to the fourth embodiment, the fifth switching element S5 is provided between the secondary side rectifier circuit 11 and the smoothing circuit 12, and the ninth diode D9 is provided between the input terminals of the smoothing circuit 12. The output adjustment in the secondary circuit and the processing of the circulating current can be realized with one switching element. Accordingly, the number of drive circuits for driving the secondary side switching elements can be reduced to one, the number of components of the drive device 22 can be reduced, and the cost can be reduced. Further, the circuit scale of the entire power conversion apparatus 100 can be reduced.

  When the first embodiment and the fourth embodiment are compared, the fourth embodiment can have one switching element on the secondary side, but the current stress flowing through one switching element increases. In applications where the current supplied to the load 30 is large, the fifth switching element S5 may generate heat, and a cooling system for preventing heat generation is required. In that case, the scale of the power converter 100 increases. Therefore, the power conversion device 100 according to Embodiment 4 is suitable for an application with a small load current. For example, when a general IGBT is adopted as the fifth switching element S5, the cooling system can be unnecessary or minimized if the current output from the switching power supply device 10 to the load 30 is 10 A or less.

  FIG. 10 is a diagram showing details of the secondary circuit of the switching power supply device 10 and the control device 20 of the power conversion device 100 according to Embodiment 5 of the present invention. The power conversion device 100 according to the fifth embodiment is different from the power conversion device 100 according to the fourth embodiment in the arrangement of the fifth switching element S5. In the fifth embodiment, the fifth switching element S5 includes a node Ng between the low potential side input terminal of the smoothing circuit 12 and the anode terminal of the ninth diode D9, and the low potential side output terminal of the secondary side rectifier circuit 11. Connected between.

  The secondary side fifth switching element S5 has both ends of the secondary winding N2 open when viewed from the primary side when all of the primary side first switching element S1 to the fourth switching element S4 are turned off. It is also arranged for the purpose of making it. Therefore, even if the fifth switching element S5 is connected between the low-potential side output terminal of the secondary rectifier circuit 11 and the low-potential side input terminal of the smoothing circuit 12, the same effect as in the fourth embodiment can be obtained.

  FIG. 11 is a diagram showing details of the secondary circuit of the switching power supply device 10 and the control device 20 of the power conversion device 100 according to Embodiment 6 of the present invention. In the power conversion device 100 according to the sixth embodiment, the secondary rectifier circuit of the power conversion device 100 according to the fourth embodiment is replaced from a full bridge type to a center tap type. The control method of switching power supply device 10 according to the sixth embodiment is the same as the control method of switching power supply device 10 according to the fourth embodiment shown in FIG.

  The secondary-side rectifier circuit 11 includes a normal center tap type rectifier circuit that does not include an active switch and uses two diodes. The anode terminal of the ninth diode D9 is connected to a node Ne between the center tap of the secondary winding N2 and the low potential side input terminal of the smoothing circuit 12, and the cathode terminal of the ninth diode D9 is connected to the smoothing circuit 12. To the high potential side input terminal. The fifth switching element S5 is connected between a node Nf between the high potential side input terminal of the smoothing circuit 12 and the cathode terminal of the ninth diode D9 and the high potential side input terminal of the smoothing circuit 12.

  As described above, even if the center tap type is adopted for the secondary side rectifier circuit 11 as in the sixth embodiment, by providing the fifth switching element S5 and the ninth diode D9, Output adjustment and circulating current processing can be realized with a single switching element. Therefore, the power conversion device 100 according to the sixth embodiment also has the same effect as the power conversion device 100 according to the fourth and fifth embodiments.

  The power conversion device 100 according to Embodiment 1-6 described above can be used for various applications. Hereinafter, examples of use in power storage systems, vehicles, and chargers will be given. In addition, it is also used for applications that require highly efficient power conversion and insulation, such as a data center power supply device.

  FIG. 12 is a diagram showing a configuration of a power storage system 400 in which the power conversion device 100 according to Embodiment 1-6 is used. A power storage system 400 illustrated in FIG. 12 includes a solar battery 200a, a storage battery 200b, a DC-DC converter 100a, a DC-DC converter 100b, and an inverter 300a. The storage battery 200b may be a stationary storage battery or a portable storage battery such as an in-vehicle battery. The DC power generated by the solar cell 200a is converted into DC power having a predetermined voltage by the DC-DC converter 100a. The DC power is converted into AC power by the inverter 300a and output to the system 500, or is converted into DC power of the storage voltage by the DC-DC converter 100b and stored in the storage battery 200b. The power conversion device 100 according to Embodiment 1-6 is used for at least one of the DC-DC converter 100a and the DC-DC converter 100b.

  Note that the storage battery 200b and the DC-DC converter 100b may be omitted. In this case, a solar power generation system without a power storage function is obtained. Further, the solar cell 200a and the DC-DC converter 100a may be omitted. In this case, the power storage system has no power generation function.

  FIG. 13 is a diagram showing a configuration of a vehicle 700 in which the power conversion device 100 according to Embodiment 1-6 is used. A vehicle 700 shown in FIG. 13 is a hybrid car (HV), a plug-in hybrid car (PHV), or an electric vehicle (EV) on which a traveling motor 600 is mounted. The motor 600 is not limited to a high-power motor that can run on its own, but may be a travel assist motor mounted on a mild hybrid car. Usually, an AC synchronous motor is used as the motor 600.

  A vehicle 700 shown in FIG. 13 includes a traveling battery 200c, an auxiliary battery 200d, a DC-DC converter 100c, a bidirectional DC-DC converter 150, an inverter 300b, and a motor 600. A storage battery such as a lithium ion battery or a nickel metal hydride battery can be used as the traveling battery 200c. During power running, bidirectional DC-DC converter 150 converts the DC power supplied from battery for traveling 200c into DC power of a predetermined voltage and outputs it to inverter 300b. Inverter 300 b converts the DC power supplied from bidirectional DC-DC converter 150 into AC power and supplies the AC power to motor 600. During regeneration, inverter 300 b converts AC power generated based on the deceleration energy into DC power and outputs the DC power to bidirectional DC-DC converter 150. The bidirectional DC-DC converter 150 converts the direct current power supplied from the inverter 300b into the direct current power of the battery voltage and charges the traveling battery 200c.

  As the auxiliary battery 200d, a 12V output lead battery is usually used. In the mild hybrid car, the traveling battery 200c is designed to output 48V, for example. The 12V system to which the auxiliary battery 200d is connected and the 48V system to which the traveling battery 200c is connected are connected via the DC-DC converter 100c. The DC-DC converter 100c boosts the voltage of the auxiliary battery 200d to the voltage of the traveling battery 200c. Thus, when the capacity of the traveling battery 200c is insufficient, power can be supplied to the motor 600 from the auxiliary battery 200d. For DC-DC converter 100c, power conversion device 100 according to Embodiment 1-6 is used.

  FIG. 14 is a diagram showing a configuration of a charger 800 in which the power conversion device 100 according to Embodiment 1-6 is used. A vehicle 700 shown in FIG. 14 is a vehicle in which a plug-in charging function is added to the vehicle shown in FIG. The charger 800 includes a rectifier circuit 810, a PFC circuit 820, and a DC-DC converter 100d. The rectifier circuit 810 rectifies the AC voltage supplied from the system 500. The PFC circuit 820 improves the power factor of the rectified power. The DC-DC converter 100d converts the input voltage from the PFC circuit 820 into a charging voltage. For DC-DC converter 100d, power conversion device 100 according to Embodiment 1-6 is used. As shown in FIG. 14, charger 800 may be a (rapid) charger installed outside the vehicle, or an in-vehicle charger mounted in vehicle 700.

  As described above, by using power conversion device 100 according to Embodiment 1-6 for the DC-DC converter used in power storage system 400, vehicle 700, or charger 800, low loss is achieved. A small power supply system can be constructed.

  The present invention has been described based on the embodiments. The embodiments are exemplifications, and it will be understood by those skilled in the art that various modifications can be made to combinations of the respective constituent elements and processing processes, and such modifications are within the scope of the present invention. .

  For example, a half-bridge type instead of a full-bridge type may be employed for the primary side inverter of the switching power supply device 10. In the first to third embodiments, the output adjustment is performed by shifting the phases of the secondary-side fifth switching element S5 and the sixth switching element S6. However, the fifth switching element S5 and the sixth switching element S6 are described. The output may be adjusted by adjusting the duty ratio.

  Moreover, you may comprise the 1st coil L1 and the 2nd coil L2 by the leakage inductance of the primary winding N1. Moreover, you may comprise the 1st capacitor | condenser C1-the 4th capacitor | condenser C4 with each parasitic capacitance of 1st switching element S1-4th switching element S4 instead of a snubber capacitor. Further, the first diode D1 to the fourth diode D4 may be constituted by the respective parasitic diodes of the first switching element S1 to the fourth switching element S4.

  The present invention can be used for DC-DC converters used in power storage systems, vehicles, and the like.

  DESCRIPTION OF SYMBOLS 100 Power converter device, 10 Switching power supply device, 20 Control apparatus, S1 1st switching element, S2 2nd switching element, S3 3rd switching element, S4 4th switching element, S5 5th switching element, S6 6th switching element, C1 first capacitor, C2 second capacitor, C3 third capacitor, C4 fourth capacitor, C5 fifth capacitor, C6 sixth capacitor, C7 seventh capacitor, D1 first diode, D2 second diode, D3 third diode, D4 4th diode, D5 5th diode, D6 6th diode, D7 7th diode, D8 8th diode, D9 9th diode, L1 1st coil, L2 2nd coil, L3 3rd coil, L4 1st Coil, L5 fifth coil, L6 sixth coil, T transformer, N1 primary winding, N2 secondary winding, E DC power supply, 11 secondary side rectifier circuit, 12 smoothing circuit, 15 output voltage detection circuit, 21 CPU, 22 driving device, 23a first gate buffer, 23b second gate buffer, 24a first high side DC-DC converter, 24b second high side DC-DC converter, 25a first low side DC-DC converter, 25b second low side DC DC converter, 26a first control logic circuit, 26b second control logic circuit, 27a first photocoupler, 27b second photocoupler, 30 load, 100a, 100b, 100c, 100d DC-DC converter, 150 bidirectional DC- DC converter 200a solar cell, 200b accumulators, 200c traveling battery, 200d auxiliary battery, 300a, 300b inverter, 400 power storage system, 500 lines, 600 motor, 700 vehicle, 800 charger, 810 rectifier circuit, 820 PFC circuit.

Claims (4)

A bridge circuit that converts DC voltage to AC voltage;
A transformer including a primary winding connected to the bridge circuit, and a secondary winding electromagnetically coupled to the primary winding;
A rectifier circuit for rectifying an AC voltage input from the secondary winding;
A smoothing circuit for smoothing the voltage rectified by the rectifier circuit;
A switching element inserted in a current path between the rectifier circuit and the smoothing circuit;
A diode having an anode terminal connected to the low potential side input terminal of the smoothing circuit and a cathode terminal connected to the high potential side input terminal of the smoothing circuit;
A switching power supply device comprising:   The switching element is connected between a node between a high potential side input terminal of the smoothing circuit and a cathode terminal of the diode and a high potential side output terminal of the rectifier circuit. The switching power supply device according to 1. The bridge circuit is
A primary-side forward switching element inserted in a path for supplying a forward current to the primary winding;
A primary-side reverse switching element inserted in a path for supplying a reverse current to the primary winding,
The rectifier circuit is composed of a diode bridge circuit,
The primary-side forward switching element and the primary-side reverse switching element operate with a fixed duty ratio and a fixed phase,
2. The switching element on the secondary side inserted in a current path between the rectifier circuit and the smoothing circuit operates with a duty ratio that varies according to an output voltage of the switching power supply device. Or the switching power supply device according to 2; A switching power supply device according to any one of claims 1 to 3,
A driving device for driving the switching element;
A power conversion device comprising:

Images