JP2013225998A - Rectifier circuit and motor drive device using the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a more highly efficient rectifier circuit with a small number of switching operations.SOLUTION: The rectifier circuit includes: an AC reactor connected to an AC power supply; a rectifier whose AC side is connected to the AC power supply through the AC reactor and whose DC side is connected to a DC load; a plurality of smoothing capacitors in series connection disposed between the DC side of the rectifier and the DC load; and a bidirectional switch circuit disposed between the AC side of the rectifier and the connection point of the plurality of smoothing capacitors in series connection. Each charge voltage of the plurality of smoothing capacitors in series connection is not balanced.

Description

  The present invention relates to a motor driving device that drives a permanent magnet synchronous motor at a variable speed and a device using the motor driving device, and more particularly to a rectifying circuit that performs a rectifying operation of an AC power source and a boosting operation of a DC voltage.

Permanent magnet synchronous motors (hereinafter referred to as “motors” or “PM motors”) have higher efficiency than induction motors, and thus have a wide range of applications from home appliances to industrial equipment or electric vehicle fields. .
In addition, the above devices are required to improve the efficiency in the normal operation area (high efficiency in the low and medium speed range) in accordance with the movement of global warming prevention and energy saving, but to improve the feeling of use of the device High output (expanding driving range in high speed range) is also required at the same time.
For example, in the case of air conditioners for home appliances, improvement of APF (Annual Performance Factor), which is an energy-saving index, and low-temperature heating capacity (heating capacity when the outside temperature is 2 ° C), which is an index of high output It is required to improve both.
For example, as a means for improving the efficiency (particularly in the low and medium speed range) by the motor drive device, there is a low-speed design of the motor by increasing the magnet amount and winding.
As one of means for driving a low-speed designed motor at high speed, there is a system for boosting a DC voltage.

Further, as a method for boosting a DC voltage by using an operation of a voltage doubler rectifier circuit of an AC power source, there are methods described in Patent Documents 1 and 2, for example. According to these, since the step-up operation can be performed with a relatively low-speed switching operation as compared with the method using the step-up chopper circuit when boosting the DC voltage, the circuit loss can be reduced.
In Patent Document 1, a rectifier connected to an AC power source via a reactor, a capacitor connected in series to the output terminal of the rectifier, and a connection between the capacitor connected to one input terminal of the rectifier and the series In a circuit configuration comprising: first switching means inserted at a point; and second switching means connected at the other input terminal of the rectifier to the connection point between the capacitors connected in series. There is disclosed a technique for converting the input current of the rectifier into a sine wave by PWM (Pulse Width Modulation) control of the first switching means and the second switching means at a low frequency of 1 to 5 kHz in a balanced manner.
Patent Document 2 also discloses a technique for boosting a DC voltage by alternately and repeatedly operating the first switching means and the second switching means in a cycle shorter than the power supply cycle with a circuit configuration similar to that of Patent Literature 1. Is disclosed.

JP 2010-68552 A JP 2005-110491 A

However, when the motor is designed at a low speed by increasing the amount of magnets and windings, the induced voltage generated in the high speed region increases. Therefore, since high-speed driving becomes difficult, there is a problem that the driving range is limited and the efficiency as a motor is greatly reduced.
In addition, in a method of boosting a DC voltage, a method of performing a high-speed switching operation by adding a boost chopper circuit to a rectifier circuit has a problem that circuit loss increases because the DC voltage is boosted.
In Patent Document 1 and Patent Document 2, as described above, the first switching means and the second switching means are alternately switched to perform a sine wave of the power supply current or a DC voltage boosting operation. However, it is necessary to switch the two switching means at a high frequency of 1 kHz or more, and there is a problem that the switching loss is large.

  Therefore, the present invention solves such problems, and an object of the present invention is to provide a rectifier circuit capable of further increasing efficiency with a small switching operation.

In order to solve the above-described problems and achieve the object of the present invention, the present invention is configured as follows.
That is, the rectifier circuit according to the first aspect of the present invention includes an AC reactor connected to an AC power source, an AC side connected to the AC power source via the AC reactor, and a DC side connected to a DC load, Bidirectionally provided between a plurality of series-connected smoothing capacitors provided between the DC side of the rectifier and the DC load, and a connection point between the AC side of the rectifier and the plurality of series-connected smoothing capacitors. A switching circuit, wherein the charging voltages of the plurality of smoothing capacitors connected in series are unbalanced.
The rectifier circuit according to the second aspect of the invention includes an AC reactor connected to an AC power source, an AC side connected to the AC power source via the AC reactor, a DC side connected to a DC load, A first smoothing capacitor and a second smoothing capacitor connected in series between a DC side of a rectifier and the DC load, the first smoothing capacitor connected in series with the AC side of the rectifier, and the A bidirectional switch circuit provided between a connection point of the second smoothing capacitor, a voltage at both ends of the second smoothing capacitor being lower than a half of a DC voltage, and the first smoothing capacitor Is characterized in that the voltage at both ends is higher than half the DC voltage.
Other means will be described in the embodiment for carrying out the invention.

  According to the present invention, it is possible to provide a rectifier circuit capable of further increasing efficiency with a small switching operation.

It is a figure which shows the circuit structure of the rectifier of 1st Embodiment of this invention, and the connection relation of alternating current power supply and load. It is a time chart which shows the power supply voltage of 1st Embodiment of this invention, an input current, a smoothing capacitor voltage, and the gate signal waveform of the switching element of a bidirectional | two-way switch circuit. It is a figure which shows the simulation result when setting to the predetermined | prescribed load of 1st Embodiment of this invention, a power supply voltage, a smoothing capacitor voltage (high voltage side), and a smoothing capacitor voltage (low voltage side). It is a figure which shows the simulation result when setting to another predetermined load of 1st Embodiment of this invention, a power supply voltage, a smoothing capacitor voltage (high voltage side), and a smoothing capacitor voltage (low voltage side). It is a figure which shows another switching method of the bidirectional | two-way switch circuit in 2nd Embodiment of this invention, (a) is the system which PWM-controlled the gate signal Gsw2, (b) is based on the switching method of 1st Embodiment. (C) is a method in which the short-circuit mode is used in combination with the switching method (a) in the short-circuit mode in which the AC power supply is short-circuited through the AC reactor. It is a figure which shows the simulation result at the time of making it operate | move by the switching system (operating condition is the same as FIG. 3) of the bidirectional | two-way switch circuit in 2nd Embodiment of this invention of FIG. It is a figure which shows the simulation result at the time of making it operate | move by the switching system (operating conditions are the same as FIG. 3) of the bidirectional | two-way switch circuit in 2nd Embodiment of this invention. It is a figure which shows the simulation result at the time of making it operate | move by the switching system (operating conditions are the same as FIG. 3) of the bidirectional | two-way switch circuit in 2nd Embodiment of this invention. It is a figure which shows the simulation result at the time of making it operate | move by the switching system (operating conditions are the same as FIG. 4) of the bidirectional | two-way switch circuit in 2nd Embodiment of this invention. It is a figure which shows the simulation result at the time of making it operate | move by the switching system (operating conditions are the same as FIG. 4) of the bidirectional | two-way switch circuit in 2nd Embodiment of this invention of FIG. It is a figure which shows the whole structure of the motor drive device of the 3rd Embodiment of this invention. It is a figure explaining operation | movement in the case of controlling a DC voltage according to the rotation speed of a motor as a 1st example in the motor drive device of the 3rd Embodiment of this invention. It is a figure explaining operation | movement in the case of switching a full wave rectification voltage and a maximum voltage with the magnitude | size of the load of a motor as a 2nd example in the motor drive device of the 3rd Embodiment of this invention.

  Hereinafter, embodiments for carrying out the present invention will be described with reference to the drawings.

(First embodiment / rectifier circuit)
A first embodiment of a rectifier circuit according to the present invention will be described with reference to FIGS. First, the basic circuit configuration and basic operation will be described with reference to FIGS.

<Circuit configuration>
FIG. 1 is a diagram illustrating a circuit configuration of the rectifier circuit 10 according to the first embodiment and a connection relationship between the AC power source 1 and the load 5.
In FIG. 1, a rectifier circuit 10 according to the first embodiment includes an AC reactor 2, a rectifier 3, a smoothing capacitor group 4, a bidirectional switch circuit 6, and a controller 9.

The AC power output from the AC power source 1 is directly connected to one end of the AC power source 1 and the other end of the AC power is connected to the AC rectifier 3 of the rectifier 3 and the rectifier 61 provided in the bidirectional switch circuit 6 (2 Terminal).
The rectifier 61 has a full-wave rectification function with diodes 611, 612, 613, and 614. The AC side terminal of the rectifier 61 includes a first terminal that is a connection point between the anode of the diode 611 and the cathode of the diode 613, and a second terminal that is a connection point between the anode of the diode 612 and the cathode of the diode 614. .
The DC side terminal of the rectifier 61 includes a third terminal that is a connection point between the cathode of the diode 611 and the cathode of the diode 612, and a fourth terminal that is a connection point between the anode of the diode 613 and the anode of the diode 614. .
Similarly to the rectifier 61, the rectifier 3 is composed of four diodes, has a full-wave rectification function, and includes an AC side terminal and a DC side terminal.

The smoothing capacitor group 4 includes a smoothing capacitor 41 (first smoothing capacitor), a smoothing capacitor 42 (second smoothing capacitor), and a smoothing capacitor 43 (third smoothing capacitor) connected in series.
The smoothing capacitor 43 is connected in parallel to both ends of the smoothing capacitor 41 and the smoothing capacitor 42 connected in series.
The DC side terminal (two terminals) of the rectifier 3 is connected to both ends of the smoothing capacitor 41 and the smoothing capacitor 42 connected in series and both ends of the smoothing capacitor 43.
The smoothing capacitor group 4 is connected to a load (DC load) 5.
The smoothing capacitors 43 connected in parallel are connected to suppress pulsation of the DC voltage. The smoothing capacitor 41 and the smoothing capacitor 42 are related to the function of the boosting operation of the present invention described later.

The bidirectional switch circuit 6 includes a rectifier 61 and switching elements 62 and 63.
The switching element 62 and the switching element 63 are connected in series and connected to the DC side terminal (two terminals) of the rectifier 61.
A connection point between the switching element 62 and the switching element 63 is connected to a connection point between the smoothing capacitor 41 and the smoothing capacitor 42 in the smoothing capacitor group 4.
The switching element 62 and the switching element 63 are ON / OFF controlled by the controller 9. Details of this control method will be described later.

<Control Method of Bidirectional Switch Circuit 6 and Operation of Rectifier Circuit 10>
Next, the control method of the bidirectional switch circuit 6 and the operation of the rectifier circuit 10 will be described.

<Outline of basic concept and control method>
In the first embodiment of the present invention, harmonic current suppression and DC voltage boosting are made compatible with a small number of switching operations.
In the process of boosting the DC voltage using the two smoothing capacitors 41, 42, there is a process of charging the two smoothing capacitors 41, 42. The difference between the power supply voltage (instantaneous value) and the charging voltage of the smoothing capacitors 41, 42 Therefore, the boosting efficiency and the generation state of the harmonic current differ depending on the charging method.
Therefore, in the region where the power supply voltage (instantaneous value of the sine wave) is small, the smoothing capacitor 42 charged to the low-voltage charging voltage Vc2 is charged. The bidirectional switch circuit 6 is operated so as to charge the smoothing capacitor 41 charged to the high voltage Vc1.

More specifically, the bidirectional switch circuit 6 is configured so that the series-connected smoothing capacitors 42 have a voltage lower than half of the DC voltage, and the series-connected smoothing capacitors 41 have a voltage higher than half of the DC voltage. (So that the charging voltage of the smoothing capacitor is unbalanced).
By this method, charging is performed in a state where there is a sufficient potential difference, so that charging efficiency is improved and generation of harmonics is reduced because charging is performed according to the state of the input waveform (sine wave) of the power supply voltage. The

The bidirectional switch circuit 6 includes a short-circuit mode in which the AC power supply 1 is short-circuited via the AC reactor 2, and a voltage doubler rectification mode in which any one of the smoothing capacitors 41 and 42 connected in series is charged in series. A full-wave rectification charging mode in which both of the connected smoothing capacitors 41 and 42 are charged.
The DC voltage is boosted using the short-circuit mode, the voltage doubler charging mode, and the full-wave rectification charging mode.
With the above configuration, the harmonic component of the input current can be suppressed even when the number of switching operations of the bidirectional switch circuit is reduced, and the voltage can be boosted to an arbitrary DC voltage.
That is, the harmonic current is suppressed and the DC voltage is boosted with a small number of times of switching.

<About control method>
Next, a more specific operation of the bidirectional switch circuit 6 will be described.
FIG. 2 is a time chart showing waveforms of the power supply voltage Vs, the input current Is, the smoothing capacitor charging voltages Vc0, Vc1, and Vc2 and the gate signals Gsw1 and Gsw2 of the switching elements of the bidirectional switch circuit.
In FIG. 2, the horizontal axis represents the transition of time, which shows one cycle of the power supply voltage Vs (the positive voltage period is arranged in the vicinity of the center), and the sine waveform of the power supply voltage Vs increases from a low voltage to a high voltage. A time zone in which the voltage shifts to voltage is referred to as region I, a time zone in which the sine waveform is relatively high is referred to as region II, and a time zone in which the sine waveform shifts from a high voltage to a low voltage is referred to as region III.
Further, the symbols Vs, Is, Vc0, Vc1, Vc2, Gsw1, and Gsw2 correspond to the symbols in FIG.

The smoothing capacitor 42 (FIG. 1) has a charging voltage Vc2 (for example, 50V) lower than half the DC voltage, and the smoothing capacitor 41 (FIG. 1) has a charging voltage Vc1 (for example 240V) higher than half the DC voltage. It is charged.
In the region I where the power supply voltage (sine wave) Vs is rising (low voltage), the switching element 62 (FIG. 1) is turned on by the control pulse P22 of the gate signal Gsw2, and is relatively low with respect to the smoothing capacitor 42. Charging is performed with voltage. At this time, since the charging voltage Vc2 is a low voltage, charging is possible.
Note that the magnitude of the voltage Vc2 charged in the smoothing capacitor 42 can be controlled by the pulse width and timing of the control pulse P22.

Further, in the region II where the power supply voltage (sine wave) Vs is high, the switching element 63 (FIG. 1) is turned on by the control pulse P11 of the gate signal Gsw1, and the smoothing capacitor 41 has a relatively high voltage. Charging is performed.
Note that the magnitude of the voltage Vc1 charged in the smoothing capacitor 41 can be controlled by the pulse width and timing of the control pulse P11.

In the region III where the power supply voltage (sine wave) Vs falls, the switching element 62 (FIG. 1) is turned on by the control pulse P23 of the gate signal Gsw2.
When the power supply voltage rises (region I), the ON time of the switching element 62 is relatively long (P22), and when the power supply voltage falls (region III), the ON time of the switching element is relatively short (P23). ) Set. With this setting, the input current Is has a waveform that matches the power supply phase, and has a waveform that is relatively close to a sine wave, thereby reducing harmonic components.

The control pulse P22 of the gate signal Gsw2 and the control pulse P11 of the gate signal Gsw1 have a positive half cycle of the power supply voltage (sine wave) Vs, but in the negative half cycle, the control pulse P24 of the gate signal Gsw2 The smoothing capacitor 42 and the smoothing capacitor 41 are charged by the control pulse P12 of P21 and the gate signal Gsw1. Since the power supply voltage (sine wave) Vs is full-wave rectified by the rectifiers 3 and 61, the same operation is performed whether the power supply voltage (sine wave) Vs is a positive half cycle or a negative half cycle.
Note that the pulse widths and insertion timings of the control pulses P21, P22, P23, and P24 of the gate signal Gsw2 and the control pulses P11 and P12 of the gate signal Gsw1 are merely examples. Actually, the pulse width and timing according to the actual situation are selected.

As described above, in the present embodiment, the charging voltage of the smoothing capacitors connected in series is controlled to be unbalanced, and in a region where the power supply voltage (instantaneous value) is small, the smoothing capacitor whose charging voltage is low is used. In a region where the power supply voltage (instantaneous value) is large so as to be charged, the bidirectional switch circuit is operated so as to charge the smoothing capacitor having a high charging voltage. As a result, even if the number of times of switching is reduced, it is possible to achieve both suppression of harmonic current and boosting of DC voltage.
FIG. 2 shows a voltage doubler rectification mode in which any one of the smoothing capacitors 42 and 43 connected in series is charged.
In addition, since the AC frequency of the power supply voltage Vs is 50 Hz or 60 Hz in Japan, the gate signals Gsw1 and Gsw2 of the bidirectional switch circuit 6 are approximately several tens Hz to several hundreds Hz, which is a conventional example. Compared to the low frequency.

<Simulation results>
Next, FIG. 3 and FIG. 4 show examples of simulation results (during the switching operation shown in FIG. 2).
FIG. 3 is a diagram showing simulation results when the load 2900 W, the power supply voltage 200 V, the smoothing capacitor voltage (high voltage side) Vc1 = 240 V, and the smoothing capacitor voltage (low voltage side) Vc2 = 50 V are set.
FIG. 4 is a diagram showing a simulation result when a load 2900 W, a power supply voltage 200 V, a smoothing capacitor voltage (high voltage side) Vc1 = 300 V, and a smoothing capacitor voltage (low voltage side) Vc2 = 50 V are set.
3 and 4, the horizontal axis indicates the transition of time, and in the vertical direction, from the top, the waveforms of voltage (Vs, Vc0, Vc1, Vc2), input current Is, and gate signals Gsw1, Gsw2.
In both the results of FIGS. 3 and 4, the waveform of the input current Is clears the Japanese harmonic suppression guideline (not shown).

1 to 4, the charging voltages Vc1 and Vc2 of the smoothing capacitors 41 and 42 are such that Vc1 on the high voltage side is larger than half of the desired DC voltage value Vc0 and Vc2 on the low voltage side is desired. What is necessary is just to set so that it may be a value smaller than half of direct-current voltage value Vc0, and a total value may become desired direct-current voltage Vc0.
In addition, as described above, when the purpose is to suppress harmonics of the input current waveform, it is desirable to set the set voltage (Vc2) on the low voltage side as small as possible.
However, in view of the capacity and load of the smoothing capacitor, it is necessary to set the charging voltage Vc2 of the smoothing capacitor on the low voltage side to a value that does not become zero within the power supply cycle.
Therefore, in the simulations of FIG. 3 and FIG. 4, it is desirable that the low-voltage setting voltage (Vc2) is, for example, about 50V.

(Second Embodiment / Switching Method)
Next, although the circuit configuration of FIG. 1 is used as it is, another switching method other than the method described in the first embodiment of the bidirectional switch circuit 6 will be described as a second embodiment.

<Another switching method>
5A and 5B are diagrams illustrating another switching method of the bidirectional switch circuit 6, in which FIG. 5A is a method in which the gate signal Gsw2 is PWM-controlled, and FIG. 5B is an AC power source based on the switching method of the first embodiment. A method in which a short-circuit mode in which 1 is short-circuited via an AC reactor 2 is used in combination, and (c) is a method in which the short-circuit mode is used in combination with the switching method in (a).

<< PWM controlled system >>
5A, the control pulses P11 and P12 of the gate signal Gsw1 of the switching element 63 (FIG. 1) by the bidirectional switch circuit 6 (FIG. 1) are the same as the control pulses P11 and P12 in FIG.
However, the gate signal Gsw2 of the switching element 62 is P21W and P22W of the PWM controlled method. By making the gate signal Gsw2 finely controlled by the PWM signal, harmonics in the rectifier circuit 10 can be suppressed.
Incidentally, when the control pulses P11 and P12 of the gate signal Gsw1 are generated, the PWM signal of the gate signal Gsw2 is stopped.
In the conventional chopper, about 16 kHz is used, but the PWM signal of the gate signal Gsw2 in FIG. 5A has a carrier of about 1 kHz. Although increased, the power consumption is lower than that of the conventional chopper method.
Details of the simulation result in the case of FIG. 5A will be described later with reference to FIG.

<Method using short-circuit mode together>
Next, in FIG. 5B, control pulses PS1 and PS2 for entering the short circuit mode are newly added to the gate signal Gsw1.
The control pulses P11 and P12 of the gate signal Gsw1 and the control pulses P21, P22, P23 and P24 of the gate signal Gsw2 are the same as those in FIG.
When the control pulse PS1 for entering the short-circuit mode is generated in the gate signal Gsw1, the control pulse P22 is also generated in the gate signal Gsw2, so that both the switching element 62 and the switching element 63 are turned on. At this time, the power supply voltage Vs of the AC power supply 1 is applied to the AC reactor 2, and the AC reactor 2 accumulates electric power.
By entering this short-circuit mode, the waveform of the input current Is becomes smooth. That is, the harmonic component of the input current Is is suppressed.

As shown in FIG. 5B, the control pulses PS1 and PS2 that enter the short-circuit mode are performed at a timing when the sine wave (AC voltage waveform) that is the voltage waveform of the power supply voltage Vs crosses 0 (0 volt). Is more desirable.
This is because, for the switching elements 62 and 63, zero cross switching leads to suppression of noise and harmonics.
Details of the simulation result in the case of FIG. 5B will be described later with reference to FIGS.

<Method using both PWM control and short-circuit mode>
Next, in FIG. 5C, control pulses PS1 and PS2 for entering the short circuit mode are newly added to the gate signal Gsw1 in addition to the control pulses P11 and P12 of FIG.
Further, the control pulse P21w and the control pulse P22w of the gate signal Gsw2 in FIG. 5A are (P31w, P22S, P32w) and (P33w, P24S, P34w) in FIG. 5C, respectively.
Since P22S and P24S are generated in the gate signal Gsw2 with respect to the control pulses PS1 and PS2 in the short circuit mode in the gate signal Gsw1, both the switching element 62 and the switching element 63 are turned on.

Therefore, when the voltage of the AC power source 1 is applied to the AC reactor 2, and the AC reactor 2 has the effect of accumulating electric power and the effect of PWM control, the peak value of the input current Is can be simultaneously reduced. The waveform becomes even smoother. That is, the harmonic component of the input current Is is suppressed.
As shown in FIG. 5C, the control pulses PS1, PS2, and P22S, P24S that are in the short-circuit mode have a sine wave (AC voltage waveform) that is a voltage waveform of the power supply voltage Vs set to 0 (0 volt). It is more desirable to carry out at the timing of crossing.
This is because, for the switching elements 62 and 63, zero cross switching leads to suppression of noise and harmonics.
Details of the simulation result in the case of FIG. 5C will be described later with reference to FIGS.

<Simulation results>
Next, FIGS. 6 to 10 show simulation results when the switching system shown in FIGS. 5A, 5B, and 5C is operated.
6 to 10, the horizontal axis indicates the flow of time, and in the vertical direction, the waveforms of voltage (Vs, Vc0, Vc1, Vc2), input current Is, and gate signals Gsw1 and Gsw2 are shown from the top. ing.

<< FIG. 6: Simulation Results of the Switching System in FIG. 5 (a) / Conditions in FIG. 3 >>
FIG. 6 shows the switching method of FIG. 5A (operating conditions are the same as in FIG. 3, load 2900 W, power supply voltage 200 V, smoothing capacitor voltage (high voltage side) Vc 1: 240 V, smoothing capacitor voltage (low voltage side) Vc 2: It is a figure which shows the simulation result at the time of making it operate | move by setting to 50V.
Compared with the current waveforms of the respective input currents Is in FIG. 6 and FIG. 3, the current waveform in FIG. 6 is smooth, and it is possible to suppress harmonic components, particularly to reduce low-order harmonic components.

<< FIG. 7: Simulation Results of the Switching Method in FIG. 5 (b) / Conditions in FIG.
FIG. 7 is a diagram illustrating a simulation result when the switching method of FIG. 5B (operating conditions are the same as those in FIG. 3).
Compared with the current waveforms of the respective input currents Is in FIG. 7 and FIG. 3, the short-circuit mode is entered, so that the rise of the input current Is is quickened, and the peak value of the input current Is can be slightly reduced. Accordingly, the generation of harmonic components is suppressed.

<< FIG. 8: Simulation Results of the Switching Method in FIG. 5 (c) / Conditions in FIG.
FIG. 8 is a diagram illustrating a simulation result when the switching method of FIG. 5C (operating conditions are the same as those in FIG. 3).
Compared with the current waveforms of the respective input currents Is in FIG. 8 and FIG. 7, the short circuit mode is entered, so that the peak value of the input current Is can be reduced and the waveform becomes smoother. That is, the harmonic component of the input current Is is suppressed.

<< FIG. 9: Simulation Results of Switching Method in FIG. 5B / Conditions in FIG. 4 >>
FIG. 9 shows the switching method of FIG. 5B (operating conditions are the same as in FIG. 4, load 2900 W, power supply voltage 200 V, smoothing capacitor voltage (high voltage side) Vc 1: 310 V, smoothing capacitor voltage (low voltage side) Vc 2: It is a figure which shows the simulation result at the time of making it operate | move by setting to 50V.
Comparing FIG. 9 and FIG. 4, since the short circuit mode is entered, the DC voltage (Vc0) can be further boosted.

<< FIG. 10: Simulation Results of the Switching Method in FIG. 5 (c) / Conditions in FIG.
FIG. 10 is a diagram illustrating a simulation result when the switching method of FIG. 5C is used (the operation condition is the same as that of FIG. 4).
Comparing FIG. 10 and FIG. 9, since the PWM operation is entered, the current waveform of the input current Is approaches a sine wave shape. The DC voltage (Vc0) can be boosted.

As described above, the current waveform can be improved and the DC voltage can be freely controlled by changing the switching method of the switching element of the bidirectional switch circuit 6.
Therefore, although not shown, the switching method can be changed according to the state of the load connected to the rectifier circuit (input current, DC voltage, etc.).
When the load is light (for example, when it is not necessary to boost the DC voltage), use only the short-circuit mode in which the two switching elements are simultaneously turned on, and operate as a full-wave rectifier circuit + current harmonic suppression mode. Is also possible.

Third Embodiment Motor Drive Device
Next, as a third embodiment, a case where the rectifier circuits of the first and second embodiments are applied as a rectifier circuit of a motor drive device will be described.

FIG. 11 is a diagram illustrating an overall configuration of a motor drive device according to a third embodiment of the present invention. The motor drive device includes a rectifier circuit 11 and an inverter device (inverter circuit) 7.
In FIG. 11, the AC power source 1, AC reactor 2, rectifier 3, smoothing capacitor group 4, and controller 9 have the same configuration, function, and operation as those in FIG.
11 differs from FIG. 1 in the circuit configuration of the bidirectional switch circuit 60 and the configuration in which the PM motor 8 and the inverter device 7 that drives the PM motor 8 are connected as a load of the rectifier circuit 11. .

The bidirectional switch circuit 60 includes a rectifier 64 having a diode 641 and a diode 642, a diode 65, and switching elements 62 and 63.
A function equivalent to that of the rectifier 61 of FIG. 1 is achieved by the rectifier 64 and the diode 65.
Therefore, the bidirectional switch circuit 60 of FIG. 11 operates in substantially the same manner as the bidirectional switch circuit 6 described in FIG.
The circuit configuration of the rectifier circuit 11 shown in FIG. 11 has an advantage that the number of diode components can be reduced by one as compared with the rectifier circuit 10 described in FIG.

Further, the inverter device 7 converts input DC power (DC voltage) into three-phase AC power having an arbitrary frequency and AC voltage, and outputs it. A PM motor (Permanent Magnet motor, permanent magnet motor, A controller and a control method capable of controlling the rotational speed of a permanent magnet synchronous motor (motor) 8 are provided.
Since the motor drive device of the third embodiment is characterized by using the rectifier circuit 11 (or rectifier circuit 10) described above, detailed description of the inverter device 7 is omitted.
Here, the operation of the rectifier circuit 11 (10) will be described using FIG. 12 and FIG. 13 from the relationship between the DC voltage boosting method and the overall efficiency when considered as a motor drive device.

<How to boost DC voltage and overall efficiency of motor drive>
12 and 13, the horizontal axis represents the motor speed or load (load state), and the vertical axis represents the DC motor and the overall efficiency of the motor driving device including the motor (three-phase AC power / rectifier 11 output by the inverter device 7. The change image of the AC power input to (1) is shown as a first example and a second example, respectively.
In FIGS. 12 and 13, the overall efficiency is obtained by changing the DC voltage (bidirectional switch circuit operating state) when the motor is driven with a full-wave rectified voltage (bidirectional switch circuit stopped state). This is indicated by the case where the motor is driven (dotted line).

<< First Example: Method for Controlling DC Voltage According to Motor Rotation Speed >>
FIG. 12 is a diagram for explaining the operation when the DC voltage is controlled in accordance with the rotational speed of the motor as a first example. The horizontal axis represents the motor rotation speed (rotation speed per unit time), and the vertical direction represents the efficiency of the rectifier circuit and the relative characteristic value of the rectified DC voltage of the rectifier circuit.
In FIG. 12, at the time of low rotation, the operation of the bidirectional switch circuit is stopped (switch circuit stop area) and used as a full-wave rectifier circuit.
At this time, the DC voltage becomes a full-wave rectified voltage (characteristic line 123A), and the motor can be driven with a low DC voltage, so that the overall efficiency is improved (the loss of the bidirectional switch circuit does not occur: characteristic line 120).
However, when the rotation speed is high, driving cannot be performed with a low DC voltage, and it is necessary to drive using field-weakening control (control that causes a large amount of reactive current to flow), resulting in a decrease in overall efficiency (characteristic line 122).

Therefore, by controlling the desired DC voltage (characteristic line 123B) by operating the bidirectional switch circuit 60 (FIG. 11) (switch circuit operation range, conduction time variable range), the motor does not use field-weakening control. Since driving is possible, efficient driving is possible (characteristic line 121).
Note that, when the rotational speed further increases, the maximum voltage (characteristic line 123 </ b> C) boosted by the combination of the bidirectional switch circuit 60 and the smoothing capacitor group 4 (FIG. 11) is reached.
Here, the switching method of the bidirectional switch circuit 60 is possible by combining the methods in the first and second embodiments.
Further, the rectifier circuit shown in the third embodiment and the first embodiment (the bidirectional switch circuit 60 is approximately 1 kHz even during PWM control, and even lower frequency when PWM control is not performed) is as fast as a boost chopper circuit. Since the voltage is not boosted by the switching operation (approximately 16 kHz), switching loss is small, and the overall efficiency can be improved as compared with the system using the boost chopper circuit.

<< Second example: Method of switching full-wave rectified voltage and maximum voltage depending on motor load >>
FIG. 13 is a diagram for explaining the operation when the full-wave rectified voltage and the maximum voltage are switched according to the load of the motor. That is, FIG. 13 does not change the DC voltage as shown in FIG. 12, and performs full-wave rectification operation (bidirectional switch circuit operation stop) and maximum voltage operation (bidirectional switch circuit operation) under a predetermined load condition. An example of a switching method is shown.
The horizontal axis represents the load value (load state), and the vertical direction represents the efficiency of the rectifier circuit and the relative characteristic value of the rectified DC voltage of the rectifier circuit.
In FIG. 13, since the load is light and the switch circuit stop region (bidirectional switch circuit operation stop) is performed, the full-wave rectification operation is performed. Therefore, the full-wave rectification voltage is substantially constant as shown by the characteristic line 133A. Like the characteristic line 130, the efficiency is relatively high.
However, when the load increases, the rectifier circuit 11 (FIG. 11) cannot supply an appropriate DC voltage to the inverter device 7 (FIG. 11), and the characteristic line 130 indicating the overall efficiency decreases.
If the load becomes heavier, the overall efficiency will decrease significantly as indicated by the characteristic line 132 when the switch circuit is stopped.

Therefore, when the predetermined load is reached, the bidirectional switch circuit 60 (FIG. 11) is operated (switch circuit operation area). Then, the rectified DC voltage of the rectifier circuit 11 becomes the maximum voltage indicated by the characteristic line 133C.
At this time, since a sufficient DC voltage is supplied to the inverter device 7, the overall efficiency is much higher than the efficiency of the characteristic line 132 when the switch circuit is stopped, as shown by the characteristic line 131 (switch circuit operation). Can be improved.
Since this method does not finely control the DC voltage, the control configuration can be simplified. Therefore, it is preferable to apply to an application (for example, a refrigerator, etc.) that is usually driven at a light load and needs to be activated at a high load.

  As described above, as shown in the first and second examples, when the rectifier circuit of the present invention is applied to a motor driving device, the operation of the bidirectional switch circuit 60 is stopped when the motor 8 is lightly loaded, and bidirectionally when the motor 8 is heavily loaded. To realize a motor drive device that can achieve both high efficiency and wide range drive by changing the operation according to the load state of the motor 8 that causes the switch circuit 60 to operate, and an apparatus using the same. Is possible.

(Other embodiments)
The present invention is not limited to the embodiment described above. Here are some examples:

<< Smoothing capacitors connected in series in three or more stages >>
In FIG. 1 and FIG. 11, the smoothing capacitor 41 and the smoothing capacitor 42 that perform the step-up operation in the smoothing capacitor group 4 are configured by two capacitors in series.
However, it is not limited to two capacitors. For example, when three (first, second, and third) capacitors (smoothing capacitors) are connected in series and the voltage of the power supply voltage Vs of the AC power supply 1 (FIG. 1) is relatively low, the first 3 is charged (Vc3). Then, the second capacitor is charged (Vc2) as when the voltage of the waveform of the power supply voltage Vs is a relatively medium voltage. Further, when the voltage of the waveform of the power supply voltage Vs is a relatively high voltage, the first capacitor is charged (Vc1).
As described above, the bidirectional switch circuit (corresponding to the bidirectional switch circuit 6 in FIG. 1) is configured so as to charge three (first, second, and third) capacitors. At this time, the rectifier in the bidirectional switch circuit may be the same as that of the rectifier 61 in FIG. 1, but at least three switching elements are required, and the control method of the controller 9 (FIG. 1) is the same as that of the charging. Design for proper control.

At this time, the DC voltage output from the smoothing capacitor group 4 is (Vc1 + Vc2 + Vc3), and the boosting efficiency is improved. The charging voltages Vc1, Vc2, and Vc3 of the three capacitors are set to be unbalanced so as to be different from each other.
Also, when charging the first, second, and third capacitors, the voltage of the waveform of the power supply voltage Vs is shared according to each of the relatively low voltage, medium voltage, and high voltage. Since the charging is performed, each charging is performed according to the change in the sine waveform of the power supply voltage Vs, so that the generation of harmonic current is further suppressed.
Therefore, it is possible to provide a rectifier circuit that can suppress harmonic current and boost DC voltage with a small number of switching operations.

Furthermore, the number (number) of series of smoothing capacitors performing the boosting operation in the smoothing capacitor group 4 is not limited to three or less. Four or more stages may be used.
At this time, further improvement in boosting efficiency and generation of harmonic current are suppressed. Note that the charging voltages of the capacitors of four or more stages are set to be unbalanced so as to be different from each other in order to improve the characteristics.

《Switching device》
In FIG. 1 and FIG. 11, the switching elements 62 and 63 in the bidirectional switch circuit (6, 60) may have a switching function, so that an IGBT (Insulated Gate Bipolar Transistor), a MOSFET (Metal-Oxide-Semiconductor). Elements and devices such as Field-Effect Transistor, Bipolar junction transistor, and BiCMOS (Bipolar Complementary Metal Oxide Semiconductor) can be applied.

<Parallel diode of switching element>
In FIG. 1 and FIG. 11, the switching elements 62 and 63 in the bidirectional switch circuit (6, 60) are shown as having diodes connected in parallel to each other. Since there is a parasitic diode, it is not necessary to add an external diode.

<Switching method of bidirectional switch circuit>
The present invention is not limited to the switching method shown in FIGS. 2 and 5 and can be applied to other switching methods. In other words, various changes and combinations of the timing of the generated pulses, the pulse width, and the presence or absence of the PWM method are possible.

《Rectifier circuit used in motor drive device》
In the motor drive device of the third embodiment, the rectifier circuit 11 has been described. However, the rectifier circuit 10 described in FIG. 1 may be used. Various control methods for the bidirectional switch circuits (60, 6) in the rectifier circuits 11, 10 are also possible.

DESCRIPTION OF SYMBOLS 1 AC power source 2 AC reactor 3, 61, 64 Rectifier 4 Smoothing capacitor group 5 Load, DC load 6, 60 Bidirectional switch circuit 7 Inverter device (inverter circuit)
8 PM motor (motor, permanent magnet synchronous motor)
9 Controller 10, 11 Rectifier circuit 41 Smoothing capacitor (first smoothing capacitor)
42 Smoothing capacitor (second smoothing capacitor)
43 Smoothing capacitor (third smoothing capacitor)
62, 63 Switching element 65, 611, 612, 613, 614, 641, 642 Diode

Claims (8)

An AC reactor connected to an AC power source;
The AC side is connected to the AC power supply via the AC reactor, the DC side is connected to a DC load, and a rectifier,
A plurality of smoothing capacitors connected in series provided between the DC side of the rectifier and the DC load;
A bidirectional switch circuit provided between the AC side of the rectifier and the connection points of the plurality of smoothing capacitors connected in series;
With
The rectifier circuit according to claim 1, wherein the charging voltages of the plurality of smoothing capacitors connected in series are unbalanced. An AC reactor connected to an AC power source;
The AC side is connected to the AC power supply via the AC reactor, the DC side is connected to a DC load, and a rectifier,
A first smoothing capacitor and a second smoothing capacitor connected in series provided between the DC side of the rectifier and the DC load;
A bidirectional switch circuit provided between the AC side of the rectifier and the connection point of the first smoothing capacitor and the second smoothing capacitor connected in series;
With
The voltage across the second smoothing capacitor is lower than half the DC voltage, and the voltage across the first smoothing capacitor is higher than half the DC voltage. The rectifier circuit according to claim 2,
The bidirectional switch circuit is
A short-circuit mode in which the AC power supply is short-circuited through the AC reactor;
A voltage doubler rectification mode for charging any one of the first and second smoothing capacitors connected in series;
A full-wave rectification charging mode for charging both the first and second smoothing capacitors connected in series;
A rectifier circuit comprising: The rectifier circuit according to claim 3,
The short circuit mode is performed at a timing when the AC voltage of the AC power supply becomes 0 volts. The rectifier circuit according to claim 2,
The bidirectional switch circuit is
In a region where the instantaneous voltage of the AC power source is lower than a predetermined value, the second smoothing capacitor is charged,
A rectifier circuit that operates to charge the first smoothing capacitor in a region where the instantaneous voltage of the AC power supply is equal to or higher than a predetermined value. The rectifier circuit according to claim 1 or 2,
When the value of the current flowing through the DC load connected to the rectifier circuit is smaller than a predetermined value, the operation of the bidirectional switch circuit is stopped. A rectifier circuit according to claim 1 or claim 2,
A motor drive device comprising an inverter circuit and a motor as the DC load is connected, and the switching operation of the bidirectional switch circuit is controlled in accordance with the load state or rotation speed of the motor. In the motor drive device according to claim 7,
A motor drive device comprising: a mode for stopping a switching operation of the bidirectional switch circuit according to a load state or a rotation speed of the motor; and a mode for performing a switching operation of the bidirectional switch circuit.