JP2013162658A - Power conversion device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power conversion device that is reliable, small, light, and of low cost, and does not require a complicated phase-shifting transformer, and is also capable of regenerative operation while suppressing increase in the number of transformers.SOLUTION: A power conversion device comprises: transformers 201-203 each of which comprises a primary winding connected with input terminals R, S, T and a secondary winding which is comprised of a plurality of mutually insulated single-phase open windings; converter cells 30U1-30W3 each of which comprises a switching element SW, and its input end is connected with each single-phase open winding, and thereby connected in parallel with the input terminal of each phase via the transformer, and its output end is connected in series with the output terminal of each phase, and each of which comprises a converter for performing three level or more voltage conversion and an inverter, and performs single-phase alternating current/single-phase alternating current conversion; and control means 601 for controlling on/off of the switching element SW.

Description

  The present invention relates to a power conversion device that converts AC power into AC power, for example, to a device that is applied to a device that drives a motor at a variable speed.

  FIG. 17 shows an example of a circuit configuration of a conventional first power converter. The power conversion device of FIG. 17 has a plurality of single-phase converters in which each AC terminal is connected in series for the purpose of obtaining a high-voltage output voltage to the motor connected to the output terminal. For the purpose of supplying power to the plurality of single-phase converters, a plurality of DC power supplies that are insulated from each other are generated by a transformer having a plurality of windings and a plurality of diode rectifiers, and Each is connected to a DC part. Moreover, in order to suppress the harmonic current on the input side, the transformer is a transformer (phase-shifting transformer) including a plurality of windings 3 to 11 whose phases are shifted from each other (for example, patents). Reference 1).

  On the other hand, FIG. 18 shows an example of a circuit configuration of a conventional second power converter. 18 is a circuit configuration in which a plurality of three-phase converters having a common DC voltage and a three-phase transformer are multiplexed and the secondary winding of the transformer is connected in series as an open winding. (For example, see Patent Document 2).

  Furthermore, FIG. 19 shows an example of a circuit configuration of a conventional third power converter. In the power conversion device of FIG. 19, the primary side of a single-phase transformer is connected in series with other single-phase transformers, and the tip of the single-phase transformer is connected to an input terminal. Each line is connected to a converter cell having a single-phase full-bridge converter / inverter composed of legs capable of two-level voltage output as shown in FIG. The AC terminal of the inverter is connected in multiple series with the AC terminal of the other inverter (see, for example, Patent Document 3).

US Pat. No. 5,625,545 (FIG. 1) Japanese Patent No. 3019655 (FIG. 1) JP 2009-106081 A (FIGS. 1 and 2)

  In the first power conversion device of FIG. 17, a transformer (phase-shifting transformer) including a plurality of windings whose phases are shifted from each other is required for the purpose of suppressing the harmonic current on the input side. This type of transformer has a problem of large size and high cost because of its complicated structure. Another disadvantage is that the diode rectifier limits the power flow in one direction.

  Further, in the second power conversion device of FIG. 18, since a transformer is used on the output side, when a load such as a motor that requires voltage change is connected to the output side, It is assumed that driving is restricted due to concerns about magnetic saturation. Specifically, it is conceivable that a low frequency voltage cannot be output. In addition, in order to generate a common DC power source, a configuration such as a self-excited converter using a diode rectifier or a switching element is considered, but when generating a DC power source from a high-voltage power source, an additional transformer, particularly For the purpose of reducing harmonics, problems such as the need for a phase-shifting transformer are assumed.

Further, in the third power conversion device of FIG. 19, since a self-excited converter is used, bidirectional power flow is possible, but since a single-phase transformer is used, the number of transformers increases. . In addition, since the single-phase transformers are directly connected in series, the primary voltage of the transformer may not be properly divided when the converter does not output voltage. Patent Document 3 also describes that a five-leg iron core three-phase transformer is used instead of a single-phase transformer.
However, even if a five-legged iron core is used, the cross-sectional areas of the fourth and fifth legs that are not wound are finite, so there is a concern that magnetic saturation may occur if control is performed without considering magnetic saturation. Is done. Since means for controlling the input current, the output voltage, and the DC bus voltage of each converter cell while preventing magnetic saturation is not known, reliability is a concern. Furthermore, since a leg capable of two-level voltage output is used for the converter cell, the output voltage per cell is small, and there is a disadvantage that the number of converter cells and the number of transformers increase.

  The present invention was made to solve the above problems, and without requiring a phase-shifting transformer with a complicated structure, regenerative operation is possible while suppressing an increase in the number of transformers, An object is to obtain a power converter with high reliability, small size, light weight, and low cost.

  A power conversion device according to the present invention is a power conversion device that performs power conversion between a multiphase AC input terminal and a polyphase AC output terminal, and includes a primary winding connected to the input terminal and a plurality of A transformer having a secondary winding composed of a single-phase open winding insulated from each other, and a switching element, an input end connected to each single-phase open winding, and an output end in series with each other to output each phase A plurality of converter cells connected to the terminal and performing conversion of single-phase alternating current / single-phase alternating current, and control means for controlling on / off of the switching element, each of the converter cells being single-phase from the input end A converter that converts an AC voltage into a DC voltage of three or more levels and outputs it to a capacitor series body, and an inverter that converts the DC voltage from the capacitor series body into a single-phase AC voltage and outputs it to the output terminal. is there.

  As described above, the power conversion device according to the present invention has a simple and light-weight structure with a small number of windings including a primary winding and a secondary winding of a single-phase open winding. The converter cell is composed of a converter and an inverter that perform voltage conversion at three or more levels, and the voltage waveform can be improved and high-voltage specifications are possible. Therefore, it is possible to reduce the number of units, and to realize a power conversion device that is small, lightweight, and low in cost.

It is a circuit diagram which shows the main circuit structure of the power converter device in Embodiment 1 of this invention. It is a figure which shows the coil | winding structure of the transformers 201-203 of FIG. It is a circuit diagram which shows the main circuit structure of the converter cells 30U1-30W3 of FIG. It is a figure explaining the internal structure of the control means 601 of FIG. FIG. 5 is a block diagram showing an input current control unit 610 in FIG. 4. FIG. 6 is a block diagram showing output voltage control means 620 in FIG. 4. It is a block diagram which shows the average voltage control means 631 of FIG. It is a block diagram which shows the interphase balance control means 632 of FIG. It is a block diagram which shows the intra-phase balance control means 633 of FIG. It is a block diagram which shows the balance control means 634 in a cell of FIG. It is a block diagram which shows the modulation means 640 of FIG. It is a timing chart explaining operation | movement of the PWM controller 801 which is a modulation means by the side of a converter. It is a timing chart explaining operation | movement of the PWM controller 802 which is a modulation means by the side of an inverter. 6 is a timing chart for explaining a phase relationship of triangular wave carriers in PWM controllers 801 and 802. It is a circuit diagram which shows the main circuit structure of the power converter device in Embodiment 2 of this invention. It is a figure which shows the coil | winding structure of the transformer 211 of FIG. It is a circuit diagram which shows an example of the circuit structure of the conventional 1st power converter device. It is a circuit diagram which shows an example of the circuit structure of the 2nd conventional power converter device. It is a circuit diagram which shows an example of the circuit structure of the conventional 3rd power converter device. FIG. 20 is a circuit diagram showing the converter cell of FIG. 19.

Embodiment 1 FIG.
FIG. 1 shows an example of a main circuit configuration of the power conversion device according to Embodiment 1 of the present invention. FIG. 1 shows an example in which a three-phase voltage source 101 is connected to input terminals R, S, and T of a power converter, and a three-phase motor 401 is connected to output terminals U, V, and W. That is, FIG. 1 shows an example in which the present invention is applied as a motor drive device.

  The main circuit of the power conversion device according to the first embodiment of the present invention includes a transformer 20n (n = 1, 2, 3,...) And a converter cell 30Xn (X = U, V, W,...). , N = 1, 2, 3,... That is, in the present invention, the polyphase alternating current applied to the input terminal and the output terminal is not limited to three phases. For example, the present invention can also be applied to an apparatus that includes three two-phase / two-phase transformers and six converter cells, converts AC two-phase from the input terminal into AC three-phase, and outputs from the output terminal. is there. Further, the number n of converter cells in series is not limited to 3, but in the example of FIG. 1 of the first embodiment, the voltage source 101 and the motor 401 are both AC three-phase, and three transformers 201 and 202 are used. 203, three units per phase, a total of nine converter cells 30U1, 30U2, 30U3, 30V1, 30V2, 30V3, 30W1, 30W2, and 30W3, which will be described below. Moreover, the control means 601 which controls ON / OFF of the switching element in a power converter device is provided.

  2A and 2B show an example of the winding configuration of the transformer 20n. The primary winding has a three-phase star connection (Y connection) winding structure, and each terminal is connected to input terminals R, S, and T of the power converter. The primary winding may use a delta connection (Δ connection). However, if the sum of the voltages applied to the secondary winding is not zero, a circulating current flows in the delta connection and the loss is reduced. Increase. Therefore, the primary winding is preferably a star connection.

  The secondary winding is a plurality of mutually isolated single-phase open windings. The secondary winding corresponds to the voltage between the terminals R, S, T on the primary side and the neutral point N of the star connection, that is, the voltage between RN, S-N, and TN. Voltages depending on the turns ratio are generated in the wire between Rs-Na, Ss-Nb, and Ts-Nc. Since the secondary winding is an open winding, one isolated voltage source is generated for each secondary winding. Thus, unlike the conventional circuit of FIG. 17, three or more secondary windings are not required for the purpose of generating one isolated voltage source.

Note that the total leakage inductance of the primary winding and the secondary winding is preferably designed to have a% impedance of 5% or more for the purpose of realizing the input current control means 610 described later.
That is, the% impedance is an important factor that determines the controllability of the current. Mainly, the controllability of the current is related to the% impedance (inductance component on the output side of the converter cell) and the switching frequency (controllable period), and the larger the both, the higher the controllability. In general, considering the target voltage class / capacity band (6.6 kV, 1 MVA, etc.), the switching frequency is limited to some extent. Therefore, it is appropriate that the% impedance is about 5% to 10%.

  Moreover, the iron core of a 3 or more leg is used for the iron core of a transformer. When a winding is wound around each of the legs of the three-legged iron core, if the total voltage of each winding is not zero, magnetic saturation may occur. Therefore, it is desirable to use a 4-legged or 5-legged iron core. However, since the effective cross-sectional area of the added leg (fourth leg or fifth leg) is finite, it is necessary to control the control means 601 described later so as not to cause magnetic saturation.

  3A and 3B show the main circuit configuration of the converter cell 30Xn. The converter cell 30Xn includes a single-phase full-bridge converter and an inverter having a leg capable of outputting a voltage of three or more levels, and performs single-phase AC / single-phase AC conversion. The DC terminals of the converter and the inverter are connected back to back to the capacitor series body CPCN. As an example, FIG. 3 shows a diode-clamped three-level conversion in which four arms comprising a switching element SW and a free-wheeling diode FD connected in antiparallel thereto are connected in series and connected to a neutral point by a clamp diode CD. It is based on the circuit of the vessel.

The diode-clamped three-level converter uses four legs. Of the four legs, two legs operate as converters.
The converter AC terminals IN1 and IN2, which are the input terminals of the converter cell, are connected to one winding on the secondary side of the transformer 201, for example, both ends Rs and Na of the single-phase open winding in FIG. . Therefore, the input ends of the converter cells are connected in parallel to the input terminals of the respective phases via the transformer.

The other two legs operate as inverters. The output terminals OUT1 and OUT2 of the inverter, which are the output terminals of the converter cell, are connected in series with the output terminals of the other converter cells and star-connected to the output terminals U, V, and W of the power converter. . Therefore, the output terminals of the converter cells are connected in series to the output terminals of the respective phases.
The phase of the output terminal to which the output ends of the converter cells connected in series with each other are connected and the phase of the input terminal to which the input ends of the converter cells are connected are the same phase.

  A series body of a positive capacitor CP and a negative capacitor CN is connected to both ends of the leg. In the following, the voltage applied to both ends of this capacitor series CPCN is the DC bus voltage, the voltage applied to the positive capacitor CP is the positive DC bus voltage, and the voltage applied to the negative capacitor CN is the negative DC bus. Defined as voltage.

  As described above, the advantage of the circuit configuration in the power conversion device of the present invention is that the self-excited converter is used, so the on / off state of the switching element on the converter side is controlled, and the harmonic current on the input side is controlled. Therefore, it is possible to reduce the number of windings because the structure is complicated, large and expensive, and no phase-shifting transformer is required, and the transformer secondary winding uses a single-phase open winding. A large number of mutually isolated voltage sources can be secured, and the number of cells can be reduced by using a leg that can output three levels of voltage to the converter cell, and the secondary winding of the transformer can be reduced. The number of can be reduced.

  In particular, there is an advantage that the number of converter cells can be reduced by half by using a leg that can output a three-level voltage compared to a case that uses a leg that can output a two-level voltage. The fact that the number of converter cells is halved reduces the number of transformer windings by half because the number of necessary isolated power supplies is halved. Furthermore, by using a leg capable of outputting three levels of voltage, the output voltage or harmonic components of the current can be reduced. This reduction of harmonic components provides further advantages to the circuit configuration of the present invention. That is, the loss of the transformer is reduced by reducing the harmonic voltage applied to the transformer and the flowing harmonic current. Therefore, the transformer can be further reduced in weight and size, contributing to energy saving.

  In recent years, legs that can output three levels of voltage, that is, a group of semiconductor elements including legs composed of four switching elements SW, a freewheeling diode FD, and two clamp diodes CD, have been housed in one module. Therefore, even if the leg is capable of outputting three levels of voltage, one converter cell has a size not much different from that of two levels. That is, the volume, weight, and cost of the entire converter cell can be reduced by the amount of the converter cell that is reduced.

  Next, the control means 601 will be described. The control means 601 makes the current flowing through the input terminal close to an ideal sine wave current (reducing harmonics), controls the motor 401 to a desired rotational speed or torque, and sets the DC bus voltage to an appropriate value. The main purpose is to prevent overvoltage breakdown of the semiconductor element by controlling the current to the current, the current flowing through the input terminal of the power converter, or the current flowing through the converter cell, the voltage at the input terminal of the power converter, and the conversion Using the sensor detection value 502 (see FIG. 2) such as the DC bus voltage (positive side DC bus voltage, negative side DC bus voltage, and the total voltage of both) of the detector cell, the converter cell is finally used. A control signal 501 for controlling on / off of the switching element is derived.

  The internal configuration of the control means 601 is shown in FIG. The control means 601 is roughly divided into four parts: an input current control means 610, an output voltage control means 620, a bus voltage control means 630, and a modulation means 640. Furthermore, the bus voltage control means 630 is composed of an average voltage control means 631 and an interphase. The balance control means 632, the in-phase balance control means 633, and the in-cell balance control means 634 are included.

  The input current control means 610 is reflected in the control on the converter side, the output voltage control means 620 is reflected in the control on the inverter side, and the average voltage control means 631 in the bus voltage control means 630 is controlled in the control on the converter side. The interphase balance control means 632 is reflected in the inverter side control, the in-phase balance control means 633 is reflected in the inverter side control, and the in-cell balance control means 634 is applied to both the converter side and the inverter side. It is reflected in control or either control. Modulation means 640 is finally reflected in the control of the switching elements on the converter side and the inverter side.

  Prior to detailed description of the control means 601, each variable is defined. First, the voltage (power supply voltage) of the input terminal is Vr, Vs, Vt, and the current flowing through the input terminal is Ir, Is, It. The currents that flow on the secondary side of the transformer are IRsn, ISsn, and ITsn. Note that n is 1, 2, 3, corresponding to the order of the transformers 201, 202, 203. The DC bus voltage of the converter cell 30Xn is assumed to be VdcXn. X is any one of U, V, and W, and n is any one of 1, 2, and 3.

  Further, the voltage command value on the converter side of the converter cell 30Xn is set to VCXn *, and as shown by an arrow in FIG. 3B, the voltage command value to the positive side switching element is set to VCXnP *, The voltage command value to the switching element on the negative electrode side is set to VCXnN *. Similarly, the voltage command value on the inverter side is VIXn *, of which the voltage command value for the positive switching element is VIXnP * and the voltage command value for the negative switching element is VIXnN *.

  A control block diagram showing an example of the input current control means 610 is shown in FIG. The main purpose of the input current control means 610 is to make the current IRsn, ISsn, ITsn flowing through the input terminals R, S, T, or the secondary side of the transformer follow the current command value. The input current control means 610 sets three converter cells connected to one transformer as one set, and performs control independently of the other sets.

  First, input currents IRsn, ISsn, and ITsn of the converter cell are detected. These are subjected to dq conversion using the power supply phase θ to derive a d-axis current Idn and a q-axis current Iqn. Assuming a case where the d-axis current corresponds to the reactive current (reactive power) and the q-axis current corresponds to the active current (active power) when the power supply voltage is three-phase balanced by the power supply phase θ and dq conversion, The following will be described. Deviations between the obtained Idn and Idq and the respective current command values Idn * and Iqn * are calculated and given to the controller Gc (s). PI control or the like is applicable to the controller Gc (s). Here, since Idn * is a command value corresponding to the reactive current, Idn * = 0 is set so that the power factor becomes approximately 1, and Iqn * corresponds to the effective current. Derived by means 631.

  A d-axis voltage Vds and a q-axis voltage Vqs of the power supply voltage are considered as feedforward amounts in the output of the controller Gc (s). Note that Vds and Vqs are obtained by performing dq conversion on the power supply voltages Vr, Vs, and Vt and multiplying by the turns ratio TR of the transformer. Thereafter, inverse dq conversion is performed to obtain voltage command values VCUn *, VCVn *, VCWn * on the converter side of the converter cell. Since a transformer is connected to the converter side, it is necessary not to output a zero-phase voltage for the purpose of preventing magnetic saturation. Alternatively, magnetic saturation may be prevented by controlling the zero-phase current derived from the sum of IRsn, ISsn, and ITsn to be zero.

  The above is an example, and it is possible to incorporate a known technique such as non-interference current control so as not to interfere with the d-axis and q-axis currents. Also, it is possible to control the active power P and the reactive power Q more strictly by using PQ conversion instead of dq conversion.

Next, a control block diagram showing an example of the output voltage control means 620 is shown in FIG. In FIG. 6, the total voltage command values VIU *, VIV *, and VIW * on the inverter side of each phase are obtained by using a known motor control technique (for example, V / f constant control, vector control, direct torque control, etc.). obtain. Furthermore, the voltage utilization factor is improved by adding a zero-phase voltage component Vz * having an output frequency of three times to these voltage command values.
Since this method itself is known, the details are omitted, but a common zero-phase voltage Vz * is added so that the amplitude of the peak value portion of each phase on the inverter side becomes small. This addition causes distortion in the voltage waveform, but the cause of the distortion waveform is the zero-phase voltage, so when it is supplied to the load with three-phase three-wires, only a clean sine wave with this distortion waveform removed is present in the load. Supplied as voltage.

  The reason why this method is not applied to the converter side is that a transformer is connected to the converter side, so when adding and outputting the zero-phase voltage, a magnetic flux that does not become zero in total for the three phases is generated in the transformer. This is because it is necessary to enlarge the iron core of the fourth and fifth legs of the transformer, which is disadvantageous.

  Thereafter, a zero-phase voltage command value Vzb * determined by an interphase balance control means 632, which will be described later, is added, and divided by the number of cells per phase (= 3), whereby a voltage command per cell on the inverter side is obtained. VIU **, VIV **, and VIW ** are output as provisional values.

  The bus voltage control means 630 is configured to control the DC bus voltage of each converter cell to a predetermined value by four control means including an average voltage control means 631, an interphase balance control means 632, an in-phase balance control means 633, and an in-cell balance control means 634. Control to voltage.

A control block diagram showing an example of the average voltage control means 631 is shown in FIG. The average voltage control means 631 predetermines an average value of DC bus voltages VdcUn, VdcVn, VdcWn of three converter cells connected to one transformer, that is, an average value VdcAVGn over three phases U, V, W. Q-axis current command value Iqn * corresponding to the input current effective component of the primary winding of the transformer is determined so as to follow the bus voltage command value Vdc *.
Specifically, the deviation between VdcAVGn and Vdc * is calculated and given to the controller Gv (s) to calculate Iqn *. A PI controller or the like can be used as the controller Gv (s). Since Iqn * is a current corresponding to active power, VdcAVGn can be made to follow Vdc *. As described above, when PQ conversion is used for the input current control means 610, the command value P * of the active power is adjusted.

  Regarding the connection of converter cells, converter cells connected in series on the inverter side are connected in parallel via a transformer on the converter side, and these converter cells connected in series and in parallel are both the same. Connected to the phase. Then, the average voltage control means 631 controls the three converter cells connected to one transformer as one set. As a result, when the average value VdcAVGn of the DC bus voltage is obtained, the voltage oscillation generated in each DC bus voltage is canceled.

  That is, for example, in general, when a single-phase voltage is output, the output voltage vibrates at twice the frequency. Therefore, the DC bus voltage also vibrates at twice the frequency. Since the vibration phases of VdcUn, VdcVn, and VdcWn are different from each other by 120 °, the three-phase average value VdcAVGn is canceled and the vibration component having the double frequency becomes zero. Therefore, the average voltage control means 631 can be realized more easily.

  Next, a control block diagram showing an example of the interphase balance control means 632 is shown in FIG. The interphase balance control means 632 adjusts the zero-phase voltage Vzb * superimposed on the voltage command value on the inverter side of each phase (see FIG. 6 above), whereby the average voltage VdcUAVG of each phase (the average value of VdcU1 to VdcU1-3). ), VdcVAVG (average value of VdcV1 to 3) and VdcWAVG (average value of VdcW1 to 3) are uniformly balanced.

  Specifically, for each of the average voltages VdcUAVG, VdcVAVG, and VdcWAVG of each phase, a deviation from the overall average voltage VdcAVG is calculated and given to the controller Gp (s) via an LPF (Low-Pass Filter). Thereafter, the product of the inverter-side voltage command values VIU *, VIV *, and VIW * is calculated for each phase, and the results are summed to obtain the zero-phase voltage command value Vzb *. The reason for applying the LPF is to remove a frequency component twice the output frequency generated in the DC bus voltage as described above. A PI controller or the like can be used as the controller Gp (s).

  When the control is performed in this way, the voltage of the phase in which the average value of the DC bus voltage has decreased during motor powering decreases, so the output power of that phase decreases, and the DC bus voltage of that phase recovers. Furthermore, the bus voltage average values of all phases are balanced. Note that the motor regeneration can be handled by reversing the polarity of the controller Gp (s).

  Next, a control block diagram showing an example of the in-phase balance control means 633 is shown in FIG. The in-phase balance control means 633 adjusts the output voltage sharing of the inverters in the phase to uniformly balance the DC bus voltages in the phase with each other. Specifically, the deviation between the DC bus voltage VdcX1 to 3 of its own and the bus voltage average value VdcXAVG in the phase is calculated and given to the controller Gb (s). The result is equivalent to the adjustment ratio of the output voltage sharing, and the adjustment range is derived by multiplying the voltage command value VIX ** provisionally determined in FIG. 6 (output voltage control means 620). Thereafter, it is added to VIX ** to derive final voltage command values VIX1 *, VIX2 *, VIX3 *.

  By performing the control as described above, the output voltage of the inverter of the converter cell having a relatively small bus voltage is reduced during powering of the motor, so that the output power can be suppressed. As a result, the DC bus voltage in the phase is reduced. Can be balanced. The motor regeneration can be handled by reversing the polarity of the controller Gb (s).

  Next, an example of the in-cell balance control means 634 is shown in FIG. The in-cell balance control means 634 adjusts the voltage ratio between the positive side leg and the negative side leg to uniformly balance the positive side DC bus voltage and the negative side DC bus voltage with each other. The converter side, the inverter side This can be realized by reflecting either or both.

  First, the converter side will be described with reference to FIG. The voltage command value VCXn * of the converter is multiplied by 1/2 to calculate the voltage command value VXnP * of the positive leg and the voltage command value VXnN * of the negative leg multiplied by -1. Also, the deviation between VdcXnN and VdcXnP is calculated and given to the controller Gcz (s) to calculate VXnCz *. Thereafter, VXnCz * is added to VXnP * and VXnN *, respectively, to calculate the final positive electrode leg voltage command value VCXnP * and the negative electrode leg voltage command value VCXnN *.

  When the control is performed as described above, the voltage command value on the capacitor side having a low voltage increases during motor power running (that is, when power is input to the converter), and the positive and negative DC buses are increased. The voltage can be balanced. Conversely, during motor regeneration, this can be dealt with by determining the polarity of the controller Gcz (s).

  The basic principle is the same for the inverter side shown in FIG. However, since the inverter outputs electric power during motor power running, the final voltage command values VIXnP * and VIXnN are obtained by subtracting VXnIZ * calculated by the controller Giz (s) from the command values of the positive and negative legs. * Calculate. At the time of motor regeneration, the polarity of the controller Giz (s) is reversed to cope with it.

  Finally, the modulation means 640 can appropriately output the voltage command values VCXnP * and VCXnN * on the converter side and the voltage command values VIXnP * and VIXnN * on the inverter side derived by the control means described above. Then, pulse width modulation (PWM) is performed to derive a gate signal for controlling on / off of each switching element. FIGS. 11 (a) and 11 (b) show an example of the modulation means 640. Each voltage command value is given to the PWM controller 801 (converter side) or the PWM controller 802 (inverter side). In addition, a dead time process is performed so as to delay the rise, and a gate signal for controlling on / off of the switching element is output.

  When attention is paid to one leg of the three-level conversion circuit, there are various known examples of the modulation means, and the specific modulation means is not limited in the present invention. The modulation means 640 in the present invention intends that the switching timing of the positive-side leg and the negative-side leg is not overlapped as much as possible, and the switching timing of the converters connected in parallel via the transformer is not overlapped as much as possible. In other words, the switching timings of the inverters connected in series are not overlapped as much as possible to obtain an input current and output voltage with less harmonic components.

Hereinafter, a case where modulation is performed on one leg by using one set of two triangular wave carriers for positive voltage output and negative voltage output will be described.
First, regarding the leg on the converter side, as shown in FIG. 12, the triangular wave carriers CarCPn and CarCNn are compared with the voltage command value VCXnP * of the positive leg and the voltage command value VCXnN * of the negative leg. CarCPn and CarCNn have the same phase, and the amplitude of CarCPn corresponds to the voltage across the positive capacitor CP of the corresponding converter cell, and the amplitude of CarCNn corresponds to the voltage across the negative capacitor CN.

  GXnCP1, GXnCP2, GXnCP3, GXnCP4, and the negative leg gate signal from the positive DC terminal side switching element GXnCN1, GXnCN2, GXnCN3 from the positive DC terminal side switching element on the converter side , GXnCN4, GXnCP1 and GXnCP3 from the magnitude relationship between CarCPn and VCXnP *, GXnCP2 and GXnCP4 from the magnitude relationship between CarCNn and VCXnP *, GXnCN1 and C from the magnitude relationship between CarCPn and VCXnN *, GXnCN2 and GXnCN4 are determined from the magnitude relationship with *.

  When the voltage command value is larger than the triangular wave carrier, the positive side switching element is turned on and the negative side switching element is turned off. When the magnitude relationship is reversed, the on / off is reversed. Finally, dead time processing is performed so as to delay the rise of each gate signal, and a final gate signal is determined. Since the dead time processing is known, a description thereof will be omitted.

  Similarly, regarding the leg on the inverter side, as shown in FIG. 13, the triangular wave carriers CarIPn and CarINn are compared with the voltage command value VIXnP * of the positive leg and the voltage command value VIXnN * of the negative leg. CarIPn and CarINn have the same phase, and the amplitude of CarIPn corresponds to the voltage across the positive-side capacitor CP of the corresponding converter cell, and the amplitude of CarINn corresponds to the voltage across the negative-side capacitor CN.

  In FIG. 13, the waveforms of the voltage command values VIXnP * and VIXnN * are distorted from the sine wave by adding the zero-phase voltage component Vz * described in FIG. 6 (output voltage control means 620). It is based on being.

  GXnIP1, GXnIP2, GXnIP3, GXnIP4, and the negative leg gate signal from the positive DC terminal side switching element GXnIN1, GXnIN2, GXnIN3 from the positive DC terminal side switching element on the inverter side , GXnIN4, GXnIP1 and GXnIP3 from the magnitude relationship between CarIPn and VIXnP *, GXnIP2 and GXnIP4 from the magnitude relationship between CarINn and VIXnP *, GXnIN1 and GXnIN3 from the magnitude relationship between CarIPn and VIXnN *, GXnIN2 and GXnIN4 are determined from the magnitude relationship with *.

  When the voltage command value is larger than the triangular wave carrier, the positive side switching element is turned on and the negative side switching element is turned off. When the magnitude relationship is reversed, the on / off is reversed. Finally, dead time processing is performed to determine a final gate signal.

  The important point in the modulation means 640 is the phase relationship of the triangular wave carrier. In the voltage output by one leg, harmonic components near the carrier frequency are dominant. When attention is paid to one converter or inverter, the voltage command values of the positive and negative legs are almost inverted (because the negative side is multiplied by -1 in FIG. 10) and therefore equivalent. Thus, the carrier frequency component is canceled out, and the harmonic component in the vicinity of twice the carrier frequency becomes dominant.

  Further, on the converter side, as shown in FIG. 14A, the phase of the triangular wave carriers CarCP1, CarCP2, and CarCP3 (CarCN1, CarCN2, and CarCN3) is shifted by 60 degrees (π / 3 rad) to the input current. It is possible to cancel harmonic components in the vicinity of a frequency twice as high as the contained carrier frequency. Finally, the number of legs × the number of parallel multiplexes, that is, in this example, harmonic components in the vicinity of the frequency 2 × 3 = 6 times the carrier frequency become dominant. Therefore, since a low-order harmonic component having a large amplitude can be canceled, an input current having a small harmonic component can be obtained. Also, the remaining harmonic components are very high in the vicinity of 6 times the carrier frequency and can be easily removed simply by adding a small filter to the input terminal or the converter side of the converter cell. Is possible.

On the other hand, on the inverter side, as shown in FIG. 14B, the phase of the triangular wave carriers CarIP1, CarIP2, and CarIP3 (CarIN1, CarIN2, and CarIN3) is shifted by 60 degrees (π / 3 rad) to the output voltage. It is possible to cancel harmonic components in the vicinity of a frequency twice as high as the contained carrier frequency. Finally, the number of legs × the number of serial multiplexes, that is, in this example, harmonic components in the vicinity of the frequency 2 × 3 = 6 times the carrier frequency become dominant. Therefore, since a low-order harmonic component having a large amplitude can be canceled, an output voltage having a small harmonic component can be obtained.
Further, since the inverter side is connected in series, the output voltage level can be increased according to the number of potentials of the capacitor by shifting the switching timing.

  As described above, when the power conversion device according to the present invention is used, a phase shift transformer having a complicated structure as in the related art is not required, and the converter cell is converted into a converter having three or more levels. Since the number of converter cells and the number of windings of the transformer can be reduced, it is possible to reduce the size, weight, and cost. Furthermore, since a self-excited converter is used for the converter cell, a regenerative operation is possible. Furthermore, the magnetic saturation of the transformer and the bus voltage are appropriately controlled by the control means, and the reliability is improved.

Embodiment 2. FIG.
FIG. 15 shows an example of the main circuit configuration of the power conversion device according to Embodiment 2 of the present invention. In FIG. 15, the transformer is different from the transformer shown in FIG. That is, the transformer 211 in the second embodiment has a secondary winding composed of a plurality of (here, three) windings with respect to the primary winding of one phase, as shown in FIG. Winding is provided, and the transformers 201, 202, and 203 in the first embodiment are integrated into one transformer 211.

  The primary winding of the transformer 211 is a three-phase star connection, the secondary winding is three single-phase open windings per phase, and one transformer 211 makes a total of nine A power supply consisting of single-phase open windings insulated from each other is secured.

  By consolidating the three transformers into one, the transformer can be further reduced in size, weight, and cost. In addition, the integration of the transformers exhibits a further effect because an open winding is used as the secondary winding in the present invention. That is to increase the degree of freedom of combination of controls. In the first embodiment, the input current control means 610 is applied to a set of converter cells (for example, a set of 30U1, 30V1, and 30W1) connected to one transformer, but one transformer is used. For example, 30U1, 30V2, and 30W3 can be combined into one set, for example, so that the control combination can be freely controlled. By using this, optimum design can be performed in consideration of insulation between control lines and control signals. It becomes.

As in the first embodiment, it is desirable to use a three-legged or more iron core as the transformer iron core and to set the leakage inductance to about 5% or more.
In Embodiments 1 and 2, the leakage inductance of the transformer is considered about 5%, but an additional reactor may be inserted. This may be inserted on the primary side of the transformer or on the secondary side. Further, a capacitor is added and an LC filter is added on the primary side or secondary side of the transformer. May be. If a reactor or LC filter is added in this way, the harmonic component of the input current can be further suppressed.

  Furthermore, in the first and second embodiments, it is assumed that an IGBT (Insulated-Gate Bipolar Transistor) is used as the switching element SW, but other types such as a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) are used. These switching elements may be used.

  Also, silicon is usually used as the material for the semiconductor elements constituting the switching elements SW, diodes FD, and CD, but a wide band gap material such as silicon carbide, gallium nitride-based material, or diamond has a larger band gap than that of silicon. Since it is possible to increase the breakdown voltage of the semiconductor element, the number of the converter cells described above can be further reduced. Furthermore, since switching can be speeded up, it is possible to obtain an input current and an output voltage with smaller harmonic components.

  It should be noted that the present invention can be freely combined with each other within the scope of the invention, and each embodiment can be appropriately modified or omitted.

101 voltage source, 20n, 201, 202, 203, 211 transformer,
30Xn, 30U1-30W3 converter cell, 401 motor, 601 control means,
610 input current control means, 620 output voltage control means, 630 bus voltage control means,
631 Average voltage control means, 632 Interphase balance control means,
633 In-phase balance control means, 634 In-cell balance control means, 640 Modulation means, 801, 802 PWM controller, R, S, T input terminals, U, V, W output terminals,
SW switching element, CP positive side capacitor, CN negative side capacitor,
CPCN series capacitor body.

Claims (16)

A power conversion device that performs power conversion between a polyphase AC input terminal and a polyphase AC output terminal,
A transformer comprising a primary winding connected to the input terminal and a secondary winding comprising a plurality of mutually isolated single-phase open windings; and a switching element having an input terminal at each single-phase open winding A plurality of converter cells that are connected to a line and whose output ends are connected in series with each other and connected to the output terminals of each phase and perform single-phase AC / single-phase AC conversion, and control means for controlling on / off of the switching element And
Each converter cell converts a single-phase AC voltage from the input terminal into a DC voltage of three or more levels and outputs it to a capacitor series body, and converts the DC voltage from the capacitor series body into a single-phase AC voltage. An electric power converter comprising: an inverter that converts and outputs the output to the output terminal. 2. The power converter according to claim 1, wherein the control means comprises bus voltage control means for controlling a DC bus voltage, which is a voltage of the capacitor series body of each converter cell, to a predetermined voltage. The bus voltage control means is configured so that an average value over the different phases of the DC bus voltages in the converter cells connected to different phases of the output terminal becomes a predetermined bus voltage command value. The power converter according to claim 2, further comprising an average voltage control means for controlling an input current effective component of the next winding. The bus voltage control means includes: the converter cells connected to different phases of the output terminals so that the DC bus voltages in the converter cells connected to different phases of the output terminals are uniformly balanced with each other. 4. The power conversion device according to claim 2, further comprising interphase balance control means for controlling a voltage command value of the inverter. The bus voltage control means is configured so that the DC bus voltages in the plurality of converter cells connected in series with each other of the output terminals are uniformly balanced with respect to each other. 5. The power conversion device according to claim 2, further comprising an in-phase balance control unit configured to control a voltage command value of the inverter. The capacitor series body is composed of a positive-side capacitor and a negative-side capacitor connected in series with each other, and the DC-bus voltage of the converter cell is applied to the positive-side capacitor and the negative-side It consists of the negative side DC bus voltage applied to the capacitor,
The bus voltage control means includes a voltage to the switching element that constitutes the converter and / or the inverter so that the positive DC bus voltage and the negative DC bus voltage are uniformly balanced with each other in each converter cell. The power converter according to any one of claims 2 to 5, further comprising an in-cell balance control means for controlling the command value. The control means includes a plurality of the converters connected in series for each phase of the output terminal so as to reduce harmonic components contained in an input current to the input terminal and / or an output voltage from the output terminal. 7. The timing for switching the converter and / or the switching element constituting the inverter in the converter cell is controlled so as to be shifted with respect to each other in the plurality of converter cells. 8. The power converter described. The control means includes modulation means for performing pulse width modulation control using a carrier signal, and the modulation means shifts the phase of the carrier signal to each other by the plurality of the converter cells, thereby performing the switching. 8. The power conversion device according to claim 7, wherein the switching timing of the elements is shifted between the plurality of converter cells. 9. The electric power according to claim 1, wherein each of the primary windings is composed of a plurality of transformers connected in parallel to the input terminal. Conversion device. The power converter according to any one of claims 1 to 8, wherein the transformer includes a plurality of the secondary windings with respect to the primary winding of one phase. The inputs of the plurality of converter cells connected to the output terminal with the same number of phases of the multiphase alternating current of the input terminal and the multiphase alternating current of the output terminal, the output ends of which are connected in series with each other The end is connected in parallel with each other through the transformer to the input terminal of the same phase as the phase of the output terminal to which the output end is connected. The power conversion device according to item 1. 12. The electric power according to claim 1, wherein the first multi-phase alternating current is a three-phase alternating current, and the primary winding of the transformer is a three-phase star connection. Conversion device. The power converter according to claim 12, wherein the iron core of the transformer is composed of four or more iron cores. The group of semiconductor elements including the switching elements and diodes constituting the converter or the inverter of each converter cell is housed in one module. The power converter device described in 1. At least one of the switching element and the diode constituting the converter or the inverter of each converter cell is formed of a wide band gap semiconductor material having a larger band gap than silicon. The power conversion device according to any one of claims 1 to 14. The power converter according to claim 15, wherein the wide gap semiconductor material is silicon carbide, a gallium nitride-based material, or diamond.

Images