JP2011050178A - Motor control device and generator control device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To stabilize a torque response when a voltage is saturated. <P>SOLUTION: A motor control device adopts direct torque control. Namely, a magnetic flux which interlinks an armature winding of a motor is estimated, and a torque is estimated from an estimated magnetic flux and a motor current so as to directly control the torque. PI control for converging an error (ΔT) between the estimated torque and a torque command value to zero is performed. Presence of the occurrence of voltage saturation is detected from a difference (θ<SB>DIF</SB>) between a phase (θ<SB>S</SB><SP>*</SP>) of a magnetic flux vector following PI control and a phase (θ<SB>S</SB>) of estimated magnetic flux. When voltage saturation occurs, an integral gain (γ<SB>i</SB>*K<SB>i</SB>) in PI control is variably set in accordance with the difference (θ<SB>DIF</SB>). <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a motor control device for controlling a motor and a generator control device for controlling a generator.

  As a motor control method, a method of controlling torque and magnetic flux is known, unlike a method of controlling a motor current using a current control system, and the latter control is called direct torque control. In direct torque control, the magnetic flux linked to the armature winding of the motor and the generated torque of the motor are estimated, and the voltage command value is created so that the estimated magnetic flux and the estimated torque follow the magnetic flux command value and the torque command value. . Then, a desired torque is obtained by applying the voltage command value to the PWM inverter.

  The magnetic flux command value can be generated using proportional-integral control so that the error between the estimated torque and the torque command value converges to zero. This proportional-integral control operates on the assumption that a voltage command value according to the magnetic flux command value, that is, a voltage command value for making the generated torque of the motor coincide with the torque command value is generated in the voltage command value generation unit. However, when the voltage saturation occurs, such as when the motor rotates at high speed, the assumption is lost, and torque control may become unstable.

  Voltage saturation refers to a state in which the magnitude of the voltage command value (designated magnitude of the applied voltage of the motor) exceeds the upper limit voltage value that can be output by the PWM inverter. When voltage saturation occurs, the voltage command value is corrected so that the magnitude of the voltage command value is less than or equal to the upper limit voltage value for the purpose of protecting the power supply. A voltage command value smaller than the command value is generated in the voltage command value generation unit. As a result, the linearity of the torque control system (the linearity of proportional-integral control) is often lost, and the responsiveness of torque control often deteriorates or becomes unstable. Typically, for example, a positive error or a negative error between the estimated torque and the torque command value is steadily generated, so that the integral value in the proportional integral control becomes excessively large, which causes a so-called windup phenomenon. It is. When the windup phenomenon occurs in the proportional-integral control, an operation delay due to accumulation of the integral value occurs, and an overshoot of the torque to be controlled occurs.

  Several methods for stabilizing the control against such voltage saturation have been proposed. For example, in Patent Document 1 below, the integral value in the q-phase integrator is rewritten to a small value when voltage saturation occurs. Further, for example, in Patent Document 2 below, when voltage saturation occurs, the motor control is changed from proportional-integral control based on current loop processing to vector angle control (the combined vector of the voltage command values vd and vq of the d-axis and q-axis The angle with respect to the q axis is controlled.

Japanese Patent Application Laid-Open No. 6-1553569 JP 2008-67582 A

  However, since the methods according to Patent Documents 1 and 2 are applicable only when the d-axis and q-axis components of the motor current are controlled using the current control system, current control systems such as direct torque control are applicable. It is not applicable to motor control that does not constitute Although the prior art relating to the motor control method has been described, the same applies to the generator control method.

  Then, an object of this invention is to provide the motor control apparatus and generator control apparatus which contribute to stabilization of torque control.

  The motor control device according to the present invention is a torque control that controls the generated torque using an integrator so that the generated torque of a motor having an armature winding follows a torque command value that is a target value of the generated torque. In the motor control apparatus including the unit, the torque control unit is a command value that is to be generated to cause the generated torque to follow the torque command value, and a command value that specifies an applied voltage of the motor. When the magnitude of the voltage command value exceeds a predetermined limit voltage value and the magnitude is limited to the limit voltage value or less, the parameter of the integrator is based on the information on the interlinkage magnetic flux of the armature winding. It is characterized by adjusting.

  A state where the magnitude of the voltage to be applied to the motor in order to make the generated torque follow the torque command value, that is, the state where the magnitude of the voltage command value exceeds a predetermined limit voltage value is called voltage saturation. If the voltage is limited as described above when voltage saturation occurs, the difference between the generated torque and the torque command value is accumulated in the integrator, the output of the integrator becomes excessively large, and the torque response becomes poor. There is a concern that torque control becomes unstable. On the other hand, the occurrence of voltage saturation can be estimated by looking at magnetic flux information corresponding to voltage integration. Considering this, the motor control device according to the present invention adjusts the parameters of the integrator based on the information on the interlinkage magnetic flux of the armature winding when the voltage is saturated. As a result, integral control suitable for voltage saturation becomes possible, and torque control can be stabilized.

  Specifically, for example, the integrator outputs an integrated value of a value corresponding to a difference between the generated torque and the torque command value, and the torque control unit uses the output of the integrator to output the difference. A voltage command value generation unit that executes control including integration control for reducing the voltage, generates the voltage command value based on the control including the integration control and the limit voltage value, and the voltage command value and the The motor control device further includes an estimation unit that estimates the interlinkage magnetic flux based on an armature current flowing through the armature winding, and the information about the interlinkage magnetic flux is estimated by the estimation unit. Generated based on the result.

  Based on control including integral control (for example, proportional integral control), the applied voltage value of the motor necessary for causing the generated torque to follow the torque command value is known. However, the applied voltage value needs to be limited to the limit voltage value or less. Therefore, the voltage command value generation unit generates a voltage command value based on the control including the integration control and the limit voltage value. When the applied voltage value of the motor necessary to make the generated torque follow the torque command value exceeds the limit voltage value, and the actual applied voltage value is limited below the limit voltage value due to this, the voltage command The flux linkage estimated from the value deviates from the target value of the flux linkage according to the control including the integral control. Therefore, it is possible to detect the occurrence of voltage saturation from the estimation result of the interlinkage magnetic flux, and it is possible to perform integral control suitable for voltage saturation using the detection result.

  More specifically, for example, the torque control unit generates a phase command value that is a target of the phase of the flux linkage on a predetermined coordinate system using the integrator so that the difference is reduced, The voltage command value generation unit is configured to set the voltage command value necessary to link the magnetic flux according to the phase command value to the armature winding, and the magnitude of the voltage command value is equal to or less than the limit voltage value. The information on the flux linkage generated under restriction includes the phase of the estimated flux linkage, which is the flux linkage estimated by the estimation unit, on the predetermined coordinate system, and the phase command value. This is phase difference information representing the difference.

  More specifically, for example, the torque control unit generates the phase command value according to an integral value obtained by multiplying the difference by an integral gain as the parameter, and is represented by the phase difference information. As the difference increases, the integral gain is decreased.

  Thereby, it is suppressed that the output of the integrator becomes excessively large at the time of voltage saturation, and the torque control is stabilized.

  Further, for example, the estimation unit also estimates the generated torque based on the voltage command value and the armature current, and the torque generation unit uses the estimated generated torque as the generated torque.

  The generator control device according to the present invention controls the generated torque using an integrator so that the generated torque of the generator having the armature winding follows a torque command value that is a target value of the generated torque. In the generator control device including a torque control unit, the torque control unit is a command value that should be generated to cause the generated torque to follow the torque command value, and specifies an output voltage of the generator. When the magnitude of the voltage command value, which is a command value, exceeds a predetermined limit voltage value, and the magnitude is limited to the limit voltage value or less, the information is based on information on the interlinkage magnetic flux of the armature winding. It is characterized by adjusting the parameters of the integrator.

  As a result, similar to the motor control device, integral control suitable for voltage saturation is possible, and torque control can be stabilized.

  ADVANTAGE OF THE INVENTION According to this invention, it becomes possible to provide the motor control apparatus and generator control apparatus which contribute to stabilization of torque control.

  The significance or effect of the present invention will become more apparent from the following description of embodiments. However, the following embodiment is only one embodiment of the present invention, and the meaning of the term of the present invention or each constituent element is not limited to that described in the following embodiment. .

1 is a schematic block diagram of a motor drive system according to a first embodiment of the present invention. It is a block diagram showing the internal structure of the motor shown by FIG. FIG. 2 is an analysis model diagram of the motor shown in FIG. 1. 1 is a detailed block diagram of a motor drive system according to a first embodiment of the present invention. It is a figure which concerns on 1st Embodiment of this invention and shows the relationship between a magnetic flux vector, (alpha) axis | shaft, and (beta) axis. It is a figure which concerns on 1st Embodiment of this invention and shows the relationship between a voltage command vector, an estimated magnetic flux vector, and a magnetic flux command vector when voltage saturation has not occurred. It is a figure which concerns on 1st Embodiment of this invention and shows the relationship of a voltage command vector, an estimated magnetic flux vector, and a magnetic flux command vector at the time of generation | occurrence | production of voltage saturation. It is an internal block diagram of the torque control part which concerns on 1st Embodiment of this invention. It is a figure which concerns on 1st Embodiment of this invention and shows the waveform of the estimated torque waveform when the torque command value is changed in steps, and the waveform of the coefficient for integral gain. It is a detailed block diagram of the electric power generation system which concerns on 2nd Embodiment of this invention.

  Hereinafter, embodiments of the present invention will be specifically described with reference to the drawings. In each of the drawings to be referred to, the same part is denoted by the same reference numeral, and redundant description regarding the same part is omitted in principle.

<< First Embodiment >>
A first embodiment of the present invention will be described. FIG. 1 is a schematic block diagram of a motor drive system according to the first embodiment. The motor drive system of FIG. 1 includes a motor 1, a voltage conversion circuit 2, and a motor control device 3. FIG. 2 is a block diagram showing the internal configuration of the motor 1.

  The motor 1 is a three-phase permanent magnet synchronous motor, and includes a rotor 1r having a permanent magnet and three-phase armature windings (that is, U-phase, V-phase, and W-phase armature windings). And a stator 1s.

The voltage conversion circuit 2 is, for example, a PWM (Pulse Width Modulation) inverter. The PWM inverter as the voltage conversion circuit 2 converts a DC voltage supplied from a DC power supply (not shown) into a three-phase AC voltage using pulse width modulation, and applies the three-phase AC voltage to the motor 1. The three-phase AC voltage applied to the motor 1 includes a U-phase voltage v _u representing a voltage applied to the U-phase armature winding, a V-phase voltage v _v representing a voltage applied to the V-phase armature winding, And a W-phase voltage v _w representing a voltage applied to the W-phase armature winding. The total voltage applied to the motor 1, which is a combined voltage of the U-phase voltage v _u , the V-phase voltage v _v and the W-phase voltage v _w , is referred to as a motor voltage (motor terminal voltage), and is represented by the symbol v.

U-phase component of the current flowing through the motor 1 by application of the motor voltage v, V-phase component and a W-phase component, i.e. U-phase, the current flowing in the armature winding of the V-phase and W-phase, respectively, the U-phase current i _u , V phase current i _v and W phase current i _w . The total current flowing through the motor 1, which is a combined current of the U-phase current i _u , the V-phase current i _v and the W-phase current i _w , is referred to as a motor current (armature current) and is represented by the symbol i.

  3A and 3B are analysis model diagrams of the motor 1. FIG. 3A shows U-phase, V-phase, and W-phase armature winding fixed axes (hereinafter also referred to as U-phase axis, V-phase axis, and W-phase axis). 1 m is a permanent magnet provided on the rotor 1 r of the motor 1. The phase of the V-phase axis is advanced by 120 degrees in electrical angle with respect to the U-phase axis, and the phase of the W-phase axis is further advanced by 120 degrees in electrical angle with respect to the V-phase axis. In a rotating coordinate system that rotates at the same speed as the rotational speed of the magnetic flux generated by the permanent magnet 1m, the direction of the magnetic flux generated by the permanent magnet 1m is taken as the d-axis, and the q-axis is taken as a phase advanced 90 degrees from the d-axis by an electrical angle. In FIGS. 3A and 3B, the counterclockwise direction corresponds to the phase advance direction. The rotational speeds of the d-axis and q-axis are represented by ω. The rotational speed represented by ω is an angular speed in electrical angle.

  FIG. 3B shows the relationship between the U-phase axis, the V-phase axis, and the W-phase axis, and the α axis and β axis that are orthogonal to each other. The α axis coincides with the U-phase axis, and the β axis advances by 90 degrees in electrical angle with respect to the α axis. The U-phase axis, the V-phase axis, the W-phase axis, the α-axis, and the β-axis are fixed axes that are fixed regardless of the rotation of the rotor 1r. The α axis and the β axis are collectively referred to as an αβ axis, and a fixed coordinate system in which the α axis and the β axis are selected as coordinate axes is referred to as an αβ coordinate system.

Furthermore, state quantities are defined as follows.
The α-axis component and β-axis component of the motor voltage v are referred to as an α-axis voltage and a β-axis voltage, respectively, and are represented by symbols vα and vβ.
The α-axis component and β-axis component of the motor current i are referred to as α-axis current and β-axis current, respectively, and are represented by symbols iα and iβ.
A combination of magnetic fluxes interlinking three-phase armature windings is called an armature interlinkage magnetic flux and represented by the symbol φ. The α-axis component and β-axis component of the armature linkage flux φ are referred to as α-axis flux and β-axis flux, respectively, and are represented by symbols φα and φβ. The armature interlinkage magnetic flux φ corresponds to a combined magnetic flux of _a field magnetic flux (corresponding to Φa described later) by the permanent magnet 1m and an armature reaction magnetic flux by the motor current i. Since φ is a two-dimensional quantity, it is also called a magnetic flux vector.

The target values of vα, vβ, φα, and φβ that should be followed by vα, vβ, φα, and φβ are the α-axis voltage command value vα ^* , β-axis voltage command value vβ ^* , α-axis magnetic flux command value φα ^*, and referred to as the β-axis magnetic flux command value φβ ^*, _{v u,} v v and v _w to be followed, v _u, the target value of _v v and v _w, respectively, U-phase voltage command value v _u ^*, V-phase voltage It is called a command value v _v ^* and a W-phase voltage command value v _w ^* .

  In the present specification, for simplicity of description, only a symbol (such as iα) may be used to express a state quantity corresponding to the symbol. That is, in this specification, for example, “iα” and “α-axis current iα” or “α-axis current value iα” indicate the same thing.

  FIG. 4 is a detailed block diagram of the motor drive system of FIG. The motor drive system shown in FIG. 4 includes a power source 4 and a current sensor 10 in addition to the motor 1, the voltage conversion circuit 2, and the motor control device 3. The motor control device 3 in FIG. 4 includes each part referred to by reference numerals 31 to 40.

Each part in the motor control device 3 can freely use each value generated in the motor control device 3. Each part forming the motor drive system shown in FIG. 4 is calculated (or detected) by a predetermined control period and output as command values (ω ^* , T ^* , θ _S ^* , | φ ^* |, φα ^{*). , φβ *, vα *, vβ} *, v u *, v v * and v, including the _w ^*) or state the amount of (i _u, i _v, including iα, iβ, φα, φβ, θ S, the ω and T ) And calculate each value according to the latest command value or state quantity.

  In the motor drive system shown in FIG. 4, the motor 1 is driven and controlled by so-called direct torque control. When using an embedded magnet type motor or the like, harmonics often exist in the magnetic flux or inductance distribution. That is, for example, the magnetic flux of the permanent magnet interlinking the U-phase armature winding ideally draws a sinusoidal waveform with respect to the phase change of the rotor, but actually the waveform includes harmonics. As a result, the induced voltage caused by the rotation of the permanent magnet is also distorted. Similarly, the d-axis inductance and the q-axis inductance of the armature winding include harmonics. Such harmonics are known to cause torque ripple.

  When the d-axis current and the q-axis current are controlled to be constant by vector control, the generated torque is constant unless harmonics are included in the magnet magnetic flux or inductance distribution. Since the wave is included, the generated torque pulsates. On the other hand, in the direct torque control, the magnetic flux interlinking the armature windings of the stator is estimated, the torque is estimated from the estimated magnetic flux and the motor current, and the torque is directly controlled. Since the influence of harmonics is reflected in the estimated magnetic flux, the influence of harmonics is reflected in the estimation of torque. In direct torque control, since control is performed based on this estimated torque, there is an effect that torque ripple can be reduced even when harmonics exist in the magnetic flux or inductance distribution.

  In addition, responsiveness to torque command is good, rotor position information (d-axis phase information) necessary for performing dq coordinate conversion is unnecessary, and motor parameters such as inductance of armature winding are not required for control There are advantages to direct torque control. Hereinafter, the function of each part in FIG. 4 will be specifically described.

Current sensor 10 outputs an analog signal representative of the current value of the U-phase current i _u and the V-phase current i _v are supplied from the voltage conversion circuit 2 to the motor 1. Current detecting unit 31 consisting of the A / D converter detects the current value of the U-phase current i _u and the V-phase current i _v based on the output signal of the current sensor 10. Incidentally, using the current value of the detected i _u and i _v, according to the relational expression "i _w = -i _u -i _v", it is also possible to calculate the current value of the W-phase current i _w.

3-phase / 2-phase converting unit 32 performs coordinate transformation on the αβ-axis U-phase current value i _u and the V-phase current value i _v, calculates and outputs the α-axis current value iα and β-axis current value iβ .

The magnetic flux / torque estimation unit 33 (hereinafter, abbreviated as the estimation unit 33) is supplied from the α-axis current value iα and β-axis current value iβ given from the three-phase / two-phase conversion unit 32 and the voltage command unit 39. The α-axis magnetic flux φα and the β-axis magnetic flux φβ are estimated based on the α-axis voltage command value vα ^* and the β-axis voltage command value vβ ^* , and the phase θ _S of the magnetic flux vector φ is calculated based on the estimated φα and φβ. calculate. Further, the torque T generated in the motor 1 is estimated based on φα, φβ, iα and iβ.

In the space vector diagram shown in FIG. 5, a vector denoted by reference numeral 301 is a magnetic flux vector φ having φα and φβ as an α-axis component and a β-axis component. The phase θ _S is a phase in electrical angle and is a phase of the magnetic flux vector φ viewed from the α axis (U phase axis). As shown in FIG. 5, the field magnetic flux by the permanent magnet 1 m (corresponding to [Phi _a in FIG. 5) and the armature reaction magnetic flux by the motor current i (corresponding to Li in FIG. 5; L represents the inductance of the armature winding) Is equivalent to φ.

Specifically, φα, φβ, θ _S and T can be calculated according to the following formulas (A1), (A2), (A3) and (A4), respectively.

The integration of each right side in the equations (A1) and (A2) is the integration with respect to the time t, and the integration interval is from the time t = 0 which is the reference time to the current time.
R _a represents the resistance value per phase of the armature winding provided in the motor 1,
P _n represents the number of pole pairs of the motor 1,
φα _{| t = 0} and φβ _{| t = 0} represent the values of φα and φβ at time t = 0 (that is, initial values of φα and φβ), respectively.
_{Incidentally, R a, P n, φα} | t = 0 and Faibeta _| a _{t = 0,} can be previously determined at the design stage of the motor drive system.

A speed command generation unit (not shown) generates a rotation speed command value ω ^* for rotating the rotor 1r of the motor 1 at a desired rotation speed (electrical angular speed). The rotational speed command value ω ^* and the rotational speed ω of the motor 1 are given to the magnetic flux / torque command unit 34. The rotational speed ω can be obtained by detecting the rotational speed using an encoder or the like. The rotational speed ω may be estimated and calculated by differentiating the phase θ _S obtained by the estimation unit 33 with respect to time.

The magnetic flux / torque command unit 34 uses proportional integral control (hereinafter referred to as PI control) or the like so that the difference (ω ^* −ω) between the rotational speed command value ω ^* and the rotational speed ω converges to zero. Torque command value T ^* is calculated and output. The torque command value T ^* is a target value of the generated torque T of the motor 1 (in other words, a target value of the torque T estimated by the estimation unit 33).

In direct torque control, it is necessary to give a command value for the amplitude of the armature linkage flux simultaneously with the torque command value T ^* . The command value for this amplitude is represented by | φ ^* |. | Φ ^* | represents a target value of the magnitude of the magnetic flux vector φ formed from φα and φβ (that is, the amplitude of the armature linkage magnetic flux φ). | Φ ^* | is, for example, a function of the torque command value T ^* or the rotational speed ω. Therefore, for example, table data representing the relationship between | φ ^* | and T ^* or ω is stored in the motor control device 3 in advance, and | φ ^* | is obtained according to the table data based on T ^* or ω. be able to.

The subtracting unit 35 calculates a difference ΔT (= T ^* −T) between the torque command value T ^* and the estimated torque T and supplies the difference to the torque calculating unit 36.

The torque calculator 36 calculates a phase correction amount Δθ _S ^* corresponding to ΔT by performing PI control for converging ΔT to zero. The adding unit 37 obtains the sum (Δθ _S ^* + θ _S ) of the phase correction amount Δθ _S ^* and the phase θ _S calculated by the estimating unit 33, and uses the obtained sum as the phase command value θ _S ^* (= ( Δθ _S ^* + θ _S )).

Flux vector generation unit 38, | φ ^* | based on the theta _S ^*, calculates the α-axis magnetic flux command value Fa ^* and β-axis magnetic flux command value Faibeta ^* according to the following equation (A5) and (A6).

The voltage command unit (voltage command value generation unit) 39 is based on φα ^* , φβ ^* , φα and φβ from the magnetic flux vector generation unit 38 and the magnetic flux / torque estimation unit 33, and (φα ^* −φα) / t _S and ( α-axis voltage command value vα ^* and β-axis voltage command value vβ ^* are calculated using φβ ^* −φβ) / t _S (t _S is the time length of the control cycle as will be described later). In this calculation, iα and iβ from the three-phase / two-phase converter 32 can also be referred to (however, the manner in which iα and iβ are given to the voltage command unit 39 is not shown).

2-phase / 3-phase conversion unit 40 performs coordinate transformation of v? ^* And v? ^* Is the voltage command value on the αβ-axis on the fixed coordinate axes of a three-phase, v _u ^*, the v _v ^* and v _w ^* A three-phase voltage command value is calculated. The voltage conversion circuit 2 converts the DC voltage from the power supply 4 into a three-phase AC voltage composed of a U-phase voltage v _u , a V-phase voltage v _v and a W-phase voltage v _w according to the three-phase voltage command value. A current corresponding to the three-phase AC voltage is supplied to the armature winding to drive the motor 1.

The calculation in the motor drive system is actually performed based on the instantaneous value of each command value or state quantity discretized in a predetermined control cycle. Therefore, considering the case where the input value and the output value of each part in the motor control device 3 are discretized in the control cycle, the operation will be described. The time length of the control cycle is represented by t _S.

Consider a case where the reference time t = 0 belongs to the 0th control cycle and the current time belongs to the kth control cycle (k is a natural number). The command value or state quantity derived in the kth control cycle is represented by a symbol with [k]. Thus, k-th Fa derived in control _{period, φβ, θ S, T,} T *, ΔT, Δθ S *, θ S *, | φ * |, φα *, φβ *, vα *, vβ * is , Φα [k], φβ [k], θ _S [k], T [k], T ^* [k], ΔT [k], Δθ _S ^* [k], θ _S ^* [k], | ^{φ * | [k], φα} * [k], φβ * [k], vα * [k], is represented by vβ ^* [k]. The same applies to other than the k-th control cycle. Therefore, for example, the command value or the state quantity derived in the (k + 1) th control cycle is represented by a symbol with [k + 1]. [Delta] [theta] _S ^* [k] is, [Delta] T [k] and the k-th than the control period of the derived in the control cycle before ΔT (ΔT [k-1], etc.) and a [Delta] [theta] _S ^* which has been derived based on , Θ _S ^* [k] = θ _S [k] + Δθ _S ^* [k]. φα ^* [k] and φβ ^* [k] is θ _S ^* [k] and | φ ^* | is derived from ^{[k], vα * [k} ] and vβ ^* [k] is φα ^* [k] and φβ ^* It is derived from [k], φα [k] and φβ [k].

The motor control device 3 has φα [k], φβ [k], θ _S [k], T [k], according to the following formulas (B1) to (B6) corresponding to the formulas (A1) to (A6). φα ^* [k] and φβ ^* [k] can be calculated.

FIG. 6 shows the relationship between the magnetic flux vector and the voltage command vector. First, the relationship when so-called voltage saturation does not occur will be described. Now, the estimated Fa [k] and φβ [k] to the magnetic flux vector phi is called the estimated magnetic flux vector φ _est [k] having as α-axis component and β-axis component, Fa ^* [k] and φβ ^* [k] Is referred to as a magnetic flux command vector φ ^* [k]. Furthermore, a voltage vector having vα ^* [k] and vβ ^* [k] as an α-axis component and a β-axis component is referred to as a voltage command vector v ^* [k].

In FIG. 6, vectors 320 and 321 represent an estimated magnetic flux vector φ _est [k] and a magnetic flux command vector φ ^* [k], respectively. The magnetic flux command vector φ ^* [k] is a target of the magnetic flux vector φ in the next control cycle, that is, the (k + 1) th control cycle. In order to make the magnetic flux vector that was φ _est [k] in the kth control cycle coincide with the magnetic flux command vector φ ^* [k] in the (k + 1) th control cycle, the voltage command unit 39 sets the vector 322 in FIG. A corresponding voltage command vector v ^* [k] according to the equation “v ^* [k] = (φ ^* [k] −φ _est [k]) / t _S ” is generated. Since the product of v ^* [k] and t _S corresponds to the difference vector 323 between the estimated magnetic flux vector φ _est [k] and the magnetic flux command vector φ ^* [k], the voltage command vector is used in the k-th control cycle. By applying a voltage according to v ^* [k] to the motor 1, the magnetic flux vector in the (k + 1) th control period (ie, φ _est [k + 1]) ideally matches φ ^* [k]. It becomes like this.

However, when voltage saturation occurs during high speed rotation of the motor 1, the voltage according to the equation “v ^* [k] = (φ ^* [k] −φ _est [k]) / t _S ” is applied to the motor 1. Can not be applied. FIG. 7 shows the relationship between the magnetic flux vector and the voltage command vector when voltage saturation occurs. The voltage saturation is the magnitude of the voltage command vector v ^* to be generated in order to make the generated torque T of the motor 1 follow the torque command value T ^* (that is, the positive square root of (vα ^{* 2} + vβ ^{* 2} )). ^Indicates a state where the value exceeds a predetermined limit voltage value V _LIM, and when voltage saturation occurs, the magnitude of the voltage command vector v ^* is corrected to be equal to or less than the limit voltage value V _LIM . The corrected voltage command vector is represented by v ^* ′. The limit voltage value V _LIM is a voltage value that depends on the maximum voltage that can be output by the PWM inverter as the voltage conversion circuit 2, and is determined based on the magnitude of the DC voltage output from the power supply 4. In FIG. 7, an arc 330 represents a boundary line that distinguishes whether or not the voltage command vector is larger than the limit voltage value V _LIM .

When voltage saturation occurs, that is, the magnitude of the voltage command vector v ^* [k] according to the equation “v ^* [k] = (φ ^* [k] −φ _est [k]) / t _S ” Is larger than the limit voltage value V _LIM , the voltage command unit 39 converts the voltage command vector v ^* [k] corresponding to the vector 322 into the voltage command vector v ^* ′ corresponding to the vector 325 as shown in FIG. Correct to [k]. In order to distinguish and show the vectors 322 and 325, in FIG. 7, the vectors are shown slightly shifted, but the starting points of the vectors are the same. The direction of the voltage command vector v ^* ′ [k] is the same as the direction of the voltage command vector v ^* [k], but the magnitude of the voltage command vector v ^* ′ [k] is the same as the limit voltage value V _LIM. The The magnitude of the voltage command vector v ^* ′ [k] can be made smaller than the limit voltage value V _LIM .

When voltage saturation occurs, the voltage command unit 39 outputs the α-axis component and the β-axis component of the corrected voltage command vector v ^* ′ [k] as vα ^* and vβ ^* . Then, the magnetic flux vector changes by an amount corresponding to the vector 326 in FIG. 7 between the k and (k + 1) th control periods, and as a result, the magnetic flux vector in the (k + 1) th control period (ie, φ _est [k + 1]). Becomes a vector 327 and deviates from φ ^* [k]. This offset appears as a phase θ _S ^* [k] and phase difference theta _DIF between the phase θ _S [k + 1] of the estimated magnetic flux vector φ _est [k + 1] of the magnetic flux command vector φ ^* [k], the deviation is The larger the phase difference θ _DIF is, the larger the phase difference is.

As described above, the torque calculation unit 36 can obtain Δθ _S ^* from ΔT using PI control. However, when voltage saturation occurs, the magnitude of the voltage command vector is set to be equal to or less than the limit voltage value V _LIM. Correction is performed. If no measures corresponding to the execution of this correction are taken, the linearity of the torque control system (PI control linearity in the torque calculation unit 36) will be disrupted when voltage saturation occurs, and the torque control response will be degraded or undesired. Invite stabilization. Typically, for example, when ΔT having the same polarity that is not zero is continuously input to the torque control system, the integral value in the PI control becomes excessively large, and a so-called windup phenomenon is caused. When a windup phenomenon occurs in PI control, an operation delay due to accumulation of an integral value occurs, and an overshoot or the like of torque to be controlled occurs.

Considering this, the motor control device 3 varies the integral gain of the integrator in the PI control of the torque calculator 36 according to the phase difference θ _DIF . This function will be described in detail with reference to FIG. FIG. 8 is an internal block diagram of the torque control unit 50 included in the motor control device 3. The torque control unit 50 is formed including the torque calculation unit 36 and the subtraction unit 37 of FIG. 4 can be considered to include the PI control unit 51, the delay unit 52, the subtraction unit 53, and the multiplication unit 54 shown in FIG.

The PI control unit 51 calculates Δθ _S ^* so that ΔT converges to zero by PI control based on ΔT. As is well known, PI control is a combination of proportional control and integral control, and is a kind of control including integral control. The relationship between ΔT, which is an input value of the PI control unit 51, and Δθ _S ^* , which is an output value, is expressed by the following equation (C1). “S” on the right side of the formula (C1) is a Laplace operator. In the PI control of the PI control unit 51, the proportional gain is K _p and the integral gain is γ _i · K _i . The values of K _p and K _i can be fixed values set in advance, but the coefficient γ _i is a variable value (details will be apparent from the description below). The integrator 55 included in the PI control unit 51 derives and outputs an integral value corresponding to the second term “(γ _i · K _i / s) ΔT” on the right side of the equation (C1). Actually, the integrated value is derived by integrating the value obtained by multiplying ΔT discretized in the control cycle by the integral gain γ _i · K _i .

The adding unit 37 obtains the sum (Δθ _S ^* + θ _S ) of the phase correction amount Δθ _S ^* from the PI control unit 51 and the phase θ _S from the estimation unit 33, and uses the obtained sum as the phase command value θ _S ^*. Output as (= (Δθ _S ^* + θ _S )).

The delay unit 52 delays the output value of the addition unit 37 by one control cycle, and gives the delayed output value of the addition unit 37 to the subtraction unit 53. The subtracting unit 53 outputs a value obtained by subtracting θ _S from the estimating unit 33 from the output value of the delay unit 52 as the phase difference θ _DIF . In k-th control period, since the delay unit 52 theta _S ^* from the estimation unit 33 with [k-1] is output theta _S [k] is output, from the subtraction unit 53, the following formula (C2) Is output as the phase difference θ _DIF [k].

The multiplier 54 variably sets the coefficient γ _i using the phase difference θ _DIF from the subtractor 53 so that the coefficient γ _i in the equation (C1) becomes γ _i according to the following equation (C3). Here, K _A is a predetermined coefficient (where K _A > 0), and | θ _DIF | is an absolute value of θ _DIF . when setting the coefficient gamma _i in the k-th control period, a theta _DIF of formula _{(C3), θ DIF [k} ] is used. Since the coefficient γ _i is 1 when the phase difference θ _DIF is zero, the integral gain γ _i · K _i becomes a constant reference gain K _i . On the other hand, when the phase difference θ _DIF is not zero, as the absolute value of θ _DIF increases from 0, the coefficient γ _i decreases from 1 to 0, so the integral gain γ _i · K _i is also zero from the reference gain K _i. Decrease towards Even if the phase difference θ _DIF is not completely zero, the coefficient γ _i is set to 1 without using the equation (C3) if | θ _DIF | is equal to or less than a positive threshold value near zero. May be.

The coefficient K _A, the gain for the anti-windup, it is possible to reduce the overshoot of the torque T by a coefficient K _A greater than zero, reduction of the overshoot by adjusting the coefficient K _A The amount can be adjusted.

FIGS. 9A to 9D show waveforms of the estimated torque T and the coefficient γ _i when the torque command value T ^* is changed stepwise. FIG. 9A shows a waveform 370 representing a step change in the torque command value T ^* . In the graphs of FIGS. 9A to 9D, the horizontal axis corresponds to time. Of the waveforms 370 to 373 and 381 to 383 shown in FIG. 9A, only the waveforms 371 and 381 are extracted, and FIG. 9B shows only the waveforms 372 and 382 extracted. FIG. 9D shows only the waveforms 373 and 383 extracted from (c).

Waveform 371 and 381, respectively, a waveform and the waveform of the coefficient gamma _i of the estimated torque T when the coefficient K _A to zero. Actually, the coefficient K _A is set to a value larger than zero. Waveforms 372 and 373 are waveforms of the estimated torque T when the coefficient K _A is set to 10 and 50, respectively. Waveforms 382 and 383 are coefficients γ when the coefficient K _A is set to 10 and 50, respectively. It is a waveform of _i . 9A to 9D also show that the overshoot in the torque step response is reduced.

As can be seen from the above description, in the motor drive system according to the present embodiment, the generated torque T is controlled by the action of the torque control unit 50 using the integrator 55 (see FIG. 8). The integrator 55 calculates and outputs an integral value (γ _i · K _i · ΔT) corresponding to the difference ΔT between the generated torque T and the torque command value T ^*, and the torque controller 50 includes the integrator 55. The PI control for reducing the difference ΔT is executed using the output of. More specifically, the torque control unit 50 uses the integrator 55 to generate the phase command value θ _S ^* that is the target of the phase θ _S of the magnetic flux vector φ so that the difference ΔT is directed to zero.

On the other hand, the voltage command unit (voltage command value generation unit) 39 determines the phase command value θ _S ^* based on the PI control of the torque control unit 50, | φ ^* |, φα and φβ, and the limit voltage value V _LIM . Based on this, voltage command values vα ^* and vβ ^* that specify the voltage value to be applied to the motor 1 are generated, and the estimation unit 33 generates α of the voltage command values vα ^* and vβ ^* and the motor current (armature current) i. Based on the shaft component and the β-axis component, φα and φβ and the generated torque T are estimated.

Here, in principle, the voltage command unit 39 attempts to generate a voltage command value (vα ^* , vβ ^* ) necessary for linking the magnetic flux according to the phase command value θ _S ^* to the armature winding. Is a voltage command vector v ^* [k according to the magnitude of the required voltage command value (ie, the equation “v ^* [k] = (φ ^* [k] −φ _est [k]) / t _S ”). ] Is larger than the limit voltage value V _LIM , the actually generated voltage command values (vα ^* , vβ ^* ) are corrected to decrease. In other words, the voltage command vector v ^* [k] is changed to the voltage command vector v so that the magnitude of the voltage command vector having vα ^* and vβ ^* as the α-axis component and the β-axis component is limited to the limit voltage value V _LIM or less. ^* Correct to '[k]' (see Fig. 7).

When the voltage is restricted as described above, the magnetic flux vector φ _est estimated from the voltage command values (vα ^* , vβ ^* ) deviates from the target φ ^{* of the} magnetic flux vector according to the PI control. The torque controller 50 estimates the occurrence of voltage saturation from the phase difference θ _DIF representing this deviation, and variably sets the integral gain γ _i · K _i that is a parameter of the integrator 55 according to the phase difference θ _DIF .

  As a result, integral control suitable for occurrence of voltage saturation is possible, and torque control can be stabilized. As an example of stabilization, reduction of overshoot at the time of torque step response is expected. Furthermore, in order to realize such stabilization, the addition of a special circuit and complicated processing are not necessary.

By the way, in the motor control apparatus 3, the presence or absence of voltage saturation is detected from phase difference (theta) _DIF which is magnetic flux information (information regarding armature linkage magnetic flux). For example, in the torque control unit 50, if the absolute value of the phase difference θ _DIF is larger than a predetermined reference phase difference θ _TH, it can be determined that voltage saturation has occurred, and the absolute value of the phase difference θ _DIF If the reference phase difference θ _TH or less, it can be determined that voltage saturation has not occurred. The reference phase difference θ _TH is typically zero, but may be a positive value near zero. The motor control device 3 uses the detection result of the occurrence of voltage saturation for adjusting the integral gain.

  Although there is a method for detecting whether or not voltage saturation has occurred, the conventional detection method detects whether or not voltage saturation has occurred from the magnitude of the voltage command vector (see Patent Documents 1 and 2). . However, since the harmonic information as described above is included in the voltage information, the detection accuracy deteriorates due to the influence of the harmonic in the detection of the presence or absence of voltage saturation based on the voltage information. On the other hand, in the presence / absence detection of voltage saturation based on magnetic flux information employed in the present embodiment, the influence of harmonics is reduced. This is because the magnetic flux corresponds to an integrated voltage, so that higher-order harmonics are attenuated during the integration process, and as a result, the harmonics included in the magnetic flux information become smaller. Therefore, detection of occurrence of voltage saturation based on magnetic flux information is expected to have higher detection accuracy than that based on conventional voltage information.

<< Second Embodiment >>
A second embodiment of the present invention will be described. The technique described in the first embodiment can also be applied to a power generation system. In the second embodiment, a power generation system to which the technique described in the first embodiment is applied will be described. FIG. 10 is a detailed block diagram of the power generation system according to the second embodiment. The power generation system of FIG. 10 includes a generator 101, a voltage conversion circuit 102, and a generator control device 103, and also includes a voltage output unit 104 and a current sensor 10.

  The generator 101 is a three-phase permanent magnet synchronous generator, for example, an interior permanent magnet synchronous generator. The generator 101 includes a rotor including a permanent magnet and a stator including U-phase, V-phase, and W-phase armature windings, and has a configuration equivalent to that of the motor 1.

  In the motor drive system of FIG. 4, the electric power supplied from the power source 4 is converted into torque (rotational force) by the motor 1, whereas in the power generation system of FIG. 10, the torque generated by the generator 101 is the electric power. Is converted to As described above, the motor drive system of FIG. 4 and the power generation system of FIG. 10 have the same configuration and function, except that electric power is converted into torque or torque is converted into electric power. For this reason, the matters described for the motor 1 and the motor drive system (including definitions for terms and symbols) are all applied to the power generation system as long as there is no contradiction.

  However, since the generator is provided in the power generation system instead of the motor in the motor drive system, when the matters described for the motor and the motor drive system are applied to the second embodiment, the difference in terms (the term motor) And the term “generator” should be replaced. That is, in this application, first, “motor” in the description of the first embodiment should be read as “generator”, and the terms (including various state quantities and command values) described regarding the motor are used. In the second embodiment, it should be interpreted as the terms described for the generator.

Therefore, for example, in the second embodiment,
v is a generator voltage corresponding to a combined voltage of the U-phase voltage v _u , V-phase voltage v _v, and W-phase voltage v _w applied to the U-phase, V-phase, and W-phase armature windings in the generator 101. Represent,
i is a generator current (corresponding to a combined current of the U-phase current i _u , the V-phase current i _v and the W-phase current i _w flowing in the U-phase, V-phase and W-phase armature windings in the generator 101. Represents the armature current of the generator)
φ represents a combination of the interlinkage magnetic fluxes of the U-phase, V-phase, and W-phase armature windings in the generator 101. For example, in the second embodiment, the α-axis component and the β-axis component of the generator voltage v are represented by vα and vβ, respectively, and the α-axis component and the β-axis component of the generator current i are respectively iα and iβ. The α-axis component and β-axis component in the armature flux linkage φ of the generator 101 are represented by φα and φβ, respectively.
In the second embodiment,
The d-axis is a rotating shaft that faces the direction of the magnetic flux of the permanent magnet provided on the rotor of the generator 101,
ω represents the rotational speed of the d axis for the generator 101 (angular speed in electrical angle).

The generator 101 is connected to a windmill through a gear, for example. And the force which rotates a windmill with wind force is transmitted to the rotor of the generator 101 via a gear, a torque generate | occur | produces in a rotor, and a rotor rotates. The induced voltage (that is, the generator voltage v) generated in the generator 101 by the rotation of the rotor is converted from the generator 101 to the voltage conversion circuit 102 as a three-phase AC voltage composed of v _u , v _v, and v _w. Is output. The voltage conversion circuit 102 is, for example, a PWM (Pulse Width Modulation) converter, and the voltage values of v _u , v _v, and v _w forming this three-phase AC voltage are supplied from the 2-phase / 3-phase conversion unit 40, respectively. The three-phase AC voltage is converted into a DC voltage while performing pulse width modulation so as to match v _u ^* , v _v ^*, and v _w ^* . The DC voltage and the electric power based on the DC voltage are output from the voltage output unit 104. Thus, the power output from the voltage output unit 104 is based on the power generated by the generator 101.

  The generator control device 103 includes each part referred to by reference numerals 31 to 40. The configuration and function of each part in the generator control device 103 are the same as the configuration and function of each part in the motor control device 3 in FIG. 4, and the current sensor 10 in the power generation system is the same as that in FIG.

That is, the current sensor 10 outputs an analog signal representative of the current value of the U-phase current i _u and the V-phase current i _v flowing between the generator 101 and the voltage conversion circuit 102.
In the generator control device 103,
Current detector 11 detects a current value of the U-phase current i _u and the V-phase current i _v based on the output signal of the current sensor 10,
The three-phase / two-phase converter 32 converts i _u and i _v to iα and iβ,
Flux torque estimating unit 33 estimates the iα and iβ and v? ^* And the phase theta _S and torque T generated by the generator 101 of the magnetic flux vector φ with estimating the φα and φβ based and v? ^*.

A speed command generation unit (not shown) generates a rotation speed command value ω ^* for rotating the rotor of the generator 101 at a desired rotation speed (electrical angular speed). The desired rotation speed is set so that as much electric power as possible is drawn from the generator 101, for example. The magnetic flux / torque command unit 34 calculates T ^* so that the speed deviation (ω ^* −ω) converges to zero, while calculating | φ ^* |, and the torque calculation unit 36 calculates (T ^* −T). Δθ _S ^* is calculated by PI control so that the value converges to zero.

Flux vector generation unit 38, | φ ^* | and _{^{θ S * (= Δθ S *}} + θ S) calculated Fa ^* and Faibeta ^* is α-axis component and β-axis component of the magnetic flux command vector phi generator 101 and , the voltage command section 39 calculates the v? ^* and v? ^* by using the (φα ^* -φα) / t _S and _{^{(φβ * -φβ) / t S}} . When voltage saturation occurs, that is, the magnitude of the voltage command vector v ^* to be generated in order to make the generated torque T of the generator 1 follow the torque command value T ^* (ie, (vα ^{* 2} + vβ ^{* 2} )) When the positive square root) exceeds the predetermined limit voltage value V _LIM , the magnitude of the voltage command vector v ^* is corrected so as to be equal to or less than the limit voltage value V _LIM, and the α-axis component of the corrected voltage command vector And β axis components are calculated and output as vα ^* and vβ ^* . Limiting voltage value V _LIM is determined based on the component rating of the PWM converter as voltage conversion circuit 102, the rating of the part that receives the output of voltage output unit 104, or the like. vα ^* and vβ ^* are converted into a three-phase voltage command value composed of v _u ^* , v _v ^*, and v _w ^* by the two-phase / three-phase converter 40, and the three-phase voltage command value is sent to the voltage conversion circuit 102. Given.

The torque control unit formed including the torque calculation unit 36 and the subtraction unit 37 in the generator control device 103 is the same as the torque control unit 50 of the first embodiment (see FIG. 8), and between θ _S ^* and θ _S. of detecting the occurrence or non-occurrence of voltage saturation from the phase difference theta _DIF, during the occurrence of voltage saturation variably setting the integral gain γ _i · K _i in the PI control in accordance with the phase difference theta _DIF. For this reason, the same effect (stabilization of torque control etc.) as a 1st embodiment is acquired.

<< Deformation, etc. >>
The specific numerical values shown in the above description are merely examples, and as a matter of course, they can be changed to various numerical values. As modifications or annotations of the above-described embodiment, notes 1 to 6 are described below. The contents described in each comment can be arbitrarily combined as long as there is no contradiction.

[Note 1]
In the motor drive system and the power generation system described above, the U-phase current i _u and the V-phase current _iv are directly detected using the current sensor 10, but the direct current that flows between the voltage conversion circuit 2 and the power supply 4. These may be detected based on a direct current flowing between the current or voltage conversion circuit 102 and the voltage output unit 104.

[Note 2]
A method for deriving all the values to be derived including the above-described various command values (φα ^* , φβ ^*, etc.) and state quantities (φα, φβ, etc.) is arbitrary. That is, for example, they may be derived by calculation in the motor control device or the generator control device, or may be derived from preset table data.

[Note 3]
Part or all of the functions of the motor control device (or the generator control device) are realized by using software (program) incorporated in a general-purpose microcomputer, for example. When the motor control device (or generator control device) is realized using software, the block diagram showing the configuration of each part of the motor control device (or generator control device) represents a functional block diagram. Of course, it is possible to form the motor control device (or generator control device) not by software (program) but by hardware alone or by a combination of software and hardware.

[Note 4]
Each of the motor and the generator is a kind of rotating machine. Therefore, each of the motor control device and the generator control device can also be called a rotating machine control device.

[Note 5]
The motor control device 3 and the motor drive system according to the present invention are suitable for all electric devices using a motor. The motor control device 3 or the motor drive system described above can be mounted on any electric device that uses a motor, thereby improving controllability during voltage saturation (for example, improving controllability during high-speed rotation). be able to. Examples of the electric device include an electric vehicle (an electric automobile, an electric motorcycle, an electric bicycle, etc.), an air conditioner (such as an indoor or in-vehicle air conditioner), a washing machine, and a compressor (a compressor for a refrigerator). Etc.) and they are driven by the rotation of the motor.

[Note 6]
The following points should be noted in the present specification and drawings. Greek letters expressed as subscripts in the description of the expressions in the brackets (formula (A1), etc.) expressed as the number above, or so-called subscripts, are in parentheses for the function of the electronic application software. Other than expressions, it can be expressed as a standard character that is not a subscript. The difference between the subscript and the standard character in this Greek letter should be ignored.

DESCRIPTION OF SYMBOLS 1 Motor 2 Voltage conversion circuit 3 Motor control apparatus 33 Magnetic flux and torque estimation part 36 Torque calculation part 38 Magnetic flux vector generation part 39 Voltage command part 50 Torque control part 51 PI control part 52 Delay part 53 Subtraction part 54 Multiplication part 55 Integrator 101 Generator 102 Voltage conversion circuit 103 Generator control device 104 Voltage output unit

Claims (6)

In a motor control device including a torque control unit that controls the generated torque using an integrator so that generated torque of a motor having an armature winding follows a torque command value that is a target value of the generated torque.
The torque control unit is a command value that is to be generated to cause the generated torque to follow the torque command value, and a voltage command value that is a command value that specifies an applied voltage of the motor has a predetermined magnitude. A motor that adjusts a parameter of the integrator based on information on an interlinkage magnetic flux of the armature winding when a limit voltage value is exceeded and the magnitude is limited to the limit voltage value or less. Control device. The integrator outputs an integral value of a value corresponding to a difference between the generated torque and the torque command value;
The torque control unit uses the output of the integrator to execute control including integration control for reducing the difference,
A voltage command value generation unit that generates the voltage command value based on the control including the integration control and the limit voltage value, and the armature current that flows in the armature winding based on the voltage command value and the armature current. The motor control device further includes an estimation unit for estimating the flux linkage,
The motor control device according to claim 1, wherein the information on the flux linkage is generated based on an estimation result of the flux linkage by the estimation unit. The torque control unit generates a phase command value that is a target of the phase of the flux linkage on a predetermined coordinate system using the integrator so that the difference is reduced,
The voltage command value generation unit is configured to set the voltage command value necessary for interlinking a magnetic flux according to the phase command value to the armature winding, and the magnitude of the voltage command value is equal to or less than the limit voltage value. Generated under the restrictions
The information on the linkage flux is phase difference information indicating a difference between the phase on the predetermined coordinate system and the phase command value of the estimated linkage flux that is the linkage flux estimated by the estimation unit. The motor control device according to claim 2, wherein the motor control device is provided. The torque control unit generates the phase command value in accordance with an integral value obtained by multiplying the difference by an integral gain as the parameter, and as the difference represented by the phase difference information increases. The motor control device according to claim 3, wherein the integral gain is decreased. The estimation unit also estimates the generated torque based on the voltage command value and the armature current, and the torque control unit uses the estimated generated torque as the generated torque. The motor control apparatus in any one of Claims 1-4. A generator control device including a torque control unit that controls the generated torque using an integrator so that the generated torque of a generator having an armature winding follows a torque command value that is a target value of the generated torque In
The torque control unit has a predetermined voltage command value that is a command value that is to be generated to cause the generated torque to follow the torque command value and that is a command value that specifies an output voltage of the generator. The integrator parameter is adjusted based on information on the interlinkage magnetic flux of the armature winding when the limit voltage value is exceeded and the magnitude is limited to the limit voltage value or less. Generator control device.

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