JP2010166681A - Ac machine controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an AC machine controller which suppresses the switching loss and switching noise of an inverter while suppressing the reduction of efficiency. <P>SOLUTION: The AC machine controller is provided with: an inverter circuit 5 which is interposed between an AC machine 2 that functions as at least an AC electric motor and a DC power source 3, and converts an output of at least the DC power source 3 into AC power for power supply to the AC machine 2; a snubber circuit 4 which is interposed between the DC power source 3 and the inverter circuit 5, and interrupts the power supply by holding the output of the DC power source 3 in a zero-voltage state over a predetermined holding period of time; and a control section 10 which controls the snubber circuit 4 not to interrupt the power supply from the DC power source 3 when switching elements 8a-8f of the inverter circuit 5 are turned on, and to interrupt the output from the DC power source 3 when the switching elements 8a-8f are turned off. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to an AC machine control device that converts AC power supplied from a DC power source into AC and controls at least an AC machine that functions as an AC motor.

  In recent years, AC motors (alternating machines) have been used for various purposes such as driving hybrid vehicles. A battery including a fuel cell and a secondary battery is often used as a power source for supplying power to such an AC motor. Since such a battery is a DC power source, the power is AC converted via an inverter and supplied to the AC motor. The inverter is configured using a semiconductor switching element such as an IGBT (insulated gate bipolar transistor). The semiconductor switching element is switched by a control signal given from the inverter control circuit, and cuts off / conducts the DC power supply. Switching a semiconductor switching element in a situation where a voltage is applied to the semiconductor switching element or a current flows is called hard switching. General switching is this hard switching. An inverter based on hard switching has a simple configuration, but produces an excessive response during switching. This transient response causes large switching noise and switching loss. In general-purpose PWM (pulse width modulation) control, the number of times of switching per unit time is large, and accordingly, switching noise and switching loss increase.

  For hard switching, there is a switching method in which the semiconductor switching element is switched in a state where the voltage applied to the semiconductor switching element and the flowing current are substantially zero. This is called soft switching. The basic configuration of soft switching is to provide an LC resonance circuit outside the switching element. Resonance is generated in the LC resonance circuit by controlling the switching timing of the semiconductor switching element, and the voltage and current when the switching element is switched to zero. Thereby, the switching noise and switching loss accompanying switching of the semiconductor switching element are greatly reduced as compared with hard switching.

  Japanese Unexamined Patent Publication No. 2003-52190 (Patent Document 1) discloses a technique of an inverter control device using such soft switching. In an inverter that converts direct current into three-phase alternating current, a capacitor is provided in parallel with each switching element, and an inductor that forms a resonance circuit with the capacitor is provided in each arm of three phases. For example, a capacitor is provided in parallel with each of six IGBTs constituting the inverter, and a resonance inductor is provided in each of the three arms. When the load current of the motor to be controlled increases, the capacity and weight of the inductor increase, which hinders weight reduction and miniaturization. Therefore, Patent Document 1 proposes an inverter with a reduced inductor. That is, an auxiliary switching circuit is provided in the inverter, and the resonance circuit is configured by sharing one inductor by switching the path by the auxiliary switching circuit.

  The soft switching method is an excellent switching method in which switching noise and switching loss are greatly reduced as compared with hard switching. However, as described above, the soft switching method increases the circuit scale and complicates the control of the inverter. Therefore, in a hybrid vehicle including an electric motor that is supplied with electric power from a storage battery and a generator that regenerates electric power to the storage battery, it is difficult to apply the soft switching method because it may lead to complicated control. . In view of this, Japanese Patent Laid-Open No. 2006-352942 (Patent Document 2) proposes an apparatus that can easily reduce switching loss even in a system including a storage battery, an electric motor, and a generator, such as a hybrid vehicle. Has been.

  In this device, an inverter capable of regenerative operation connected to an electric motor and a rectifier circuit also having an inverter function connected to a generator are connected to a DC power source that is a rechargeable storage battery and two conductors (plus power line) , Minus power line). A soft switching circuit common to the inverter and the rectifier circuit is provided between the storage battery and the inverter, between the storage battery and the rectifier circuit, that is, between the two conductors. This soft switching circuit is provided to achieve soft switching when the switching elements of the inverter and the rectifier circuit are turned on. In the soft switching circuit, the voltage between the storage battery and the inverter and between the storage battery and the rectifier circuit, that is, between the two conductors connected to the DC power source, at the timing when the switching elements of the inverter and the rectifier circuit are turned on. By controlling to almost zero, ZVS (zero voltage switching) as soft switching is achieved. Note that a snubber capacitor is connected in parallel to each switching element of the inverter and the rectifier circuit in the apparatus of Patent Document 2. The electric charge stored in these capacitors is released by LC resonance with the inductor of the soft switch circuit.

JP 2003-52190 A (2-5, 9, 19-23 paragraphs, FIG. 1, 15 etc.) Japanese Patent Laying-Open No. 2006-352942 (paragraphs 11 to 24, 53 to 64, FIG.

  Not only hybrid vehicles but also reducing the size and weight of AC motors is beneficial for improving mounting efficiency and fuel consumption rate. On the other hand, when an AC motor is miniaturized and high torque is required for its output, nonlinearity of the d-axis inductance Ld and the q-axis inductance Lq in vector control (field oriented control: FOC) becomes strong, and control becomes difficult. FIG. 15 shows an example of inductance that changes nonlinearly. FIG. 15A shows the d-axis inductance Ld, and FIG. 15B shows the q-axis inductance Lq. As shown in FIG. 15, since Ld and Lq change nonlinearly depending on the rotation angle of the motor and the motor current, it becomes difficult to stabilize the target torque. It is possible to suppress the influence of inductance that varies nonlinearly by feedforward control to some extent. However, when the carrier frequency of the pulse width modulation that controls the switching element of the inverter is low, the response of the control is low, and pulsation (torque ripple) occurs in the output AC motor torque. Torque responsiveness in motor control is improved by increasing the carrier frequency of pulse width modulation. However, increasing the carrier frequency increases the number of times of switching per unit time. As the number of times of switching increases, switching noise and switching loss also increase.

  As described above, switching noise and switching loss can be suppressed by adding a soft switching circuit. However, when the carrier frequency increases, a state in which the voltage between the two conductors connected to the DC power source is controlled to almost zero, as in Patent Document 2, a so-called “instantaneous power failure” state frequently occurs. Efficiency will be reduced.

  The present invention was devised in view of the above problems, and an object thereof is to provide an AC machine control device capable of suppressing switching loss and switching noise of a switching element of an inverter while suppressing a decrease in efficiency. To do.

In order to achieve the above object, the characteristic configuration of the AC machine control device according to the present invention
An inverter circuit interposed between at least an alternating current machine functioning as an alternating current motor and a direct current power supply, converting at least the output of the direct current power supply into alternating current and supplying the alternating current machine;
A snubber circuit that is interposed between the DC power supply and the inverter circuit, and holds the output of the DC power supply in a substantially zero state for a predetermined holding time to cut off the supply of power;
When the switching element of the inverter circuit is turned on, the power supply from the DC power supply is not cut off, and when the switching element of the inverter circuit is turned off, the output from the DC power supply is cut off. And a control unit that controls the snubber circuit.

  An IGBT is often used as a switching element in an inverter circuit. The IGBT has a structure in which a P collector is added to the drain side of an N-channel vertical MOSFET (metal oxide semiconductor field effect transistor). By injecting holes from the P collector, the conductivity of the N base layer is modulated and the resistance is lowered. For this reason, compared with MOSFET, it is suitable for the use of a high voltage. On the other hand, since it takes time for the injected carriers to disappear, the turn-off time becomes long. As a result, a so-called tail current continues for a long time. A large collector loss (switching loss) occurs due to the overlap between the collector voltage restored to the power supply voltage side by the turn-off and the collector current continuously flowing as the tail current after the turn-off. At the time of turn-on, the collector current is delayed due to the influence of the motor (alternator) which is an inductance load. The collector current starts to flow with a delay with respect to the collector voltage dropped from the power supply voltage by the turn-on. Zero current switching (ZCS) can be realized with almost no overlap between collector voltage and collector current. Therefore, the collector loss at turn-on is very small compared to the collector loss at turn-off. For example, when soft switching is performed at the time of turn-on, charging / discharging is performed with respect to the inductor of the soft switching circuit.

  According to this configuration, the power supply from the DC power supply is not cut off when the switching element of the inverter circuit is turned on, and the output from the DC power supply is cut off when the switching element of the inverter circuit is turned off. In addition, the snubber circuit is controlled. In other words, soft switching is performed at the time of turn-off with a large switching loss to largely suppress the total loss, and normal hard switching is performed at the time of turn-on with a small switching loss to suppress a decrease in overall efficiency. By suppressing the switching loss and the overall efficiency, it is possible to increase the carrier frequency for pulse width modulation. As a result, the torque response related to the nonlinearity of the inductance can be improved, and the motor (alternator) can be downsized. Further, since the snubber circuit is not built in the inverter circuit but interposed between the DC power supply and the inverter circuit, it is not necessary to change the inverter circuit. That is, the inverter circuit can maintain a circuit configuration operable by normal hard switching, and the switching control signal provided to the switching element of the inverter circuit can also maintain the hard switching timing. Therefore, the control of the inverter circuit is not complicated.

  Further, when any trouble occurs in the snubber circuit, the inverter circuit can be normally operated by hard switching by stopping the shut-off function by the snubber circuit. That is, the snubber circuit itself can have a fail-safe function for the snubber circuit. In this case, since the effect of reducing the switching loss is lost, it is preferable to limit the output torque or rotational speed of the AC machine. On the other hand, when a malfunction occurs in the inverter circuit or the AC machine, the power supply to the inverter circuit and the AC machine can be cut off by using the snubber circuit cutoff function. That is, the snubber circuit can have a fail-safe function for the inverter circuit and the AC machine.

  Further, the control unit of the AC machine control device according to the present invention controls the snubber circuit so as to cut off the output from the DC power supply during a dead time set when the switching element of the inverter circuit is turned off. It is preferable.

  In the inverter circuit, a dead time is set in order to prevent the high-side switching element and the low-side switching element connected in series between the power source and the ground from being simultaneously turned on and short-circuiting the power source. That is, when one of the switching elements on the high side and the low side is turned off, if the other switching element is turned on at the same time, there is a possibility that both switching elements are temporarily turned on due to overlapping transition times. . In order to prevent this, a certain dead time is set after one switching element is turned off, and the other switching element is turned on after the dead time has elapsed. Therefore, no turn-on occurs during the dead time period, and there is no effect even if the supply of power to the inverter circuit is interrupted during the dead time period. Therefore, it is preferable that the snubber circuit is controlled so as to cut off the output from the DC power supply during the dead time. Further, if the power supply is cut off and restored by the snubber circuit in a short time, a large surge voltage may be generated due to a transient response. By providing a certain period during the dead time and performing the cutoff and return, the change in power per unit time can be moderated and the surge voltage can be suppressed.

Further, the snubber circuit of the AC machine control device of the present invention is:
A main switching element capable of interrupting reception of power supplied from the DC power supply;
In the subsequent stage of the main switching element, two switching elements, a high-side switching element and a low-side switching element, which are connected in series with the DC power source with an inductor interposed therebetween,
A resonant capacitor connected in parallel with the low-side switching element;
An output capacitor connected in parallel to a series circuit of the high-side switching element, the inductor, and the low-side switching element;
A regenerative diode connected in parallel to each of the switching elements and the output capacitor;
Comprising
The control unit performs switching control of the switching elements to control the snubber circuit,
Steady state of transmitting power supplied from the DC power source to the inverter circuit via the snubber circuit;
Following the steady state, the power supplied from the DC power supply is transmitted to the inverter circuit and the inductor is charged using the power to charge the inductor, and
Following the inductor charging state, the reception of power supplied from the DC power supply is cut off, and the voltage between the terminals of the output capacitor is lowered by the resonance between the inductor and the resonant capacitor, and the output of the snubber circuit A resonant state that reduces the voltage;
Following the resonance state, holding the voltage between the terminals of the output capacitor whose voltage has dropped to substantially zero, and holding the output voltage of the snubber circuit in a substantially zero state, and a zero voltage holding state,
Following the zero voltage holding state, recharging the output capacitor to increase the voltage across the output capacitor and returning the output voltage of the snubber circuit;
Following the charge recovery state, the power is regenerated from the resonant capacitor and the inductor to the DC power supply, and the reception of the power supplied from the DC power supply is resumed, and the power regeneration that returns to the steady state after the regeneration ends State,
It is suitable to control to.

  As described above, if the power supply is cut off and restored by the snubber circuit in a short time, a surge may occur due to a transient response. However, the change in power per unit time can be moderated by providing a certain period of time and performing the shutoff and return. That is, in the resonance state, the output voltage of the snubber circuit can be reduced relatively slowly. In addition, the output voltage of the snubber circuit can be gradually recovered in the charge recovery state. Then, in the vicinity of the time when the output voltage of the snubber circuit recovers, the supply of power from the DC power supply is resumed, so that a large inrush current does not occur. Further, by having the power regeneration state, energy that has conventionally been a surge or noise can be regenerated to the power source. Conventionally used noise filter circuits using capacitors and resistors consume energy absorbed by the capacitors. Therefore, even if the noise can be reduced, energy loss occurs, and as a result, the use efficiency of the power source is reduced. However, according to this structure, the energy stored in order to achieve the soft switching for reducing the noise in the inverter circuit is regenerated to the DC power source. Therefore, the snubber circuit can reduce noise and loss in the inverter circuit and improve the use efficiency of the DC power supply by regeneration.

Block diagram schematically showing a configuration example of an AC machine control device Waveform diagram schematically showing an example of transient response in hard switching Waveform diagram schematically showing an example of transient response when soft switching is performed at turn-off Explanatory diagram schematically showing the state transition at zero voltage switching The circuit diagram which shows typically an example of the logic circuit which produces | generates the gate drive signal which controls each IGBT of a snubber circuit Timing chart showing the relationship of gate drive signals for controlling each IGBT of the snubber circuit Waveform diagram showing operation of snubber circuit Explanatory drawing which shows typically operation | movement of the snubber circuit in each state shown in FIG. Explanatory drawing which shows typically operation | movement of the snubber circuit in each state shown in FIG. Explanatory drawing which shows typically operation | movement of the snubber circuit in each state shown in FIG. Explanatory drawing which shows typically operation | movement of the snubber circuit in each state shown in FIG. Explanatory drawing which shows typically operation | movement of the snubber circuit in each state shown in FIG. Explanatory drawing which shows typically operation | movement of the snubber circuit in each state shown in FIG. Explanatory drawing which shows typically operation | movement of the snubber circuit in each state shown in FIG. Graph showing characteristics of inductance ((a) d-axis, (b) q-axis)

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram schematically showing a configuration example according to an embodiment of an AC machine control device of the present invention. This AC machine control device is used to drive an AC motor as a power source of, for example, an electric vehicle or a hybrid vehicle. Some hybrid vehicles also include an alternator that functions as a generator based on the driving force of the internal combustion engine. Here, for the sake of easy explanation, an alternator control device that controls one alternator is provided. This will be described as an example.

  As shown in FIG. 1, the AC machine control device 1 receives supply of electric power from a battery 3 serving as a DC power supply, and drives and controls a motor 2 that functions as a power source for the vehicle. The motor 2 is an AC motor that functions as an AC motor or an AC motor and an AC generator. For example, the AC machine control device 1 can be configured to regenerate power to the battery 3 when the vehicle is braked.

  Between the battery 3 and the AC machine 2, an inverter circuit 5 that converts the output of the battery 3 into AC and supplies it to the motor 2 is provided. A converter circuit (not shown) that boosts the output voltage of the battery 3 may be further provided between the battery 3 and the inverter circuit 5. In this case, the battery 3 and the converter circuit correspond to the DC power source of the present invention.

  The inverter circuit 5 is configured by a three-phase bridge circuit. Two switching elements are connected in series between the input plus side (P) and the input minus side (N) of the inverter circuit 5, and this series circuit is connected in parallel in three lines. It is preferable to use IGBTs or MOSFETs as the switching elements 8a to 8f of the inverter circuit 5. In this embodiment, the case where IGBT is used is illustrated. That is, a bridge circuit in which a set of series circuits corresponds to each of the stator coil U phase, V phase, and W phase of the motor 2 is configured. In FIG. 1, IGBTs 8 a, 8 b, and 8 c are upper side switching elements corresponding to the U phase, the V phase, and the W phase, respectively. The IGBTs 8d, 8e, and 8f are lower-stage switching elements corresponding to the U phase, the V phase, and the W phase, respectively.

  The collectors of the IGBTs 8a, 8b, 8c on the upper side of each phase are connected to the input plus side P of the inverter circuit 5, and the emitters are connected to the collectors of the IGBTs 8d, 8e, 8f on the lower side of each phase. Further, the emitters of the IGBTs 8d, 8e, 8f on the lower side of each phase are connected to the input minus side N (ground) of the inverter circuit 5. The middle points (IGBT connection points) 9u, 9v, 9w of the series circuit of IGBTs (8a, 8d), (8b, 8e), (8c, 8f) of each phase are the U phase of the motor 2, V It is connected to the phase and W phase stator windings, respectively. The gates of the IGBTs 8a to 8f are connected to the control unit 10 and are individually controlled for switching. The control unit 10 is configured with a logic circuit such as a microcomputer as a core. A drive signal input to the gate of an IGBT or MOSFET that switches a high voltage requires a voltage higher than that of a general logic circuit, and thus is input via a driver circuit. In FIG. 1, for simplification, this driver circuit is omitted as being included in the control unit 10.

  The controller 10 supplies the motor 2 with a three-phase AC drive current by performing PWM control on the IGBTs 8 a to 8 f based on the target torque and the rotation speed for the motor 2. Thereby, the motor 2 performs powering according to the rotation speed and the target torque. The motor 2 functions as a generator, and the regeneration when the inverter circuit 5 receives power from the motor 2 side is the same as during powering. The control unit 10 converts the generated power into direct current by performing PWM control on the IGBTs 8a to 8f based on the target torque and the rotation speed for the motor 2. It is also possible to simply rectify using a flywheel diode (regenerative diode) connected in parallel to the IGBTs 8a to 8f.

  Further, the control unit 10 performs feedback control based on the rotation speed and the motor current. As shown in FIG. 1, a current sensor 13 is provided to measure the drive current supplied to the U-phase, V-phase, and W-phase stator windings of the motor 2. The control value received by the current sensor 13 is received by the control unit 10 and used for feedback control. In this example, a configuration is shown in which the current of all three phases is measured. However, since the three phases are in an equilibrium state and the sum of instantaneous current values is zero, the current of only two phases is measured and the control unit In 10, the remaining one-phase current may be obtained by calculation.

  The motor 2 is provided with a rotation detection sensor 11 such as a resolver, and detects the rotation angle (mechanical angle) of the rotor of the motor 2. The rotation detection sensor 11 functions as a rotation detection unit alone or in cooperation with the control unit 10. The rotation detection sensor 11 is set according to the number of poles (the number of pole pairs) of the rotor, and can also convert the rotation angle of the rotor into an electrical angle θ and output a signal corresponding to the electrical angle θ. The control unit 10 calculates the rotation speed (angular velocity ω) of the motor 2 and the control timing of each IGBT 8a to 8f of the inverter circuit 5 based on this rotation angle.

  As described above, the inverter circuit 5 is configured using semiconductor switching elements such as IGBTs 8a to 8f, and is switched by the drive signal supplied from the control unit 10 to cut off / conduct the DC power supply P-N. Switching the switching element in a situation where a voltage is applied to the switching element or a current flows is called hard switching. General switching is this hard switching. The inverter circuit based on hard switching has a simple configuration as shown in FIG. 1, but causes an excessive response during switching.

FIG. 2 is a waveform diagram schematically showing an example of an excessive response in hard switching. In FIG. 2, time t1 indicates the time when the gate drive signal of the IGBT is turned on, and time t2 indicates the time when the gate drive signal is turned off. At this time, the potential difference between the input plus side P and the input minus side N of the inverter circuit 5 is the voltage V1 of the DC power supply (battery 3 or the output of the converter). In FIG. 2, the collector voltage (collector-emitter voltage) V _CE and the collector current I _C are overlapped for easy understanding. The vertical axis is the collector voltage V _CE voltage value, the collector current I _C is a current value.

At the turn-on time, the collector voltage immediately drops from time t1, but the collector current I _C is delayed in rising due to the influence of the motor 2 which is an inductive load. In other words, the collector current I _C starts to flow with a delay with respect to the collector voltage V _CE that drops from the power supply voltage V1 due to the turn-on. So-called current zero switching (ZCS) can be realized with almost no overlap between the collector voltage V _CE and the collector current I _C. As a result, the collector loss P _C1 generated at turn-on is relatively small.

At the time of turn-off, the collector voltage V _CE immediately rises from the time t2, but the tail current having a long base continues in the collector current I _C. The IGBT has a structure in which a P collector is added to the drain side of the N-channel vertical MOSFET. By injecting holes from the P collector, the conductivity of the N base layer is modulated and the resistance is lowered. For this reason, compared with MOSFET, it is suitable for a high voltage application. However, since it takes time for the injected carriers to disappear, the turn-off time becomes long, and the tail current continues for a long time. Then, the collector voltage V _CE restored to the power supply voltage V1 side by the turn-off and the collector current I _C that continuously flows as the tail current after the turn-off overlap for a long time. As a result, as shown in FIG. 2, a larger collector loss P _C2 is generated than at the time of turn-on. In addition, since a large surge voltage is generated in the collector voltage V _CE while the tail current continues, the energy of EMI (electro-magnetic interference) noise, which is said to be caused by dV / di, also increases.

  As described above, in hard switching, excessive switching causes large switching loss and switching noise. In general-purpose PWM control, the number of times of switching per unit time is large, and accordingly, switching noise and switching loss increase. Therefore, the control unit 10 reduces the switching loss and the switching noise by performing soft switching at the time of turn-off that causes a large switching loss and switching noise. Specifically, the AC machine control device 1 is interposed between the DC power supply (battery 3 or the output of the converter) and the inverter circuit 5, and the voltage of the DC power supply is substantially zero for a predetermined holding time T4. A snubber circuit 4 is provided that keeps the state and cuts off the supply of power. And the control part 10 controls the snubber circuit 4 so that the output from DC power supply may be interrupted when IGBT8a-8f of the inverter circuit 5 turns off. Note that the control unit 10 controls the snubber circuit 4 so as not to cut off the supply of power from the DC power source when the IGBTs 8a to 8f of the inverter circuit 5 are turned on so as not to reduce the overall efficiency.

By controlling the snubber circuit 4 in this way by the control unit 10, the transient response at the time of switching of the IGBT is as shown in FIG. At the time of turn-on, a switching loss P _C1 occurs as in FIG. 2, but at the time of turn-off, the input voltage P−N of the inverter circuit 5 is controlled to zero over the period T4. As a result, the collector voltage V _CE does not rise immediately due to turn-off, but rises with a certain delay. As a result, the period in which the collector voltage V _CE overlaps with the collector current I _C that continuously flows as a tail current after turn-off is shortened. As shown in FIG. 3, at the time of turn-off, a large collector loss P _C2 as shown in FIG. 2 does not occur. That is, zero voltage switching (ZVS) is achieved.

Further, by the operation of the snubber circuit 4, the input voltage of the inverter circuit 5 after the turn-off is gradually increased, the surge voltage generated at the time of the rise of the collector voltage V _CE is small and becomes a lower frequency. Therefore, EMI noise which is said to be caused by dV / di is reduced. Further, since the collector voltage V _CE rises after the tail current is almost settled and a surge voltage is generated, the energy of EMI noise is also reduced.

  The detailed operation of the snubber circuit 4 will be described below. As shown in FIG. 1, the snubber circuit 4 includes four switching elements 6m, 6a, and 6b. In this example, the switching elements 6m, 6a, 6b are configured by IGBTs. The IGBT 6m is a switching element that directly supplies or cuts off a DC power source as a main switch. Hereinafter, the IGBT 6m is appropriately referred to as a main IGBT (main switching element). IGBT 6a and IGBT 6b are connected in series via inductor L1. The IGBT 6a is an upper switching element, and the collector terminal is connected to the emitter terminal of the IGBT 6a as a main switch. Hereinafter, the IGBT 6a is appropriately referred to as a high side IGBT (high side switching element). The IGBT 6b is a lower-stage switching element, and the emitter terminal is connected to the ground side (the negative side of the DC power supply). Hereinafter, the IGBT 6b is appropriately referred to as a low-side IGBT (low-side switching element).

  A resonance capacitor C1 is connected in parallel to the low-side IGBT 6b. That is, a parallel circuit of the low side IGBT 6b and the resonance capacitor C1, a series circuit of the inductor L1 and the high side IGBT 6a is formed. A capacitor C2 and a diode D1 are connected in parallel with this series circuit. In addition, each IGBT6a, 6b, 6m is each provided with the flywheel diode (regenerative diode) 7a, 7b, 7m in parallel, respectively.

  FIG. 4 is an explanatory diagram schematically showing state transition at the time of zero voltage switching. Vout indicates the output of the snubber circuit 4. Periods T1 and T9 are in a steady state. As will be described in detail later, in the steady state, the main IGBT 6m is turned on, and the output Vout of the snubber circuit 4 is substantially the voltage V1 of the battery 3. The period T2 is a charging state of the inductor L1. The period T3 is a resonance state between the inductor L1 and the resonance capacitor C1. The period T4 is a zero voltage holding state in which the output Vout of the snubber circuit 4 is held at zero. Period T5 is a state in which the output capacitor C2 is restored to charge. The period T6 and the period T7, that is, the period T8 are power regeneration states in which regeneration to the battery 3 is performed. The period T6 is a first power regeneration state in which the resonance capacitor C1 and the inductor L1 are regenerated to the battery 3. The period T7 is a second power regeneration state in which regeneration is performed from the inductor L1 to the battery 3.

  Each of these states is realized by switching control of the main IGBT 6m, the high side IGBT 6a, and the low side IGBT 6b at a predetermined timing by the control unit 10. FIG. 5 is a circuit diagram schematically illustrating an example of a logic circuit that generates a gate drive signal for controlling each IGBT of the snubber circuit 4. FIG. 6 is a timing chart showing the relationship of gate drive signals that control each IGBT of the snubber circuit 4. Here, the TRG signal is a signal indicating basic timing for controlling the snubber circuit 4. The TRG signal is also the gate drive signal G_hi of the high-side IGBT 6a as shown in FIGS. As shown in FIGS. 5 and 6, the gate drive signal G_low of the low-side IGBT 6b is generated by delaying the TRG signal by a delay element Td by a delay element and differentiating the rising edge of the TRG signal. The gate drive signal G_main of the main IGBT 6m is generated by taking an exclusive OR of the gate drive signal G_hi and the gate drive signal G_low and further taking a negative logic thereof.

FIG. 7 is a waveform diagram schematically showing the operation of the snubber circuit 4 based on the operation of each IGBT of the snubber circuit. The third waveform from the top is a waveform related to the main IGBT 6m, and “main” is indicated in parentheses. The next three waveforms are related to the high-side IGBT 6a, and are indicated as “hi” in parentheses. The next three waveforms are related to the low-side IGBT 6b, and are indicated as “low” in parentheses. The waveform at the bottom is the waveform of the output Vout of the snubber circuit 4. The waveforms related to each IGBT are a gate drive signal, a collector voltage (collector-emitter voltage) V _CE , and a collector current I _C from the top. Although referred to as collector current I _{C for} convenience, it is not a collector current in a strict sense. Including the current flowing through the flywheel diodes (7a, 7b, 7m) and refers to the current flowing through the IGBT module configured and a IGBT and flywheel diodes and the collector current I _C.

8 to 14 are explanatory diagrams schematically showing the operation of the snubber circuit 4 in each state transition state shown in FIG. 4, and arrows in the circuit diagrams schematically show a rough flow of typical currents. It shows. 8 to 14 show equivalent circuits, and the load corresponds to the inverter circuit 5 and the motor 2. For example, a computer simulation can be implemented by simulating a load with a resistance using such an equivalent circuit. Here, the collector current I _C of the main IGBT6m a circuit current in the steady state is a constant value. FIG. 7 shows a waveform diagram in such an equivalent circuit. In FIG. 7, the collector current I _C of each IGBT indicates the current flowing through each IGBT module as described above. Specifically, shows the node A in FIGS. 8 to 14, B, the current in M as the collector current I _C.

  FIG. 8 shows a steady state corresponding to the periods T1 and T9 in FIG. FIG. 9 shows the state of charge of the inductor L1 corresponding to the period T2 in FIG. FIG. 10 shows a resonance state between the inductor L1 and the resonance capacitor C1 corresponding to the period T3 in FIG. FIG. 11 shows a zero voltage holding state corresponding to the period T4 in FIG. FIG. 12 shows a state in which charging of the output capacitor C2 is restored corresponding to the period T5 in FIG. FIG. 13 shows a first power regeneration state corresponding to the period T6 of FIG. FIG. 14 shows a second power regeneration state corresponding to the period T7 in FIG.

Hereinafter, the operation of the snubber circuit 4 will be described with reference to FIGS. A period T1 corresponding to time t00 to time t10 in FIG. 7 is a steady state. Therefore, as shown in FIGS. 7 and 8, the main IGBT 6m is on and the other IGBTs 6a and 6b are off. Since the main IGBT 6m is on, the collector voltage V _CE of the main IGBT 6m is almost zero, and the collector current I _C is I _N which is a steady value. The steady value I _N of the collector current I _C is, for example, 300 A in the simulation.

When the gate drive signal G_hi of the high-side IGBT 6a and the gate drive signal G_low of the low-side IGBT 6b become active at time t10, the state shifts from the steady state to the charged state. That is, the high side IGBT 6a and the low side IGBT 6b are turned on, and a current flows through a series circuit of the high side IGBT 6a, the inductor L1, and the low side IGBT 6b as shown in FIG. As shown in FIG. 7, a large amount of current is drawn from the battery 3 to supply the charging current to the inductor L1, and the collector current I _C of the main IGBT 6m rapidly increases after time t10. The rise from the steady value I _N appears as the collector current I _C of the high-side IGBT 6a and the low-side IGBT 6b. This increased current is the charging current of the inductor L1, and the charging state of the inductor L1 is in the period T2 from time t10 to time t30.

When the low-side IGBT 6b is turned off at time t30 and the main IGBT 6m is turned off, the resonance state is entered. The high-side IGBT 6a is kept on. Since the low-side IGBT 6b is turned off, the inductor L1 and the resonance capacitor C1 resonate as shown in FIGS. 7 and 10, and a current flows from the resonance circuit to the output capacitor C2. Since the main IGBT 6m is in the off state, the current to the output capacitor C2 is a closed loop. For this reason, the voltage across the terminals of the output capacitor C2 gradually decreases, and finally decreases until the diode D1 connected in parallel with the output capacitor C2 becomes conductive. During this time, as shown in FIG. 7, the voltage on the emitter side of the main IGBT 6m decreases, so that the collector voltage V _CE finally rises to the steady voltage V _N. The steady voltage V _N is the voltage V1 of the DC power supply in this example, and is 300 V, for example, in the simulation.

When the collector voltage V _CE of the main IGBT6m rises to a steady voltage V _N, the output voltage Vout of the snubber circuit 4 gradually decreases to zero from constant voltage V _N. That is, the output Vout of the snubber circuit 4 decreases to zero in the resonance state in the period T3 from time t30 to time t34.

Period in which the inductor L1 emits current, i.e., a period in which the collector current I _C of the high-side IGBT6a indicates a positive value in FIG. 7, the period T4 of the time t34~ time t48 is a voltage zero holding state . The collector voltage V _CE of the low-side IGBT 6b in the off state rises greatly exceeding the steady voltage V _N in the resonance state and the zero voltage holding state from time t30 to time t48. The collector voltage V _CE of the low-side IGBT 6b reaches its maximum value at time t48 when the inductor L1 has completely discharged current.

After time t48, discharging from the resonant capacitor C1 is started, and charging to the output capacitor C2 is started. As shown in FIGS. 7 and 12, the direction of the current flowing through the high-side IGBT 6a is reversed. This is continued until the recharging of the capacitor C2 is completed. When the recharging of the capacitor C2 is completed, the output voltage Vout of the snubber circuit 4 rises and returns to the steady voltage V _N. As shown in FIG. 7, in this example, recharging is completed at time t60. A period T5 from time t48 to time t60 is in a charging return state. At this timing when the recharging is completed, the high side IGBT 6a is turned off and the main IGBT 6m is turned on. Since the collector voltage V _CE of the main IGBT 6m gradually returns to the steady voltage V _N , it can smoothly return from the zero voltage control for soft switching.

As shown in FIG. 7, at time t60, the resonant capacitor C1 is not completely discharged, and the collector voltage V _CE of the low-side IGBT 6b is not zero. Also, the collector current I _C of the high side IGBT 6a is not zero. During a period T6 from time t60 to time t64 when the collector voltage V _CE of the low-side IGBT 6b becomes zero, power is regenerated from the resonant capacitor C1 to the battery 3 via the flywheel diodes 7a and 7m, as shown in FIG. The first power regeneration state.

  Even after the resonant capacitor C1 is completely discharged at time t64, energy is still accumulated in the inductor L1. A period T7 from time t64 to time t78 is a second power regeneration state in which power is regenerated from the inductor L1 via the flywheel diodes 7a, 7m, and 7b, as shown in FIG.

  By having the power regeneration state in the period T8 from the time t60 to the time t78, it is possible to regenerate energy that has conventionally been surge or noise to the power source. Conventionally used noise filter circuits using capacitors and resistors consume energy absorbed by the capacitors. Therefore, even if the noise can be reduced, energy loss occurs, and as a result, the use efficiency of the power source is reduced. However, as described above, the snubber circuit 4 regenerates energy stored in order to achieve soft switching for reducing noise in the inverter circuit 5 to the DC power source. Therefore, the snubber circuit 4 can reduce noise and loss in the inverter circuit 5 and improve the use efficiency of the DC power supply by regeneration.

Further, as apparent from FIG. 7, the collector voltage V _CE and the collector current I _C of each IGBT of the snubber circuit 4 do not overlap. Therefore, the snubber circuit 4 is configured with low loss.

  The TRG signal for operating the snubber circuit 4 may be generated based on the gate drive signals of the IGBTs 8a to 8f of the inverter circuit 5 at a timing at which a voltage zero state can be realized in the dead time of the IGBT. That is, the TGR signal can be generated based on the logical sum of the gate drive signals of the IGBTs 8a or 8f of the inverter circuit 5. The conventional control of the inverter circuit 5 is left as it is, and the TGR signal is generated based on the control timing of the IGBTs 8a to 8f of the inverter circuit 5, and the soft switching at the turn-off time of the IGBTs 8a to 8f can be achieved based on the TRG signal. . In order to apply the soft switching, the control of the inverter circuit 5 is not changed, and the control of the snubber circuit 4 for the soft switching can be performed using the control of the inverter circuit 5. Therefore, soft switching at turn-off can be realized very easily.

  Further, when an unexpected situation such as failure of the snubber circuit 4 occurs, the inverter circuit 5 can be operated as usual by always turning on the main IGBT 6m if the main IGBT 6m is safe. Is possible. Since switching loss cannot be suppressed, the speed and torque of the motor 2 are limited to some extent, but a device driven by the motor 2 such as a vehicle can be operated at a minimum. On the other hand, when a failure occurs in the inverter circuit 5 or the motor 2, the power supply can be cut off by always turning off the main IGBT 6m. Various fail safes can be realized by the snubber circuit 4.

  According to the computer simulation by the inventor, it was confirmed that by using such a snubber circuit 4, the loss in the inverter circuit 5 can be reduced by 93% at the peak of switching. Moreover, it was confirmed that the average loss in the inverter circuit 5 can be reduced by 64%.

  By reducing the loss, for example, the modulation frequency (carrier frequency) of the pulse width modulation when the inverter circuit 5 is PWM-controlled can be increased. As described above, when the motor is downsized and high torque is obtained, the non-linearity of the d-axis and L-axis inductances becomes strong as shown in FIG. Even if feedback control is performed based on the detection results of a current sensor, a rotation sensor, etc., if the carrier frequency of pulse width modulation is low, the followability is poor, and there is a limit to improving the response. However, increasing the carrier frequency of the pulse width modulation leads to an increase in the number of switching per unit time, and the increase in the number of switching is directly related to the switching loss, which is difficult to realize. However, as described above, if switching loss can be greatly reduced, the carrier frequency of pulse width modulation can be increased. As a result, the motor can be reduced in size, increased in torque, and improved in response.

  Further, since the response is improved by increasing the carrier frequency of the pulse width modulation, the torque ripple of the motor 2 and the ripple component of the current flowing through the stator coil are also reduced. Therefore, a capacitor provided for absorbing and smoothing the ripple current can be reduced in capacity. For example, a large electric field capacitor can be changed to a smaller film capacitor or the like.

Sleeve Reduction of switching loss also contributes to suppression of heat generation. Generally, a loss in a semiconductor is released as heat. Therefore, the amount of heat generated by the inverter circuit 5 is reduced by reducing the switching loss. As a result, the heat capacity of the heat sink added to the inverter circuit 5 can be reduced. Conventionally, when cooling by water cooling or oil cooling, there is a possibility that it can be replaced by air cooling, leading to cost reduction.

  As described above, according to the present invention, it is possible to provide an AC machine control device that can suppress switching loss and switching noise of a switching element of an inverter while suppressing a decrease in efficiency.

1: AC machine control device 2: Motor (alternator)
3: Battery (DC power supply)
4: Snubber circuit 5: Inverter circuits 8a to 8f: IGBT (switching element of inverter circuit)
10: Control unit

Claims (3)

An inverter circuit interposed between at least an alternating current machine functioning as an alternating current motor and a direct current power supply, converting at least the output of the direct current power supply into alternating current and supplying the alternating current machine;
A snubber circuit that is interposed between the DC power supply and the inverter circuit, and holds the output of the DC power supply in a substantially zero state for a predetermined holding time to cut off the supply of power;
When the switching element of the inverter circuit is turned on, the supply of power from the DC power supply is not interrupted, and when the switching element of the inverter circuit is turned off, the output from the DC power supply is interrupted. An AC machine control device comprising: a control unit that controls the snubber circuit.   2. The AC machine control device according to claim 1, wherein the control unit controls the snubber circuit so as to cut off an output from the DC power supply during a dead time set when a switching element of the inverter circuit is turned off. . The snubber circuit is
A main switching element capable of interrupting reception of power supplied from the DC power supply;
In the subsequent stage of the main switching element, two switching elements, a high-side switching element and a low-side switching element, which are connected in series with the DC power source with an inductor interposed therebetween,
A resonant capacitor connected in parallel with the low-side switching element;
An output capacitor connected in parallel to a series circuit of the high-side switching element, the inductor, and the low-side switching element;
A regenerative diode connected in parallel to each of the switching elements and the output capacitor;
Comprising
The control unit controls the switching of the switching elements to control the snubber circuit.
Steady state of transmitting power supplied from the DC power source to the inverter circuit via the snubber circuit;
Following the steady state, the power supplied from the DC power source is transmitted to the inverter circuit and the inductor is charged using the power to charge the inductor, and
Following the inductor charging state, the reception of power supplied from the DC power supply is cut off, and the voltage between the terminals of the output capacitor is lowered by the resonance between the inductor and the resonant capacitor, and the output of the snubber circuit A resonant state that reduces the voltage;
Following the resonance state, holding the voltage between the terminals of the output capacitor whose voltage has dropped to substantially zero, and holding the output voltage of the snubber circuit in a substantially zero state, and a zero voltage holding state,
Following the zero voltage holding state, recharging the output capacitor to increase the voltage across the output capacitor and returning the output voltage of the snubber circuit;
Following the charge recovery state, the power is regenerated from the resonant capacitor and the inductor to the DC power supply, and the reception of the power supplied from the DC power supply is resumed, and the power regeneration that returns to the steady state after the regeneration ends State and
The alternator control device according to claim 1 or 2, which is controlled to the above.

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