JP2001197737A - High voltage power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce dimensions of a high voltage power supply circuit, simplify the circuit and reduce the peak of a current applied to switching devices to reduce a switching noise, a switching loss and, further, a power loss in the ON period of the switching device. SOLUTION: After a DC voltage supplied from a DC input power supply 1 is stabilized by a regulator 2, an isolation high voltage transformer 4 is driven by a pair of switching devices 3 of a push-pull inverter to obtain a DC high voltage output with a rectifier circuit 5 and a filter capacitor 7 of the secondary circuit of the high voltage transformer 4, and the DC high voltage is supplied to a load 8. A resonance circuit which resonates partially with the current applied to the switching devices 3, at an instant when the current is turned on, is composed of the leakage inductance 9 and a distributed capacitance 10 between secondary windings. Further, the leakage inductance 9 or the distributed capacitance 10 between the secondary windings is prescribed so as to obtain the optimum value of the current applied to the switching device 3.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a high-voltage power supply circuit, and more particularly to a push-pull inverter type high-voltage power supply using an insulated high-voltage transformer for deriving an output corresponding to a DC input voltage such as a commercial full-wave rectified voltage or a DC power supply. Circuit.

[0002]

2. Description of the Related Art FIG. 8 is a block diagram showing a conventional example of a power supply circuit having a push-pull inverter. In the power supply circuit shown in FIG. 1, for example, a DC voltage supplied from a DC input power supply 1 obtained by rectifying and smoothing a DC power supply or a commercial AC power supply is stabilized by a regulator 2. Thereafter, a DC high-voltage output is obtained by a smoothing circuit composed of a rectifier circuit 5, a choke coil 6, and a smoothing capacitor 7 via a step-up insulating high-voltage transformer 4 driven by a pair of switching elements 3 of the push-pull inverter. This output is supplied to the load 8.

Since the voltage fluctuation of the DC input power supply 1 obtained by rectifying and smoothing a DC power supply or a commercial AC power supply is suppressed by the regulator 2, the push-pull inverter drives the insulating high-voltage transformer 4 to obtain a high-voltage output. No stabilizing operation is performed. Therefore, the drive duty of the switching element 3 may be fixed. In a general low-voltage power supply circuit, a technology that does not require the regulator 2 by providing a stabilizing function to the push-pull inverter itself by changing the drive duty or the switching frequency of the switching element 3 with respect to the fluctuation of the output voltage. There is also.

However, the insulating high-voltage transformer 4 used in the high-voltage power supply circuit has a large step-up ratio (primary / secondary turns ratio) and requires a high withstand voltage.
Since the leakage inductance between the secondary and the secondary and the distributed capacitance between the secondary windings cannot be ignored, the control tends to be unstable. Therefore, when high stability of the output voltage is required in this type of inverter circuit used in the high-voltage power supply circuit, the regulator 2 is provided in the preceding stage of the push-pull inverter to stabilize the output voltage.

In the conventional push-pull inverter type power supply, the operating current has a rectangular wave, so that there is a problem that switching noise is large and power loss at the time of turn-on and turn-off of the switching element becomes large in high-frequency operation. However, in recent years, there has been proposed a method of reducing noise and switching loss by making a current or voltage waveform close to a sine wave using resonance. These are described, for example, in JP-A-64-43062, JP-A-1-91659, and JP-A-5-917.
40, JP-A-5-304775, and the like.

FIG. 9 shows operation waveforms of respective parts of a conventional power supply circuit of a push-pull inverter type. First, FIG.
(A) shows the waveform of each part of the rectangular wave current inverter that does not use resonance. Since both the operating voltage (VSW) and the operating current (ISW) of the switching element of the inverter are rectangular waves, the switching noise is large, and the power loss (PLOSS) when the switching element is turned on and off is increased.

FIG. 8 shows an example of the prior art of the current resonance type push-pull inverter system. In the figure, the leakage inductance 9 of the insulating high-voltage transformer 4 is used as the inductance of the resonance circuit.
As a capacitor of the resonance circuit, a current resonance capacitor 11 is added to the output of the rectifier circuit 5. The choke coil 6 is set to a constant serving as a constant current source so that the smoothing capacitor 7 does not affect the current resonance capacitor 11.

FIG. 9B shows waveforms of respective parts of the current resonance type inverter. When the switching element 3 is turned on, the diode of the rectifier circuit 5 is turned on in the forward direction, and a charging current flows. Here, since the inductance of the choke coil 6 is set to a value that is sufficiently large with respect to the secondary conversion value of the leakage inductance 9, the current (ISW) flowing through the turned on switching element 3 is equal to 2 of the leakage inductance 9. It becomes a sinusoidal resonance current by the secondary conversion value and the current resonance capacitor 11.

Therefore, in this current resonance type inverter system, although the operating voltage (VSW) remains a rectangular wave, the switching noise is small and the power loss (PLOSS) when the switching element is turned off is also small. . In the current resonance type inverter, the switching loss occurs at the moment of turn-on.
This is due to the influence of the parasitic capacitance of the switching element.

[0010]

As shown in FIG. 9A, in a rectangular wave current inverter that does not use resonance, the switching element is turned off when the current is large. Therefore, current noise is large and power loss (PLOSS) at the time of turn-on and turn-off is large.

In the current resonance type push-pull inverter system shown in FIG. 8 configured to improve this, the switching element is turned off when the current changed sinusoidally by current resonance becomes zero. .
Therefore, the current noise is small, and the power loss (PLOSS) at the time of turn-on and turn-off is also small.

However, when the current is completely resonated, the peak value of the current (ISW) flowing while the switching element 3 is on is approximately 3 to 5 times the current (ISW) flowing in the case of the rectangular wave current inverter. Since the switching element 3 and the diode for the secondary-side rectifier circuit 5 must have a current capacity of 3 to 5 times the average current value, the secondary-side smoothing circuit must be selected. In this case, the inductance of the choke coil 6 needs to be set to a value that is sufficiently large with respect to the secondary conversion value of the leakage inductance 9, so that the circuit scale becomes large.

Further, since the current peak value flowing during the period when the switching element 3 is on increases due to resonance, when the switching element 3 is a bipolar transistor, it depends on the product of the flowing current and the collector saturation voltage. In the case of a MOSFET, there is a disadvantage that the power loss (PLOSS) due to the product of the square of the flowing current and the on-resistance is extremely large in the case of a MOSFET.

The present invention has been made to solve the above-mentioned drawbacks of the prior art, and its object is to reduce the size of the circuit.
While simplifying, and reducing the peak value of the current flowing through the switching element from the conventional current resonance type, to reduce the switching noise and the switching loss,
An object of the present invention is to provide a push-pull inverter type power supply circuit in which power loss during the ON period of a switching element is also reduced.

[0015]

SUMMARY OF THE INVENTION A high-voltage power supply circuit according to the present invention is a high-voltage power supply circuit for generating a high DC voltage from DC input voltage insulation, and includes a regulator for suppressing input voltage fluctuations and output load fluctuations. A pair of switching elements that operate in a push-pull mode with a fixed on-duty; an insulating high-voltage transformer whose primary side is driven by the pair of switching elements; a secondary-side distributed capacitor of the insulating high-voltage transformer; And a resonance circuit formed by a leakage inductance of the high-voltage transformer, wherein a current flowing through the switching element is resonated. The resonance circuit produces a resonance action at a switching timing of the switching element.

The resonance operation of the resonance circuit is optimized by adjusting a leakage inductance value of the insulating high-voltage transformer. In this case, the value of the leakage inductance is adjusted according to the number of turns of the winding of the insulating high-voltage transformer.

Further, the resonance action is optimized by adjusting the value of the secondary side distribution capacitance. In this case, the value of the leakage inductance is adjusted according to the number of turns of the secondary winding of the insulating high-voltage transformer.

Further, in addition to the leakage inductance of the insulating high-voltage transformer, an external inductor added to the primary wiring of the insulating high-voltage transformer may be further included to optimize the resonance operation. In addition to the secondary distributed capacitance of the insulating high-voltage transformer, an external capacitor added between secondary windings of the insulating high-voltage transformer may be further included to optimize the resonance operation.

In short, the present power supply circuit includes a regulator for suppressing a change in input voltage and a change in load of output,
A pair of switching elements operating in a push-pull mode, an insulating high-voltage transformer driven by the pair of switching elements, and connected to the secondary side of the insulating high-voltage transformer; A rectifying and smoothing circuit for obtaining an output voltage;
The secondary side distributed capacitance and the leakage inductance of the insulated high-voltage transformer constitute a resonance circuit that resonates (partially acts) the current only at the moment when the switching element is turned on (switching timing), and resonates the current flowing through the insulated switching element. In this case, the values of the leakage inductance and the secondary-side distributed capacitance of the insulating high-voltage transformer are optimized.

In an insulating high-voltage transformer used in a high-voltage power supply circuit, a step-up ratio (primary / secondary turns ratio) is large and a leakage inductance between the primary and the secondary is required due to a structure requiring a dielectric strength voltage. Also, the distributed capacitance between the secondary windings cannot be ignored. In the present invention, when designing a transformer, the leakage inductance between the primary and secondary and the distributed capacitance between the secondary windings are optimized. By doing so, the current resonates only at the moment when the switching element is turned on, but the peak value of the current in this resonance region is made lower than in the case of the conventional current resonance type inverter, and the turn-off is performed when the switching element is turned off. The operation is performed in a state where the current is lower than that of the conventional square wave current inverter. As a result, switching noise at the moment of turn-on and off can be reduced, and switching loss can be reduced.

In addition, by the above optimization, the current flowing through the switching element during the ON period is smaller than that of the conventional current resonance type inverter, so that the power loss during the ON period can be reduced. Further, according to the present invention, since the distributed capacitance between the secondary windings of the high voltage transformer is used as the capacitor of the resonance circuit, the influence of the smoothing capacitor at the subsequent stage of the rectifier circuit on the resonance action is eliminated. A choke coil on the secondary side is not required, and the circuit configuration can be simplified and the element can be downsized, in addition to the downsizing of the element accompanying the reduction of the current value.

[0022]

Next, an embodiment of the present invention will be described with reference to the drawings. In the drawings referred to in the following description, the same parts as those in the other drawings are denoted by the same reference numerals.

FIG. 1 is a block diagram showing an embodiment of a high-voltage power supply circuit according to the present invention. 8, the same parts as those in FIG. 8 are denoted by the same reference numerals, and detailed description of those parts will be omitted.

As in the above-described conventional example, the DC input power supply 1
After the DC voltage supplied from the inverter 2 is stabilized by the regulator 2, the insulating high-voltage transformer 4 is driven by the pair of switching elements 3 of the push-pull inverter.
In the secondary circuit, the rectifier circuit 5 and the smoothing capacitor 7
After the DC high voltage output is obtained, the DC voltage is supplied to the load 8. Since the insulating high-voltage transformer 4 is required to have a large step-up ratio and a high withstand voltage, a leakage inductance 9 and a distributed capacitance 10 between secondary windings exist as distributed constants in the configuration. The leakage inductance 9 and the distributed capacitance 10 between the secondary windings constitute a resonance circuit for current resonance.

In this embodiment, the distributed capacitance between the secondary windings of the insulating high-voltage transformer 4 is used as the capacitor of the resonance circuit. For this reason, the capacitor of the resonance circuit is provided before the rectifier circuit. Therefore, in the secondary-side smoothing circuit, the smoothing capacitor 7 does not affect the above-described resonance operation, and thus does not require a choke coil having a large inductance value as in the related art.

FIG. 2 is a diagram showing an equivalent circuit of the push-pull type insulated high-voltage transformer 4. An equivalent circuit of the push-pull insulated high-voltage transformer 4 having the turns ratio N (= secondary winding number / primary winding number) shown in FIG. In contrast to the ideal transformer shown in FIG. 2A, with reference to FIG.
Leakage inductance 9 (L1), primary winding copper loss resistance 1
1 (R1) and an exciting inductance 13 (L2).

On the secondary side of the transformer, a secondary winding copper loss resistor 12 (R2) and a secondary winding distributed capacitance 10 (C
1) exists. In the circuit design, the step-up ratio of the transformer is determined to be a fixed value, but the turns ratio N (= secondary turns /
If the number of primary turns is fixed, the excitation inductance (L
2) itself can be varied. The leakage inductance 9 (L1) changes due to the coupling between the primary winding and the secondary winding. If the primary and secondary insulating structures (insulating layers) are kept constant among the conditions relating to this coupling, it is possible to adjust the number of turns of the winding. Further, the distribution capacitance 10 (C1) between the secondary windings can be adjusted by the winding method, the number of layers, and the number of turns of the secondary winding.

The operation of this embodiment will be described below with reference to the drawings. FIG. 3 is a primary-side converted equivalent circuit diagram showing the resonance action of the circuit of FIG. However, this equivalent circuit shows only a half cycle from the time when the switching element on one side of the push-pull inverter transitions from off to on and then turns off, and the diode of the rectifier circuit 5 has only one arm. It is shown.

In FIG. 3, the same parts as those in FIGS. 1 and 2 are denoted by the same reference numerals. However, the value converted to the primary side is shown in the figure for the secondary side circuit constant.

Assuming that the turns ratio of the transformer is N (= secondary turns / primary turns), the primary-side converted value of the secondary winding copper loss resistor 12 is R2 / N ² , and the secondary-side distributed capacitance 10 The primary conversion value is C1 · N ² , the primary conversion value of the secondary smoothing capacitor 7 is C2 · N ² , and the primary conversion value of the output current is Io · N.

The DC power supply 16 indicates the output of the regulator 2 in FIG. 1, and its output voltage is set to VIN. Since C1 · N ² << C2 · N ² , the primary-side converted value Io · N of the output current is set to the constant current source 14. In this equivalent circuit, the current flowing through the switching element at the time of turn-on is such that the diode of the rectifier circuit 5 is not conducting, so that the leakage inductance 9 (L1) and the primary-side converted value (C1 · N) of the secondary-side distributed capacitance 10 are obtained. ² ) Resonant action is caused by the resonant circuit formed by ² ). In the equivalent circuit, L2 is ignored from L1 << L2, and R = R1 + R2
Assuming that / N ² , after the switching element is turned on, it flows through the switching element during a period in which the resonance action by the leakage inductance 9 (L1) and the primary conversion value (C1 · N ² ) of the secondary distribution capacitor 10 continues. Time change of current I
(T) is as follows when the moment of turn-on is t = 0.

[0032] I (t) = VIN · [ {L1 / (C1 · N 2) -R 2/4} -0.5] · exp (-R · t / 2L1) · sin [{(L1 · C1 · N 2 ) ⁻¹ − (R ² / 4L1 ² )｝ ^0.5 · t] [A] Here, the peak value Ip of the current flowing through the switching element
For eak, R (=
R1 + R2 / N ² ) is several tens of mΩ to 100 mΩ, so that the following is approximately obtained.
It is determined by the resonance action of the resonance circuit constituted by the secondary side distribution capacitance 10.

Ipeak ≒ VIN ｛｛L1 / ( ^{C1ＮN 2} ) ^-0.5 [A] However, in practice, the voltage applied to the secondary-side distributed capacitance 10 is a steady value (the secondary winding output voltage component). ), The diode of the rectifier circuit 5 is turned on near the peak value of the current, and the resonance ends. After the diode of the rectifier circuit 5 is turned on, the energy accumulated in the leakage inductance 9 (L1) during the period from t = 0 until the diode of the rectifier circuit 5 is turned on is superimposed on the input from the DC power supply 16 and output. Is done.
Focusing on the current flowing through the leakage inductance 9, the secondary side distributed capacitance 10, and the diode of the rectifier circuit 5,
This is similar to the operation of a well-known step-up DC / DC converter.

FIG. 4 is a diagram showing a secondary-side equivalent circuit (for a half cycle), a voltage-current waveform diagram of each part thereof, and a step-up DC / DC converter for explaining the operation of FIG. 1 in comparison with a step-up DC / DC converter. FIG. 3 is a circuit diagram of a / DC converter and a current waveform diagram of each part thereof. 1, 2 and 3 are denoted by the same reference numerals. However, the primary-side circuit constant is shown in the figure as a value converted to the secondary side. Secondary converted value of the leakage inductance 9 becomes L1 · N ^2.

In the step-up DC / DC converter, the output of the DC power supply 16 is short-circuited via the choke coil while the switching element 17 is on, and energy is stored in the choke coil. Next, while the switching element 17 is off, the stored energy is superimposed on the input DC power supply 16 and output. In the current waveform of each part of the step-up DC / DC converter shown in FIG. 4B, the current IQ flows during the period in which the switching element 17 is on (t21) and the current IQ flows in the diode during the period when the switching element 17 is off (t22). The sum of the current ID is the current I flowing through the choke coil.
L. Here, the output current Io is an average value of the current IL flowing through the choke coil. In the case of a step-up DC / DC converter, the peak value of the current IL flowing through the choke coil is determined by the operation duty of the switching element 17.

Next, in this embodiment, when the rectangular wave pulse voltage source 15 having an amplitude corresponding to the output voltage of the secondary winding rises, the voltage applied to the distributed capacitance 10 between the secondary windings becomes a steady value. During the period (resonance region) until the voltage reaches the amplitude of the secondary winding output voltage (resonance region), the diode of the rectifier circuit 5 is off, and the secondary conversion value L1 · N of the leakage inductance 9 is obtained.
² store energy. Next, the secondary winding distributed capacitance 1
When the voltage applied to zero reaches a steady value, the diode of the rectifier circuit 5 is turned on and the leakage inductance 9
Is superimposed on the input and output.

In the current-voltage waveform of each part in FIG. 4A, a period until the voltage VC applied to the secondary winding distributed capacitor 10 reaches a steady value (the amplitude of the secondary winding output voltage). The sum of the current IC flowing through the distributed capacitance 10 between the secondary windings in the (resonance region: t11) and the current ID flowing through the diode during the period when the diode of the rectifier circuit 5 is on (linear region: t12) is 2 of the leakage inductance 9. a current flows to the next side converted value L1 · N ² IL (= the current flowing through the transformer secondary winding).

In this embodiment, the peak value Ipeak of the current IL is, as described above, the leakage inductance 9
And the resonance action of the resonance circuit constituted by the secondary-side distributed capacitance 10. The current ID flowing through the diode of the next rectifier circuit 5 during the ON period (linear region) of the diode is calculated from the peak value Ipeak by using the Ipeak
It changes linearly with a slope that depends on the value of k. The current value Ioff converges until the rectangular wave pulse voltage source 15 falls (actually, the polarity of the output of the secondary winding is inverted) and the diode for one arm of the rectifier circuit 5 is turned off.
Is the energy stored in the leakage inductance 9 during the period when the diode is off (the peak value Ipeak of IL)
And the output current Io.

The ON period of the switching element is set to ton.
Assuming (= switching cycle × on duty), the converging current value Ioff can be expressed as follows from the above expressions of I (t) and Ipeak.

Ipeak ≒ VINｐ {L1 / (C1 ・ N ² )｝ ^-0.5 [A] t11 ≒ π ≒ (L1ＣC1 ・ N ² ) ^0.5 / 2 [sec], ton = t11 + t12 [sec] In this case, Ioff ≒ 2Io + (√2 / 2) · Ipeak · {ton / (ton−t11)} − (1 + √2 / 2) · Ipeak [A].

Therefore, by changing the value of the leakage inductance 9 (L1) or the value of the secondary side distribution capacitance 10 (C1), the peak value Ipeak of the IL and the current value Ioff converging before the diode is turned off are reduced. It can be seen that the adjustment can be performed, that is, the current waveform flowing through the primary-side switching element can be adjusted.

Next, in order to explain the optimization of the resonance operation of the power supply circuit of this embodiment, a description will be given with reference to FIG. FIG. 3 shows current waveform diagrams of respective parts when the leakage inductance is fixed and the distribution capacitance between the secondary windings is changed.

5 (A), 5 (B) and 5 (C), the current IC flowing through the above-mentioned secondary winding distributed capacitance and the rectification The current ISW flowing through the primary-side switching element is shown in addition to the current ID flowing through the diode while the diode of the circuit is on. Io in the figure is a secondary output current value.

FIG. 5 (B) shows current waveforms at various parts when the distributed capacitance between the secondary windings is small. In this case, the current IC flowing through the distributed capacitor is small, and the peak value IB1 of the current ISW flowing through the primary side switching element and the current value IB2 at the time when the switching element is turned off are almost equal, and the operation close to the rectangular wave current inverter is performed. Do. Therefore, it is impossible to reduce the current noise and the switching loss at the time of turning on and turning off.

FIG. 5C shows the current waveforms of the respective parts when the distribution capacitance between the secondary windings is large. In this case, the current IC flowing through the distributed capacitance becomes large, and the peak value IC1 of the current ISW flowing through the primary-side switching element is several times as large as that of the rectangular wave current inverter. If the current value IC2 at the time when the switching element is turned off is not set to the perfect current resonance as in the related art but is set so that the current convergence value in the linear region becomes zero, the turn-off at zero current can be realized. However, the peak value I in this case is
Since C1 rises to about the same level as the current resonance type inverter in the related art, it is impossible to reduce the power loss during the ON period of the switching element.

FIG. 5A shows waveforms of respective parts when the distribution capacitance between the secondary windings is set to an appropriate value. In this case, the current ISW flowing through the primary-side switching element has a sinusoidal current waveform due to current resonance in the resonance region until the diode of the secondary-side rectifier circuit is turned on by the above-described resonance action. The peak value IA1 is about 1.5 to 1.8 times as large as that of the square wave current inverter, but is much smaller than that of the conventional complete current resonance inverter.

Next, in the linear region, the switching element is turned off when the current flowing through the diode of the secondary side rectifier circuit changes linearly from the peak value IA1 and converges to the current value IA2. This current value IA2 is the convergence current value Ioff of FIG. 4A, and is about 1/2 of that of the rectangular wave current inverter by optimizing the peak value IA1 by adjusting the distribution capacitance. The switching element can be turned off with a small value. As a result, reduction of the current noise at the time of turning on and off and reduction of the switching loss can be realized. Further, the current ISW flowing through the switching element including the current peak value IA1 is smaller than that of the conventional complete current resonance type inverter, so that the power loss during the ON period of the switching element can be reduced.

Therefore, as shown in the PLOSS waveform indicating the product of ISW and VSW in FIG. 5, the total loss (PLOSS), which is the sum of the switching loss and the on-period loss of the switching element, is reduced by the above-described optimization. It can be reduced compared to current inverters and complete current resonant inverters. In the above description, an example was shown in which the distribution capacitance between the secondary windings of the insulating high-voltage transformer was changed, but the same effect can be naturally obtained by adjusting the leakage inductance of the transformer.

In this embodiment, although the circuit scale is slightly increased, as shown in FIG. 6, in addition to the leakage inductance of the insulating high-voltage transformer, an external inductor is connected to the primary wiring of the insulating high-voltage transformer. By adding 90 in series, the resonance action can be adjusted and optimized.

As shown in FIG. 7, resonance is achieved by adding an external capacitor in parallel between the secondary windings of the insulating high-voltage transformer in addition to the secondary-side distributed capacitance of the insulating high-voltage transformer. The effect can be adjusted and optimized.

In a general low-voltage power supply circuit, an external inductor may be added to the primary wiring of the transformer,
It is possible to adjust the resonance action by adding an external capacitor 100 between the secondary windings of the transformer.

[0052]

As described above, according to the present invention, the current resonates only at the moment when the switching element is turned on by the leakage inductance of the high-voltage transformer and the distributed capacitance between the secondary windings, and the current peaks in this resonance region. By adjusting the leakage inductance or the distributed capacitance between the secondary windings at the time of designing the high-voltage transformer so that the value becomes lower than that of the conventional current resonance type inverter, the current of the switching element can be reduced. There is an effect that it can be turned off in a lower state. As a result, switching noise at the moment of turn-on and off can be reduced, and switching loss can be reduced. Moreover, since the current flowing through the switching element during the ON period is smaller than that of the conventional current resonance type inverter,
There is an effect that the power loss during the ON period can be reduced.

Further, since the distributed capacitance between the secondary windings of the high voltage transformer is used as the capacitor of the resonance circuit, the effect of the smoothing capacitor at the subsequent stage of the rectifier circuit on the resonance operation is eliminated. Since a choke coil is not required, there is an effect that the circuit configuration can be simplified and the element can be downsized, in addition to the downsizing of the element accompanying the reduction of the current value.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a configuration of a high-voltage power supply circuit according to an embodiment of the present invention.

FIG. 2 is a diagram showing an equivalent circuit of a push-pull type insulated high-voltage transformer.

FIG. 3 is a diagram illustrating a resonance operation of the high-voltage power supply circuit of FIG. 1;
It is a secondary side equivalent circuit diagram.

4A is a secondary-side equivalent circuit (for a half cycle) for explaining the operation of the high-voltage power supply circuit of FIG. 1 and a voltage-current waveform diagram of each part thereof, and FIG. 4B is a step-up DC / DC converter FIG. 3 is a circuit diagram of FIG.

5A and 5B are waveform diagrams of respective parts when the distribution capacitance between secondary windings is changed for explaining the optimization of the resonance operation of the high-voltage power supply circuit of FIG. 1; FIG. (B) shows each part current waveform when the distributed capacitance is small, and (C) shows each part current waveform when the distributed capacitance is large or when an external capacitor is added.

FIG. 6 is a block diagram showing a configuration of a high-voltage power supply circuit according to another embodiment of the present invention.

FIG. 7 is a block diagram showing a configuration of a high-voltage power supply circuit according to another embodiment of the present invention.

FIG. 8 is a block diagram showing a configuration of a conventional high-voltage power supply circuit.

FIG. 9A is a waveform diagram of each part of a switching element of a rectangular wave current inverter in a conventional high-voltage power supply circuit, and FIG.
FIG. 4 is a waveform diagram of each part of a switching element of a current resonance inverter in a conventional high-voltage power supply circuit.

[Explanation of symbols]

 Reference Signs List 1 DC input power supply 2 Regulator 3 Push-pull inverter switching element, 4 Insulated high-voltage transformer 5 Rectifier circuit 6 Choke coil 7 Smoothing capacitor 8 Load 9 Transformer leakage inductance 10 Transformer secondary winding distributed capacitance 11 Current resonance capacitor 12 Transformer Secondary winding copper loss resistance 13 Excitation inductance of transformer 14 Constant current source 15 Square wave pulse voltage source 16 DC power supply 17 Switching element

Claims (8)

[Claims] 1. A high-voltage power supply circuit for generating a DC high voltage from a DC input voltage insulation, comprising: a regulator for suppressing input voltage fluctuation and output load fluctuation; and a push-pull mode having a fixed on-duty. A resonance circuit including a pair of operating switching elements, an insulating high-voltage transformer whose primary side is driven by the pair of switching elements, a secondary-side distributed capacitance of the insulating high-voltage transformer, and a leakage inductance of the insulating high-voltage transformer; Wherein the current flowing through the switching element is resonated. 2. The high-voltage power supply circuit according to claim 1, wherein the resonance circuit generates a resonance action at a switching timing of the switching element. 3. The high-voltage power supply circuit according to claim 1, wherein the resonance circuit optimizes a resonance operation by adjusting a leakage inductance value of the insulating high-voltage transformer. 4. The high-voltage power supply circuit according to claim 3, wherein said leakage inductance value is adjusted by the number of windings of said insulating high-voltage transformer. 5. The high-voltage power supply circuit according to claim 1, wherein the resonance operation of the resonance circuit is optimized by adjusting a value of the secondary-side distributed capacitance. 6. The high-voltage power supply circuit according to claim 5, wherein said leakage inductance value is adjusted by the number of turns of a secondary winding of said insulating high-voltage transformer. 7. The resonance circuit further includes, in addition to a leakage inductance of the insulating high-voltage transformer, an external inductor added to a primary wiring of the insulating high-voltage transformer, so as to optimize the resonance operation. The high-voltage power supply circuit according to any one of claims 1 to 6, wherein: 8. The resonance high-voltage transformer includes a secondary-side distributed capacitance of the high-voltage isolation transformer and a secondary-side distribution capacitance of the high-voltage isolation transformer.
It further includes an external capacitor added between the secondary windings,
7. The high-voltage power supply circuit according to claim 1, wherein the resonance action is optimized.