JP2001178130A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve power conversion efficiency and to simplify a circuit by eliminating a current limiter resistor of a large capacity and a low resistance at an AC input line or an efficiency improvement, by forming a circuit for short-circuiting the resistor of an electromagnetic relay or a thyristor triac or the like. SOLUTION: A rush current, when the power supply is turned on, is limited with a soft switching circuit 13, using a transistor together with a power factor improving circuit 10 for improving the power factor by feeding back the switching output to the rectified current path by electrostatically or magnetically coupling and interrupting the rectified current.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices, and more particularly to a switching power supply circuit having a configuration for improving a power factor.

[0002]

2. Description of the Related Art The applicant has previously proposed various power supply circuits having a resonance type converter on the primary side. In addition, various types of power supply circuits including a power factor improvement circuit for improving a power factor of a current resonance type converter have been proposed.

FIG. 5 is a circuit diagram showing an example of a switching power supply circuit constructed based on the invention previously filed by the present applicant. This power supply circuit has a configuration in which a power factor improvement circuit for improving a power factor is provided for a self-excited current resonance type switching converter.

The power supply circuit shown in FIG. 1 includes a bridge rectifier circuit Di for full-wave rectification of a commercial AC power supply AC. In this case, the rectified output rectified by the bridge rectifier circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 10, and both ends of the smoothing capacitor Ci correspond to the level of one time of the AC input voltage VAC. The rectified smoothed voltage Ei is obtained. Also, in the rectifying / smoothing circuit (Di, Ci), an inrush current limiting resistor Ri is inserted in the rectification current path so as to suppress an inrush current flowing into the smoothing capacitor when the power is turned on, for example. ing. The switch PS is a power-on switch.

In the power factor correction circuit 10 shown in FIG. 1, a filter choke coil LN, a high-speed recovery type diode D1, and a choke coil LS are provided between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. They are connected in series and inserted. The filter capacitor CN is inserted between the anode side of the high-speed recovery diode D1 and the positive terminal of the smoothing capacitor Ci to form a normal mode low-pass filter together with the filter choke coil LN.

In the power factor correction circuit 10, an end of a primary side series resonance circuit, which will be described later, is connected to a connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS. The switching output obtained in the series resonance circuit is fed back. The power factor improving operation by the power factor improving circuit 10 will be described later.

This power supply circuit includes a self-excited current resonance type converter using a rectified smoothed voltage Ei, which is a voltage across a smoothing capacitor Ci, as an operating power supply. In this current resonance type converter, as shown in the figure, switching elements Q1 and Q2 of two bipolar transistors are half-bridge-coupled and then inserted between the connection point on the positive side of the smoothing capacitor Ci and ground. It is connected. Starting resistors RS1 and RS2 are inserted between the collectors and the bases of the switching elements Q1 and Q2, respectively. The resistors RB1 and RB2 connected to the bases of the switching elements Q1 and Q2 set the base current (drive current) of the switching elements Q1 and Q2. Further, clamp diodes DD1 and Q2 are connected between the base and the emitter of the switching elements Q1 and Q2, respectively.
DD2 is inserted. The clamp diodes DD1 and DD2 are
A current path for a clamp current flowing through the base-emitter is formed during a period in which the switching elements Q1 and Q2 are turned off. And the resonance capacitors CB1, C
B2 is a drive winding N of a drive transformer PRT described below.
A series resonance circuit (self-excited oscillation drive circuit) for self-excited oscillation is formed together with B1 and NB2, and the switching elements Q1, Q2
Determine the switching frequency of 2.

[0008] Drive transformer PRT (Power Regulati
ng Transformer) is provided to drive the switching elements Q1 and Q2 and to perform constant voltage control by variably controlling the switching frequency. In this case, the drive windings NB1 and NB2 and the resonance current detection are performed. Winding N
D is wound, and a control winding N is provided for each of these windings.
C is an orthogonal saturable reactor wound in the orthogonal direction. One end of the drive winding NB1 of the drive transformer PRT is connected to the base of the switching element Q1 via a series connection of the resistor RB1 and the resonance capacitor CB1, and the other end is connected to the emitter of the switching element Q1. One end of the drive winding NB2 is grounded, and the other end is connected to the base of the switching element Q2 via a series connection of a resistor RB2 and a resonance capacitor CB2. The drive winding NB1 and the drive winding NB2 are wound so that voltages of opposite polarities are generated.

Insulated converter transformer PIT (Power Is
olation Transformer) is the switching element Q1, Q2
Is transmitted to the secondary side. One end of the primary winding N1 of the insulation converter transformer PIT is connected to the contact point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, thereby providing a switching output. Is obtained.

The other end of the primary winding N1 is connected to the connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS in the power factor correction circuit 10 via the series resonance capacitor C1. ing.

In this case, the series resonance capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonance capacitor C1 and the primary winding N1 are connected.
Converter transformer PIT including (series resonance winding)
The primary side series resonance circuit for making the operation of the switching converter a current resonance type is formed by the leakage inductance (leakage inductance) component.

On the secondary side of the insulated converter transformer PIT shown in FIG. 1, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, D
By connecting O3, DO4 and smoothing capacitors CO1, CO2 as shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT as a control current, as will be described later. The constant voltage control is performed as described above.

In the switching operation of the power supply circuit having the above configuration, when a commercial AC power supply is first turned on by the switch PS, a starting current is supplied to the bases of the switching elements Q1 and Q2 via the starting resistors RS1 and RS2, for example. That is, for example, if the switching element Q1 is turned on first, the switching element Q2 is controlled to be turned off. As an output of the switching element Q1, a resonance current flows through the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1, and when the resonance current becomes zero, the switching element Q2 is turned on and the switching element Q1 is turned on. It is controlled to be off. Then, a resonance current flows in the opposite direction through the switching element Q2. Thereafter, a self-excited switching operation in which the switching elements Q1 and Q2 are turned on alternately is started. As described above, the switching elements Q1 and Q2 are alternately opened and closed by using the terminal voltage of the smoothing capacitor Ci as the operating power supply.
A drive current close to the resonance current waveform is supplied to the primary winding N1, and an alternating output is obtained to the secondary winding N2.

The constant voltage control by the drive transformer PRT is performed as follows. For example, assuming that the secondary output voltage EO1 fluctuates so as to increase with the AC input voltage level or load fluctuation, the level of the control current flowing through the control winding NC as described above is also changed to the secondary output voltage EO1.
Is controlled so as to increase in accordance with the rise of. Under the influence of the magnetic flux generated in the drive transformer PRT by this control current, the drive transformer PRT tends to approach a saturation state, and acts to reduce the inductance of the drive windings NB1 and NB2. The switching frequency is controlled so as to increase by changing the condition of the oscillation circuit. In this power supply circuit, the switching frequency is set in a frequency region higher than the resonance frequency of the series resonance circuit of the series resonance capacitor C1 and the primary winding N1 (upper side control). Then, the switching frequency is separated from the resonance frequency of the series resonance circuit.
Thereby, the resonance impedance of the series resonance circuit with respect to the switching output increases.

By increasing the resonance impedance in this way, the drive current supplied to the primary winding N1 of the primary series resonance circuit is suppressed, and as a result, the secondary output voltage is suppressed. , Constant voltage control is achieved. Hereinafter, the constant voltage control method using the above method will be referred to as “switching frequency control method”.

The power factor improving operation by the power factor improving circuit 10 is as follows. In the configuration of the power factor correction circuit 10 shown in this figure, the switching output supplied to the series resonance circuit (N1, C1) is rectified by a so-called magnetic coupling type through an inductive reactance that is assumed to be included in the choke coil LS itself. Return to the path.

The switching output fed back as described above causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. The superimposed voltage of the switching cycle causes the high speed recovery type diode D1 to be superimposed. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN and choke coil LS also increases due to the intermittent action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

FIG. 6 is a circuit diagram showing another example of the configuration of a switching power supply circuit that can be configured based on the invention previously proposed by the present applicant. This power supply circuit is also 2
A current resonance type converter in which the switching elements are half-bridge-coupled is provided, and its driving method is separately excited. Also in this case, a power factor improving circuit for improving the power factor is provided. The same parts as those in FIG. 5 are denoted by the same reference numerals, and description thereof will be omitted.

The primary-side current resonance type converter shown in FIG. 1 includes two switching elements Q11 and Q12, for example, MOS-FETs. here,
By connecting the drain of the switching element Q11 to the line of the rectified smoothing voltage Ei, connecting the source of the switching element Q11 and the drain of the switching element Q12, and connecting the source of the switching element Q12 to the primary side ground, it is compatible with separately excited type. A half-bridge connection is obtained. These switching elements Q11 and Q12 are switching-driven by the oscillation drive circuit 2 so that on / off operations are alternately repeated, and intermittently output the rectified smoothed voltage Ei as a switching output. In this case, a clamp diode DD1, which is connected between the drain and source of each of the switching elements Q11 and Q12 in the direction shown in FIG.
DD2 is provided.

In this case, the switching element Q
11. By connecting one end of the primary winding N1 of the isolated converter transformer PIT to the source-drain connection point (switching output point) of Q12, the switching output is supplied to the primary winding N1. Is done. The other end of the primary winding N1 is connected via a series resonance capacitor C1 to a filter choke coil LN of a power factor correction circuit 11 described below.
And the anode of the high speed recovery type diode D1.

Also in this case, a series resonance circuit for making the switching power supply circuit a current resonance type is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance component of the insulating converter transformer PIT including the primary winding N1. .

In this case, the control circuit 1 outputs a control signal having a level corresponding to the fluctuation of the DC output voltage EO1 to the oscillation drive circuit 2, for example. The oscillation drive circuit 2 changes the frequency of the switching drive signal supplied from the oscillation drive circuit 2 to each gate of the switching elements Q11 and Q12 based on the control signal supplied from the control circuit 1,
The switching frequency is made variable. Also, in the power supply circuit shown in this figure, similarly to the power supply circuit shown in FIG. 5, the switching frequency is set as a region higher than the series resonance frequency. When the DC output voltage EO1 rises, for example, the control circuit 1 controls the oscillation drive circuit 2 to increase the switching frequency according to the level. This allows
Constant voltage control is performed in the same manner as described with reference to FIG. The starting circuit 3 is provided for detecting the voltage or current obtained on the rectifying / smoothing line immediately after the power is turned on, and for starting the oscillation drive circuit 2. Is input as an operation power supply.

In the power factor correction circuit 11 shown in this figure, the positive output terminal of the bridge rectifier circuit Di and the smoothing capacitor Ci
Of the filter choke coil LN−
A fast recovery diode D1 is inserted in series. Here, the filter capacitor CN is provided in parallel with the series connection circuit of the filter choke coil LN and the high-speed recovery type diode D1. Also in this connection form, the filter capacitor CN forms a normal mode low-pass filter together with the filter choke coil LN. The resonance capacitor C3
Are provided in parallel with the high-speed recovery type diode D1. Although a detailed description is omitted here, for example, the resonance capacitor C3 forms a parallel resonance circuit together with, for example, the filter choke coil LN and the like, so that its resonance frequency is substantially equal to the resonance frequency of the series resonance circuit described later. Is set. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. As described above, for the power factor improving circuit 11, the end of the series resonance circuit (N1, C1) is connected to the connection point between the filter choke coil LN and the anode of the high-speed recovery type diode D1. Is connected.

In such a connection form, the switching output obtained on the primary winding N1 is connected to the series resonance capacitor C1.
The switching output is fed back to the rectified current path through the capacitive coupling of the switch. In this case, the switching output is applied to the connection point between the filter choke coil LN and the anode of the fast recovery diode D1 so that the resonance current obtained in the primary winding N1 flows. become.

As the switching output is fed back as described above, the alternating voltage of the switching cycle is superimposed on the rectified current path. In the diode D1, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN also increases due to the interrupting action. Further, a voltage is generated at both ends of the resonance capacitor C3 when a current of a switching period flows through the resonance capacitor C3. However, the level of the rectified smoothed voltage Ei is reduced by the voltage across the resonance capacitor C3. Accordingly, even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, the smoothing capacitor C
The charging current to i flows. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, the conduction angle of the AC input current is expanded, and the power factor is also improved.

As described above, in the power supply circuits shown in FIGS. 5 and 6, the power factor can be improved by providing the power factor improvement circuits (10, 11). Since the power factor improving circuit is formed with a small number of components, it has the advantage that the power factor can be improved with high efficiency, low noise, small size, light weight, and low cost.

[0028]

FIG. 7 shows the configuration of the switch P in the power supply circuit shown in FIGS.
The graph shows the time change of the AC input current IAC when S is turned on at the peak of the AC input voltage VAC. When the switch PS is turned on at the peak of the AC input voltage VAC, an excessive rush charge current of 100 A or more is supplied to the smoothing capacitor Ci through the bridge rectifier circuit Di, the filter choke coil LN, the fast recovery diode D1, and the choke coil LS. And the current exceeds the maximum allowable current of the bridge rectifier circuit Di, the high-speed recovery type diode D1, and the smoothing capacitor Ci. Therefore, as shown in FIGS. 5 and 6, a large-capacity inrush current limiting resistor Ri, which is a large-capacity and low resistance, is inserted into the AC line so that the inrush charging current becomes 50 A or less as shown in FIG. Has restricted.

However, when the load power is a heavy load of 200 W or more, the power loss increases and the efficiency decreases. AC
In the 100V system, the inrush current limiting resistor Ri is set to 1Ω / 30
As shown in FIG. 7, due to the large cement resistance of W, the AC input current IAC = 50 A at time t = 0, and the AC input current IAC = 5 A at time t = 60 (msec).
However, the power conversion efficiency is reduced due to the power loss.

As a countermeasure, there is also a method in which the rush current limiting resistor Ri is short-circuited by an electromagnetic power relay or a semiconductor switch such as a thyristor or a triac one second after the switch PS is turned on, thereby reducing power loss. . However, in this case, by providing an electromagnetic power relay or a semiconductor switch and its peripheral circuit elements,
There are difficulties in terms of the number of components and cost.

FIG. 8 shows the relationship between the AC input voltage VAC and the power factor PF. Here, the maximum load power Pomax = 120 W and the minimum load power Pomin = 40 W
The characteristics under each condition of time are shown. As shown in this figure, it can be seen that the power factor PF decreases proportionally as the AC input voltage VAC increases. As the power factor PF under the condition of the minimum load power Pomin = 40 W, the maximum load power Pomax = 120
The power factor is lower than W. That is, the power factor PF decreases due to fluctuations in the AC input voltage VAC and the load power Po. This is not a problem in home electric appliances in which the AC input voltage VAC and load conditions are specified, such as a color television receiver, but in office equipment and information equipment in which the AC input voltage VAC and load conditions may fluctuate. Means that the power supply circuit cannot be used.

The characteristics shown in FIG. 8 are shown as operating waveforms as shown in FIG. Here, FIG.
9B shows the AC input voltage VAC and the AC input current IAC when the AC input voltage VAC = 100 V and the maximum load power Pomax = 120 W. FIGS. 9C and 9D show the AC input voltage VAC = The AC input voltage VAC and the AC input current IAC at 100 V and the minimum load power Pomin = 40 W are shown.
Assuming that the half cycle of the AC input voltage VAC is 10 ms, when the maximum load power Pomax = 120 W, the AC input current IA
The conduction period τ of C is actually about 5 ms, and the power factor is PF = 0.85. On the other hand, when the minimum load power Pomin is 40 W, the conduction period τ of the AC input current IAC is reduced to about 2.5 ms, and the power factor is reduced to about PF = 0.65. The value of the power factor PF obtained when the minimum load power Pomin = 40 W may not satisfy the value as the power factor required for practical use.

[0033]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a rectifying diode for inputting and rectifying commercial AC power and a rectifying / smoothing means for generating a rectified / smoothed voltage with a smoothing capacitor, an insulating converter transformer provided for transmitting a primary output to a secondary side, and Switching means for intermittently outputting a voltage by a switching element to output to a primary winding of the insulating converter transformer, and at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a series resonance capacitor. A primary-side series resonance circuit that is formed to make the operation of the switching means a current resonance type, and a switching output obtained in the primary winding is fed back to a rectified current path by magnetic coupling or electrostatic coupling, and this feedback is provided. Power by intermittent commutation current based on switching output A power factor improving means adapted improve, disposed in front of the power factor correction unit,
At the time of turning on the power, a soft switch means formed by using a transistor and an alternating voltage obtained in a secondary winding of the insulating converter transformer are input so that an inrush current to the smoothing capacitor can be limited. DC output voltage generating means configured to generate a secondary DC output voltage by performing a rectifying operation, and a constant voltage control for the secondary DC output voltage in accordance with the level of the secondary DC output voltage. And a constant voltage control means adapted to perform the control.

The soft switch means includes an NPN transistor connected to the positive electrode of the rectifier diode, a resistor arranged between the collector and base of the NPN transistor, and a resistor connected between the base of the NPN transistor and ground. And a capacitor arranged. Alternatively, the soft switch means includes a PNP transistor connected to a negative electrode side of the rectifier diode,
It is formed to have a resistor disposed between the collector and the base of the PNP transistor and a capacitor disposed between the base and the ground of the PNP transistor.

According to the above configuration, the inrush current at the time of turning on the power can be limited by the soft switch means using the transistor, whereby a large-capacity low-resistance current limiting resistor can be provided on the AC input line. It is not necessary to improve the efficiency by forming a circuit for short-circuiting the current limiting resistor by using an electromagnetic relay, a thyristor, or a triac.

[0036]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention. The switching power supply circuit shown in this figure is configured in the same manner as the switching power supply circuit shown in FIG. 5, and the same parts are denoted by the same reference numerals. That is, a full-wave rectification circuit including a bridge rectification circuit Di and a smoothing capacitor Ci is provided as a rectification / smoothing circuit for inputting a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage. A rectified smoothed voltage Ei corresponding to the level of one time is generated. A configuration including a parallel resonance circuit formed by a leakage inductance component including the primary winding N1 of the insulating converter transformer PIT and the capacitance of the parallel resonance capacitor C1 to make the operation of the switching elements Q1 and Q2 a current resonance type. Is done. As a power factor improving means, a power feedback type magnetic coupling type power factor improving circuit 10 is provided. However, in this rectifying / smoothing circuit, the inrush current limiting resistor (Ri) as shown in FIGS. 5 and 6 is not inserted into the AC input line. In this embodiment, the positive side of the bridge rectifier circuit Di and the power factor improving circuit 10
A soft switch circuit 13 is provided therebetween.

The soft switch circuit 13 includes an NPN transistor Q3, a resistor R1, and a small-capacity electrolytic capacitor C.
10. It is composed of a diode D3. The collector of the transistor Q3 is connected to the positive electrode of the bridge rectifier circuit Di, and the emitter of the transistor Q3 is connected to the power factor correction circuit 10. The resistor R1 is arranged between the collector and the base of the transistor Q3, and the capacitor C10 is arranged between the base and the primary side ground of the transistor Q3. A diode D3 is connected between the emitter and the base of the transistor Q3 to prevent a reverse voltage of the base-emitter voltage VBE.

FIG. 2 shows operation waveforms of various parts in the switching power supply circuit shown in FIG. As shown in FIG. 2A, assuming that an AC input voltage VAC is input at a commercial power supply cycle of, for example, 50H, the rectified current is supplied to the bridge rectifier circuit Di as a rectified current as shown in FIG. When the input current IAC flows, the high-speed recovery diode D1 performs a switching operation as shown by the switching current ID in FIG. In the steady state, transistor Q3 is on,
The positive and negative periods of the AC input voltage VAC shown in FIG. 2A correspond to the collector current Ic as shown in FIG.
Flows continuously, and the collector-emitter voltage VCE is in a saturation voltage state. The resistance value of the resistor R1 is Ic <
hFE · (base current IB). In a steady state where the transistor Q3 is turned on, the collector-emitter voltage VCE and the collector current Ic of the transistor Q3 are as shown in FIGS. Note that FIG.
(F) shows the feedback voltage V1 supplied to the anode side of the fast recovery diode D1 in the switching cycle.

FIG. 3 shows that the switch PS has the AC input voltage V
The state of the change of the AC input current IAC when it is turned on at the peak of AC is shown. The resistance value of the resistor R1 is such that the ASO region of the transistor Q3 has a collector current Ic <16A.
, The collector current Ic is determined to be 15A. The capacitor C10 is composed of the resistor R1 and the capacitor C
The time constant of 10 has an effect of delaying the rise of the AC input current IAC after the switch PS is turned on. Therefore, the soft start function by the soft start circuit 13 works, and the switch PS is connected to the AC input voltage VAC as shown in FIG.
, The AC input current IAC does not flow to the smoothing capacitor Ci as an excessive rush charging current. Therefore, even if a current limiting resistor is not provided in the AC input line as in the present embodiment, the AC input current IAC does not flow. The input current IAC does not exceed the maximum allowable current of the bridge rectifier circuit Di, the fast recovery type diode D1, and the smoothing capacitor Ci.

That is, since the large-capacity low-resistance inrush current limiting resistor Ri inserted into the AC line as shown in FIGS. 5 and 6 is not required, the inrush current limiting resistor Ri is not required.
And the power conversion efficiency is improved. Also, of course, the switch PS
When one second elapses after the switch is turned on, an inrush current limiting resistor Ri is short-circuited by a semiconductor switch such as an electromagnetic power relay or a thyristor or a triac, and a circuit unit for reducing power loss is not required.

As described above, the power supply circuit according to the present embodiment has a characteristic that the power factor is substantially constant with respect to the fluctuation of the AC input voltage, and the power factor increases with a decrease in the load power. Can be For this reason, the power supply circuit according to the present embodiment is not limited to a television receiver in which an AC input voltage or a load condition is specified, but is mounted on, for example, office equipment or a personal computer in which the load condition varies. Is practically sufficient.

The secondary side DC output voltage level EO1 of 50
The Hz ripple voltage component is also increased only about twice as much as that when the power factor correction circuit 10 is not provided, and is in a range where there is no practical problem as a power supply circuit used for a color television or the like. The operation of the power factor correction circuit 10 is low noise because a smooth sinusoidal waveform is obtained. Further, in practice, the choke coil LS has a small inductance value, so that a small and light element can be selected, so that the circuit can be reduced in size and weight and cost can be reduced.

In the above embodiment, an orthogonal control transformer is used as a control transformer for performing constant voltage control under a configuration having a self-excited resonance converter for the primary side. Instead of the orthogonal control transformer, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, for example, as in the case of the orthogonal control transformer, a three-dimensional structure is obtained by combining two sets of double U-shaped cores having four magnetic legs. Form a mold core. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, one of the control winding NC and the driving winding NB is wound around two magnetic legs in a positional relationship adjacent to each other among the four magnetic legs, The other winding is wound around two magnetic legs which are considered to be in a diagonal positional relationship. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. Thereby, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced.

Next, a second embodiment of the present invention will be described with reference to FIG. The power supply circuit shown in FIG. 4 is also provided with a current resonance type converter in which two switching elements are half-bridge-coupled, and the drive system is separately excited. Also in this case, as a power factor improving means,
This is an example in which a power feedback type electrostatic coupling type power factor improving circuit 11 is provided, and a soft switch circuit 13A is further provided. Note that the same parts as those in FIG.

The soft switch circuit 13A shown in FIG.
Is composed of a PNP transistor Q4, a resistor R1, a small-capacity electrolytic capacitor C10, and a diode D3. The collector of the transistor Q4 is connected to the negative side of the bridge rectifier circuit Di. The resistor R1 is arranged between the collector and the base of the transistor Q4, and the capacitor C10
Is arranged between the base and the primary side ground of the transistor Q4. Also, between the emitter and the base of the transistor Q4,
A diode D3 is connected to prevent a reverse voltage of the base-emitter voltage VBE.

That is, the soft switch circuit 13A
Has a configuration using a PNP transistor, and is inserted on the negative electrode side of the bridge rectifier circuit Di. The operation is the same as that of the soft switch circuit 13 of FIG. 1, and the rise of the AC input current IAC after the switch PS is turned on is delayed by the soft switch circuit 13A, so that the switch PS is turned on at the peak of the AC input voltage VAC. Even if it is, the AC input current IAC does not flow to the smoothing capacitor Ci as an excessive inrush charging current.
Therefore, it is not necessary to provide a current limiting resistor in the AC input line.

In the soft switch circuit 13A, the transistor Q4 may have a low-speed switching characteristic. In practice, a low-voltage, high-hFE, inexpensive general-purpose transistor can be employed.

In each of the above embodiments, a so-called half-bridge coupling system in which two switching elements are alternately switched is employed as the primary side current resonance type converter. The present invention can also be applied to a full-bridge coupling system in which two switching elements are alternately switched.

Further, in each of the above embodiments, the switching frequency control system for controlling the switching frequency has been described as an example of the constant voltage control means. It is also possible to adopt a so-called series resonance frequency control method for performing control. In the series resonance frequency control method,
When the DC output voltage output to the secondary side of the insulating converter transformer rises, the control circuit 1 controls so as to reduce the level of the current flowing through the control winding according to the level. This control lowers the series resonance frequency of the series resonance circuit formed by the leakage inductance of the insulating converter transformer including the primary winding N1 and the series resonance capacitor C1. As a result, the difference between the switching frequency and the series resonance frequency of the power supply circuit increases, the resonance impedance increases, and the resonance current flowing as the drive current through the primary winding N1 decreases. As a result, the transmission output to the secondary side of the insulated converter transformer decreases, and constant voltage control of the DC output voltage EO1 and the DC output voltage EO2 is performed.

[0050]

As described above, according to the present invention, in a switching power supply circuit provided with a power factor improving circuit, the power factor becomes substantially constant with respect to the fluctuation of the AC input voltage, and the power factor is reduced as the load power decreases. As a result, a power supply circuit that can be practically used sufficiently can be provided for office equipment and personal computers in which the load conditions fluctuate. The soft start means eliminates the need to insert an inrush current limiting resistor in the AC input line, thereby eliminating the need for a large high-temperature heat-generating component, and improving power conversion efficiency by eliminating power loss due to the inrush current limiting resistor. Note that the soft start means using an NPN-type or PNP-type transistor having a low withstand voltage and a low saturation voltage has a low loss and does not require a heat sink. Therefore, when an electromagnetic power relay is provided to short-circuit the inrush current limiting resistor, the driving power 0.5
The collector loss is substantially equal to W, and the power loss due to the soft start means of the present invention is of a level that does not pose a practical problem. Further, since there is no need to provide an electromagnetic power relay associated with the inrush current limiting resistor, it is not necessary to take measures against secondary defects such as when the electromagnetic power relay is open and fails. Also, compared with the provision of a rush current limiting resistor and the associated resistance short circuit system using electromagnetic power relays and semiconductor switches, the circuit configuration is very simple, and it is significantly advantageous in terms of the number of components and cost. Becomes

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of the switching power supply circuit shown in FIG.

FIG. 3 is an explanatory diagram of an input AC current when power is turned on in the switching power supply circuit of the present embodiment.

FIG. 4 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a second embodiment.

FIG. 5 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 6 is a circuit diagram showing a configuration of another power supply circuit as a prior art.

FIG. 7 is an explanatory diagram of an input AC current at the time of power-on of a power supply circuit of the related art.

FIG. 8 is a characteristic diagram illustrating a relationship between an AC input voltage and a power factor of a power supply circuit according to the related art.

FIG. 9 is a waveform diagram showing an operation of a prior art power supply circuit with respect to an input of a commercial AC power supply according to load power.

[Explanation of symbols]

Reference Signs List 1 control circuit, 2 oscillation / drive circuit, 3 start-up circuit, 10, 11 power factor improvement circuit, 13, 13A soft switch circuit, Di bridge rectifier circuit, Ci smoothing capacitor, D1 high speed recovery type diode, C1 series resonance capacitor, PRT Orthogonal control transformer, PIT
Insulation converter transformer, Q1, Q2, Q11, Q12 switching element

──────────────────────────────────────────────────続 き Continued on the front page (51) Int.Cl. ⁷ Identification symbol FI Theme coat ゛ (Reference) H02M 7/12 H02M 7/12 PF term (Reference) 5G065 AA01 AA06 BA04 DA06 DA07 EA06 FA02 GA06 GA07 HA04 HA12 HA13 JA01 LA01 MA01 MA03 MA10 NA01 NA02 NA04 NA06 NA09 5H006 AA02 AA05 CA01 CA07 CA12 CA13 CB01 CB05 CC02 DA02 DA04 DB01 DC05 FA02 GA02 5H730 AA14 AA18 AA20 AS01 BB26 BB62 CC04 DD02 DD23 DD35 EE03 XX07 XX07 XX

Claims (3)

[Claims] 1. A rectifying diode for inputting and rectifying commercial AC power and a rectifying / smoothing means for generating a rectified / smoothed voltage by a smoothing capacitor; an insulating converter transformer provided for transmitting a primary-side output to a secondary-side; Switching means for intermittently outputting the rectified smoothed voltage by a switching element and outputting the rectified smoothed voltage to a primary winding of the insulating converter transformer; and at least a leakage inductance component including a primary winding of the insulating converter transformer and a series resonance capacitor. A primary side series resonance circuit formed by a capacitance and making the operation of the switching means a current resonance type, and a switching output obtained in the primary winding is fed back to a rectified current path by magnetic coupling or electrostatic coupling. Intermittent rectification current based on the feedback switching output. A power factor improving means arranged to improve the power factor, and a transistor arranged before the power factor improving means, so that a rush current to the smoothing capacitor can be limited at power-on. And a DC output voltage generated by inputting an alternating voltage obtained to a secondary winding of the insulating converter transformer and performing a rectification operation to generate a secondary side DC output voltage. Switching power supply, comprising: generating means; and constant voltage control means configured to perform constant voltage control on the secondary DC output voltage in accordance with the level of the secondary DC output voltage. circuit. 2. The soft switch means includes: an NPN transistor connected to a positive electrode of the rectifier diode; a resistor arranged between a collector and a base of the NPN transistor; The switching power supply circuit according to claim 1, wherein the switching power supply circuit is formed having: 3. The soft switch means includes: a PNP transistor connected to a negative electrode of the rectifier diode; a resistor arranged between a collector and a base of the PNP transistor; The switching power supply circuit according to claim 1, wherein the switching power supply circuit is formed having: