FR2921210A1 - Forward chopping converter for energy distribution chain, has control unit controlling open and closed states of switches at chopping frequency, where voltages at terminals place diodes in conducting and blocked states, respectively - Google Patents

Abstract

The converter has a main switch (I2) connected to an input terminal (A) and a free end of primary windings (E1, E2) of transformers (T1, T2), and a secondary switch (I1) connected to the main switch. An output circuit has secondary windings (S1, S2) respectively forming current rectifier circuits with rectifier diodes (D1, D2) to provide continuous current in a load (R). Voltages appearing in one direction at terminals of the windings (S1, S2) place the diodes in conducting and blocked states, respectively. A control unit controls open and closed states of the switches at chopping frequency.

Description

ELECTRICAL CONVERTER WITH ENERGY STORAGE

The present invention relates to a switching power converter, energy storage, similar to high-integration Forward type structure converters.

Energy distribution chains, and in particular those massively distributed, use integrated switching converters. State-of-the-art switching converters usually used, such as Forward or Flyback converters, have disadvantages. They generate in particular significant switching noise (a few tens of MHz). In addition, they do not have sufficient integration for some current applications.

In order to overcome the drawbacks of state-of-the-art converters, the invention proposes a switching converter at a frequency f having two input terminals intended to be connected to the poles of an input DC voltage source. voltage Ve, two output terminals for supplying a load with electrical power under a DC output voltage Vs, characterized in that it comprises at least two transformers each having a primary winding and a secondary winding, a main switch, a secondary switch each having a control input for putting said switches in an open state, or in a closed state, a holding capacitor, a first and a second rectifying diodes forming, - a switching primary circuit comprising the main switch connected by one of its ends to one of the input terminals and by its other end to one of the ends lib res of the two primary windings 30 in series with the holding capacity connected by its free end to the other input terminal, the secondary switch being connected by one of its ends to the common point between the main switch and the windings primary series of transformers and by its other end at the end of the holding capacity connected to the other input terminal, - a secondary circuit having the two secondary windings of the transformers each forming with a respective rectifying diode a circuit rectifier connected to the output terminals for supplying a direct current in the load, voltages in one direction across the secondary windings putting one of the two rectifying diodes in the on state, the other rectifying diode being in the blocked state, voltages in the other direction at the terminals of said secondary windings reversing the state of said diodes, control means of the state of the switches at the cutting frequency f.

In one embodiment of the switching converter, the control means are electronic means configured to simultaneously put the two main and secondary switches during a first phase of duration T, one in the open state, the other in the closed state, then during a second following phase of duration (T- 'c) in the inverse states, T being a period of time T = 1 / f.

In another embodiment, the first diode being connected by its cathode to one end of the secondary winding of the first transformer, the second diode is connected by its cathode to one end of the secondary winding of the second transformer, the ends of the connected windings. the cathodes of the diodes being those having an opposite potential, the anodes of the two rectifying diodes being connected to one of the output terminals of the converter, the two other free ends of the secondary of the transformers being connected to the other output terminal of said converter.

In another embodiment, the converter operating in steady state, the transformers work at zero average magnetizing current. A main object of the invention is to provide a converter with a high level of integration.

Another aim is to reduce the switching noise of the converter and thus improve the electromagnetic compatibility.

The invention will be better understood by the description of embodiments of embodiments of converters according to the invention with the aid of indexed drawings in which: FIG. 1 shows an exemplary embodiment of a switching converter according to the invention ; FIGS. 2a and 2b respectively show the control signals of the converter of FIG. 1 as a function of time. FIG. 3 shows a variant of the converter of FIG. 1; FIG. 4a shows a simplified diagram of the converter of FIG. 1 during a first phase of operation; FIG. 4b shows an equivalent diagram of the converter of FIG. 4a during this first phase; - Figure 5a shows a simplified diagram of the figurel converter during a second phase of operation following the first; FIG. 5b shows an equivalent diagram of the converter of FIG. 5a during this second phase; FIG. 6 shows the law of variation of the output voltage of the converter of FIG. 1 as a function of the duty ratio of the control signals of the converter; FIG. 7a shows the variation of the primary current in the converter of FIG. 1; Figure 7b shows the variation of the secondary current in the converter of Figure 1; - Figures 8a and 8b recall the control phases of the switches of the converter of Figure 1.

Figure 1 shows an exemplary embodiment of a switching converter according to the invention. The converter comprises, two input terminals, a first input terminal A and a second input terminal B intended to be connected to the poles of an input DC voltage source Ve and two output terminals C, D a first output terminal C and a second output terminal D for supplying a load R under a DC output voltage Vs. The input voltage cutoff Ve is performed by an inverter circuit (or primary circuit) of the converter comprising a main electronic switch 12, a secondary electronic switch I1, two transformers, a first transformer T1, a second transformer T2 and a holding capacitor C. The first transformer T1 comprises a primary winding (or winding) El of n1 turns and a secondary winding S1 n3 ~ o turns. The transformation ratio m1 of the first transformer Ti is given by the ratio m1 = n3 / n1. The second transformer T2 comprises a primary winding E2 of n2 turns and a secondary winding S2 of n4 turns. The transformation ratio m2 of the second transformer T2 is given by the ratio 15 m2 = n4 / n2. The main switch 12, the primary winding E1 of the first transformer T1, the primary winding E2 of the second transformer T2 and the holding capacitor C form a series circuit connected between the input terminals A and B. The free terminal of the main switch is connected to the first input terminal A and the free terminal of the holding capacitor C being connected to the second input terminal B. The output voltage Vs of the converter is obtained by an output circuit (or secondary circuit) comprising the secondary windings Si, S2 of the two transformers Ti, T2 and two rectifying diodes, a first rectifying diode D1 connected by its cathode to an end 2s of the secondary winding Si of the first transformer and a second rectifying diode D2 connected by its cathode to one end 3s of the secondary winding S2 of the second transformer, the ends of the secondary windings Si, S2 connec The cathodes 30 of the first and second diodes are those having an opposite potential and, in this embodiment, both ends of the nearest secondary windings 2s, 3s. The anodes of the two rectifying diodes D1, D2 are connected together to the second output terminal D of the converter, the two other free ends 1s, 4s of the secondary windings Si, S2 of the transformers T1, T2 being connected to the first terminal of output C of the converter. The output circuit comprises a filtering capacitor Cf connected between the output terminals C, D of the converter.

A control device not shown in the figures provides control signals Cdl1, Cd12 of the respective switches I1, 12 to put them either in an open state or in a closed state according to a first and a second control cycle forming a period For this purpose, the main switch 12 and the secondary switch I1 each have a control input (not shown in the figures). The control input of the secondary switch I1 receives the control signal Cd1, the control input of the main switch 12 receiving the control signal Cd2. FIGS. 2a and 2b show the control signals CdI1 and CdI2 of the switches of the converter of FIG. 1 as a function of time t. The control of the converter is performed at the switching frequency f of period T = 1 / f. With reference to FIGS. 2a and 2b, between a time t0 and a time the secondary switch 11 is controlled to be closed (state 1 in FIG. 2a), the main switch 12 being, during the same period of time, in the complementary state, be open (state 0 in Figure 2b). At the time up to the time T corresponding to a switching period of the converter, the states of the switches are reversed, the main switch 12 being controlled to be closed (state 1 in FIG. 2b), the secondary switch 11 being controlled to be open (state 0 in Figure 2a). The function of the switches is usually performed by field effect transistors, the control input being the gate G of the transistor. FIG. 3 shows a variant of the converter of FIG. 1. The converter of FIG. 3 comprises, in place of the holding capacitor C, a capacitive bridge comprising two capacitors C1, C2 in series connected between the input terminals A. and B, the common point between the two capacitors being connected to the free terminal 4p of the primary windings in series, of the two transformers T1, T2.

We will then describe the operation of the converter of Figure 1 simpler, according to the invention, during the control cycles of the main switch 12, and secondary I1 using diagrams of voltages and currents in the various elements of the converter. For this purpose, the following parameters define the characteristics of the transformers Ti and T2: n1: number of turns of the primary of the first transformer Ti, 10 n2: number of turns of the primary of the second transformer T2, n3: number of turns of the secondary of the first transformer Ti, n4: number of turns of the secondary of the second transformer T2, 15 m1 = n3 / n1: transformation ratio of the first transformer T1, m2 = n4 / n2: transformation ratio of the second transformer T2, Ti: first transformer, 20 T2: second transformer And operating parameters of the circuit is: Is: current delivered by the converter, Im1: magnetizing current in the first transformer Ti seen from the primary of the transformer, Im2: magnetizing current in the second transformer T2 seen from the primary of transformer, VL1 voltage across the primary E1 of the first transformer Ti, 30 VL2 voltage across the primary E2 of the second transformer T2, VL3 voltage across the secondary If the first transformer T1, VL4 voltage across the secondary S2 of the second transformer Ti, Vc: voltage across the holding capacitor C of the converter, T: period of the frequency f of cutting of the converter, ie T = 1 / f For the description that follows, we will consider that the charge of the holding capacitor C is constant which limits the circuits studied during the different operating phases to first order circuits while in the reality is second order resonant circuits (LC type).

FIG. 4a shows a simplified diagram of the converter of FIG. 3 during a first phase of operation, between times t0 and T (see FIGS. 2a and 2b).

Figure 4b shows an equivalent diagram of the converter of Figure 4a during this first phase. In the first phase, the secondary switch 11 is controlled to be closed (state 1) and the main switch 12 to be open (state 0). The voltages VL3, VL4 appearing in one direction across the secondary S1, S2 of the transformers T1, T2 produce the blocking of the first diode D1 and the conduction of the second diode D2. The second transformer T2 supplies a current in the load R connected to the output terminals C, D. The voltage VL2 applied across the inductor L2 of the primary of the second transformer T2 is the output voltage Vs brought back to the primary E2 of the second transformer. T2. The converter of FIG. 4a, during the first phase, is in the form of a closed primary circuit comprising the inductors L1, L2 of the two transformers and the holding capacitance C in series, under the voltage Vc. The output circuit of the converter, in this first phase, comprises the secondary winding S2 of the second transformer in series with the second rectifying diode D2. The first transformer Ti, during this first phase, behaves like a simple self-inducting choke L1 that is comparable to a self-forwarding bus 8 brought back to the primary of the first transformer Ti and consequently it is the self-inductance L1 which sets the variation of the primary current ip ( t) in the primary circuit. FIG. 4b shows an equivalent diagram of the circuit of FIG. 4a during this first phase, comprising a closed circuit with the two inductors L1 and L2 in series with the holding capacitor C traversed by the primary current Ip. An M2 transform ratio transformer M2 connected across the primary inductor L2 of the second transformer T2 represents the transformation of the output voltage Vs into voltage VL2 across the primary inductor L2 of the second transformer T2.

The holding capacitance C is chosen to be large enough so that the resonant frequency fo of the primary circuit is less than the switching frequency of the converter. fo = 1 ù f 2 • r • 1 • C the variation Aip of the primary current ip (for 0 _ <tz) is written: Aip Vc-m2 • Vs = - 20 The variation Aim2 of the magnetising current im2 in the self magnetising of the second transformer T2 (self L2) (for 0 _ <t ≤r) is written: m2 • Vs Aim2 = ù L2 Knowing that: Aip = Aim2 Ais m2

We obtain: L1 25 30 Ais Vc - m2 • Vs m2 • Vs 9 z m2 L1 L2 Dis> 0, the current increases m2 5

FIG. 5a shows a simplified diagram of the converter of FIG. 1 during a second phase of operation of the converter according to the first between times t to T. FIG. 5b shows an equivalent diagram of the converter of FIG. 5a during this second phase.

In the second phase, the secondary switch I1 is open (state 0) and the main switch 12 is closed (state 1). The voltages VL3, VL4, 15 appearing in a direction opposite to that of the first phase, across the secondary S1, S2 of the two transformers T1, T2 produce the blocking of the second diode D2 and the conduction of the first diode D1. The first transformer T1 supplies a current in the load R connected to the output terminals C, D. The inductor L1 again becomes a transformer and the second transformer T2 behaves like a choke L2. The magnetizing current in the second transformer T2 is reversed as for an active clamp converter. The voltage applied across the inductor L1 is the voltage Vs brought back to the primary SI of the first transformer Ti. During this second phase, it is the self L2 of the second transformer T2 which sets the evolution of the primary current ip (t). The converter of FIG. 5a, during the second phase, is in the form of a closed primary circuit comprising the inductors L1, L2, the holding capacitor C under the voltage Vc and a generator E1 of the input voltage Ve serial. The output circuit of the converter, in this second phase, comprises the secondary winding SI of the first transformer Ti in series with the first rectifying diode D1.

The second transformer T2 behaves, during this second phase, as a simple self L2 self equivalent to a self forward brought back to the primary of the second transformer T2 and therefore it is the self L2 which sets the variation of the primary current ip (t) .

FIG. 5b represents an equivalent diagram of the circuit of FIG. 5a during this second phase, comprising a closed circuit with the two inductors L1 and L2 and a generator E2 of voltage Ve-Vc in series. A transformation ratio transformer M1 connected to the terminals of the primary inductor of the first transformer T1 represents the transformation of the output voltage Vs into voltage VL1 across the primary inductor L1 of the first transformer T1. The holding capacitance C is large enough for the resonant frequency fo of the primary circuit to be smaller than the switching frequency f. fo = 1 f 2 • ~ c • '• • C

the variation Aip of the primary current ip (for r <_ t T 0) is written: Aip _ Ve - Vc - ml • Vs T ûz L2

The variation Aim1 of the magnetising current im1 in the magnetising self of Ti (self L1) (for z 5 t T) is written:

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d being the duty cycle of a control signal during the duration period T.

Starting from equation (1) we obtain: ml • Vs Vc-m2 • Vs L1 (T ~) = L1 • z ie: Vc = ml • Vs (3) d Starting from equation (2) we get: m2 • Vs Ve-Vc-ml Vs • (T -7-) that is: L2 L2 Vs = (VeùVc) • 1 ù d (4) m It can be deduced from expressions (3) and (4): Vs = Ve • d • (1ûd) (5) m using equation (5) equation (3) becomes: Vc = Ve • (1 - d) We can then deduce the maximum voltage Vc, ie Vcmax at FIG. 6 shows the law of variation of the output voltage Vs of the converter of FIG. 1 (or the product m.Ve) as a function of the duty cycle d. It can be seen in FIG. 6 that the output voltage Vs for a ratio m and a determined input voltage Ve goes through a maximum for d = 0.5.

If it is desired to regulate the output voltage Vs constant as a function of a variable input voltage Ve, the transformation ratio m 10 of the transformers will have to be determined as a function of the minimum input voltage Vemin applied to the input of the converter for a maximum duty cycle dmax is: Vs dmax. V emin

The maximum duty cycle, dmax, corresponds to the Vemin input voltage.

FIG. 7a shows the variation of the primary current in the converter of FIG. 1 and FIG. 7b shows the variation of the secondary current in the converter of FIG. 1. FIGS. 8a and 8b recall the control phases of the main switches 12 and secondary 11 of the converter of FIG.

During the first phase, between the times t = 0 and t = -r, the secondary switch 11 is closed and the main switch 12 is open. The primary current ip increases by Aim2 then decreases abruptly at the moment of the control reversal of the switches 11, 12, then, during the second phase between the times t = i, and t = T decreases, reversing with sign, 30 of Aim1, then the cycle starts again. The current is at the output of the converter varies around a mean value with a ripple of Ais.

Subsequently is explained the search of the operating point of the converter for a variation of the output current Ais = 0: m 15 by: The current seen from the primary, during the first phase is given Ais = Aip-Aim2 or Dis = Vc - m • Vs ûm • Vs LL Ais = 0 for Vc-2.m.Vs = 0 m • Vs d -2 • m • Vs = 0 from or: d = 1/2 which corresponds to minimum Ve.

The converter topology described according to the invention is comparable to a bridge resonance converter, energy recovery. The originality of this structure lies, among other things, in the fact that the Forward inductance, normally located at the secondary of the transformer, is brought back to the primary. This inductor is actually a transformer, which in the restitution phase (main switch 12 closed) can recover the energy stored during the Forward phase (as does the self in a Forward 20 state of the art). The holding voltage Vc (or clamp in English) is significantly reduced (by a factor of 2) and the switching losses become negligible. Another advantage of the converter according to the invention is that the voltage appearing across the primary of each of the transformers T1, T2 is a reduced voltage value m.Vs furthermore the average magnetizing current is zero centered on a zero current and symmetrical . The stored energy and the amplitude of the magnetizing current are therefore lower than in state-of-the-art converters and consequently the volume of the transformer can be reduced.

When one of the transformers supplies an output current, the other transformer operates in self, under the same conditions as a Forward inductance. The transformer supports a voltage Ve-m.Vs instead of Ve / m-Vs with however a significantly reduced current of a transformation factor m.

In the case of a step-down converter according to the invention, and in the case of a high output current, the winding of the transformers are not dimensioned as a function of the losses in the copper of the windings, as in the case of a converter. Forward of the state of the art, but only according to the energy stored in the magnetic circuit. The operation of the converter in soft switching is facilitated by the low amplitude of the voltages across the switches between Ovolt and the input voltage Ve. The voltage resistance of the switches and the holding capacitance C is lower than in the active clamp converters. The structure with two transformers according to the invention is well suited for producing single-voltage converters with high output current.

Claims (9)

1. Switching converter at a frequency f having two input terminals (A, B) intended to be connected to the poles of a voltage input DC voltage source Ve, two output terminals (C, D) for supplying a load (R) with an electric power under a DC output voltage Vs, characterized in that it comprises at least two transformers (T1, T2) each having a primary winding (El, E2) and a secondary winding (SI , S2), a main switch (12), a secondary switch (I1) each having a control input for putting said switches in an open state, or in a closed state, a holding capacity (C), a first ( D1) and a second (D2) rectifying diode forming - a switching primary circuit having the main switch 12 connected at one of its ends to one of the input terminals and at its other end to one of the free ends of the two windings imaires (E1, E2) in series with the holding capacitance (C) connected by its free end to the other input terminal, the secondary switch (I1) being connected by one of its ends to the common point between main switch (12) and the primary windings in series of the transformers and at its other end at the end of the holding capacitance (C) connected to the other input terminal, - a secondary circuit comprising the two secondary windings (Si, S2) transformers each forming with a respective rectification diode (D1, D2) a current rectifying circuit connected to the output terminals (C, D) for supplying a direct current in the load (R), voltages (VL3, VL4) in one direction at the terminals of the secondary windings (SI, S2) putting one of the two rectifying diodes in the on state, the other rectifying diode being in the locked state, voltages in the other sense at the terminals of said dry windings transforming the state of said diodes, - control means of the state of the switches at the cutting frequency f. 2. Switching converter according to claim 1, characterized in that the control means are electronic means configured to simultaneously put the two main switches (12) and secondary (I1) during a first phase of duration z, the one in the open state, the other in the closed state, and then during a second subsequent phase of duration (T-T) in the inverse states, T being a period of time T = 1 / f. 3. Switching converter according to one of claims 1 or 2, characterized in that the first diode (D1) being connected by its cathode to an end 2s of the secondary winding (Si) of the first transformer (Ti), the second diode (D2) is connected by its cathode to one end 3s of the secondary winding (S2) of the second transformer, the ends of the windings connected to the cathodes of the diodes being those having an opposite potential (2s, 3s), the anodes of the two rectifying diodes being connected to one of the output terminals of the converter (C), the two other free ends (1s, 4s) of the secondary of the transformers being connected to the other output terminal of said converter (D). 4. Switching converter according to one of claims 1 to 3 characterized in that in steady state operation, transformers Ti, T2 work at zero average magnetizing current, im1 being the magnetizing current in the first transformer Ti, im2 being the magnetizing current in the second transformer T2, 5. Switching converter according to claim 4, characterized in that in steady-state operation: during the first phase of operation of the converter for 0 <t <r, the absolute value of the variation Aip of the current ip in the circuit primary must be equal to the absolute value of the variation Aim1 of the magnetising current im1 of the first transformer (Ti) and, during the second phase of operation of the converter for • r <t <T, the absolute value of the variation Aip of the current in the primary circuit must be equal to the absolute value of the variation Aim2 of the magnetising current im2 of the second transformer (T2). expressed as equations: I Aip I (first phase) = Aim1 1 (second phase) (1) Aim2 1 (first phase) = Aip 1 (second phase) (2) 10 6. Switching converter according to one of claims 4 or 5, characterized in that, the ratio of transformations m and the values of the inductors L1, L2 of the two transformers T1, T2 being identical is m1 = m2 = m 15 and LI = L2 = L the duration of the first phase being expressed by: = dT d being the duty cycle of a control signal during the duration period T. the output voltage Vs and the input voltage Ve of the converter are expressed by: Vs = Ve • d • (1 d) (5) m 7. Switching converter according to claim 6, characterized in that the voltage across the holding capacitor (C) is given by: Vc = Ve (1 - d) the maximum voltage Vcmax across the capacitance of maintaining (C) being then: 35 Vcmax = Ve Switching converter according to one of Claims 6 to 7, characterized in that the transformation ratio m of the transformers is determined as a function of the minimum input voltage Vemin applied to the input of the converter for a maximum duty cycle. dmax is: Vs m = ù dmax.Vemin 9. Switching converter according to claim 1 to 8, characterized in that the holding capacitance (C) is chosen to be sufficiently large so that the resonant frequency fo of the primary circuit is lower than the frequency of the converter, f fo = 1 ù f 2 • 7r • i • C the first transformer T1 being assimilable to a self LI brought back to the primary of the first transformer T1 and fo = 1 ù f 2.7r •, fi -,: a .0 the second transformer T2 being comparable to an inductor L2 brought back to the primary of the second transformer T2.25