CN117914202A - Full-speed domain rotor information estimation method of permanent magnet synchronous motor - Google Patents
Full-speed domain rotor information estimation method of permanent magnet synchronous motor Download PDFInfo
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Abstract
Compared with the traditional estimation method without a position sensor, the method for estimating the rotor information of the full-speed domain of the permanent magnet synchronous motor utilizes an average weighting scheme to combine a high-frequency injection scheme with a nonlinear flux linkage observer scheme, so that the problem of overlarge rotating speed pulsation in the switching process of low speed, medium speed and high speed is avoided; in the medium-high speed range, three-phase current of the motor is monitored, current and voltage under a two-phase static coordinate system are obtained through transformation, a nonlinear flux linkage observer is constructed, rotor flux linkage based on the two-phase coordinate system is obtained to observe the position and the speed of the rotor, and speed closed-loop control can be directly realized in the medium-high speed range, so that the motor stably operates.
Description
Technical Field
The invention relates to the technical field of permanent magnet synchronous motor control, in particular to a full-speed domain rotor information estimation method of a permanent magnet synchronous motor.
Background
The permanent magnet synchronous motor has the characteristics of large power factor, high torque inertia ratio, high power density, simple structure, convenient maintenance and the like, and is widely applied to the fields of servo systems, industrial control and the like. In the control of a permanent magnet synchronous motor, in order to observe the position and speed information of a rotor, a mechanical sensor (a hall sensor, a rotary transformer, a grating encoder and the like) is required to be installed on the motor so as to realize closed-loop control of the motor. The motor body is additionally provided with the position sensor, so that the motor volume is increased, the structure of a control system is more complicated, the maintenance is inconvenient, the stability of the system is reduced, and the anti-interference capability is deteriorated; meanwhile, the sensor has specific applicable conditions, can only operate under the appropriate conditions, and cannot be normally used under some severe environments. Therefore, a control method without a position sensor is adopted by many permanent magnet synchronous motors at present, so that the cost of the motors is reduced, and the stability of the system is improved.
Currently, field Oriented Control (FOC) is one of the common methods of permanent magnet synchronous motor control. Through magnetic field directional control, stator current of the permanent magnet synchronous motor can be decoupled into exciting current and torque current, and the exciting current and the torque current are respectively controlled, so that accurate control of torque and speed of the motor is realized. The magnetic field orientation control requires knowledge of the rotor position and speed information to accurately control the excitation current and torque current; meanwhile, according to the position and speed information of the rotor, the speed ring and the current ring of the motor can be correspondingly regulated and controlled. In order to realize accurate control of the permanent magnet synchronous motor, accurate rotor position and speed information needs to be acquired by adopting a proper estimation method and applied to magnetic field orientation control.
The main stream estimation method of the rotor information of the permanent magnet synchronous motor is a sliding mode observer, the sliding mode observer utilizes the back electromotive force information of the motor, the motor has good observation effect at medium and high speeds, the control effect is not ideal at low speeds, open loop control is generally adopted, and the rotor position needs to be positioned during starting. The flux linkage observer can be directly and rapidly started in a closed loop, and the control effect at a low speed is better than that of a sliding mode observer. Depending on the type of motor used, there are two classes of flux linkage observers based on voltage models and current models.
For a voltage model flux linkage observer, because the voltage model flux linkage observer is irrelevant to rotor resistance and does not need motor rotation speed information, the voltage model flux linkage observer is particularly suitable for a vector control system without a speed sensor, but comprises a pure integral term, the initial phase and direct current bias of the integral term can influence an integral result, and the observer performance is poor at a low speed.
For a current model flux linkage observer, the current model flux linkage observer is used for observing based on the relation between the rotor flux linkage and the stator current, and has a simpler structure; however, the current model flux linkage observer has strong dependence on motor parameters, and if the motor parameters are inaccurate or change, the observation performance of the current model flux linkage observer can be affected.
Disclosure of Invention
Aiming at the technical defects existing in the control of the existing permanent magnet synchronous motor, the invention provides a full-speed domain rotor information estimation method of the permanent magnet synchronous motor.
The invention provides a method for estimating rotor information of a full-speed domain of a permanent magnet synchronous motor,
When the motor rotation speed omega is less than or equal to omega l, a high-frequency injection scheme is adopted, a high-frequency voltage signal is injected to the d axis in a two-phase rotation coordinate system, and sinusoidal current for motor rotation control is generated after iPak conversion and SVPWM module; sampling to obtain a current component i a、ib、ic under a three-phase static coordinate system, then obtaining a current component i d、iq under a two-phase rotating coordinate system through Clark conversion and Park conversion, and multiplying a current component i q of a q-axis with a modulation signal cos omega ht·sin2ωh t after bandwidth filtering to obtain a high-frequency current componentWherein ω h is the frequency of the injected high-frequency voltage signal; then the rotor speed/>, is obtained after low-pass filtering and PI regulationControl/>Zero, so that the error of the estimated value and the actual value of the rotor position/>Gradually becomes 0, and the rotor position/> is obtained through an integrator
When the motor rotation speed omega is more than omega h, a nonlinear flux linkage observer scheme is adopted, a current component i a、ib、ic under a three-phase static coordinate system is obtained by sampling, a current component i α、iβ under a two-phase static coordinate system obtained by Clark conversion and a voltage component u α、uβ under the two-phase static coordinate system output by iPark conversion are used as inputs of a flux linkage observer, state variables and output variables of the observer are defined, vector function values of the state variables are calculated, a difference value between an estimated flux linkage amplitude and an actual flux linkage amplitude is used as an estimated flux linkage component compensation term, and a rotor position is obtained according to the flux linkageAnd speed/>At the time of obtaining the rotor positionAnd speed/>Generating sinusoidal current for motor rotation control through an iPak conversion and SVPWM module;
When the rotating speed of the motor is omega l＜ω≤ωh, an average weighting scheme is adopted, and a high-frequency injection scheme is combined with a nonlinear flux linkage observer scheme to obtain the position and speed information of the rotor so as to ensure the smooth rotation of the rotor in the transition stage.
Further, in the high-frequency injection scheme, the d-axis injection high-frequency voltage signal in the two-phase rotation coordinate system is u=k·ucos ω h t, where K is a weight coefficient of the high-frequency injection scheme when the motor rotation speed is ω l＜ω≤ωh, and U is an amplitude of the injected high-frequency voltage signal.
Further, in the scheme of the nonlinear flux linkage observer, a flux linkage equation under a two-phase static coordinate system is defined as a state variable, and observation, derivation and analysis are carried out on flux linkage of the motor; taking the derivative of the state variable as an output variable, taking the estimated state variable as a vector function, constructing an observer to obtain a nonlinear flux linkage observer equation, and then performing discretization to obtain a processing equation executable by a program.
Further, the flux linkage under the two-phase stationary coordinate system is calculated through the observation of the flux linkage by the nonlinear flux linkage observer, and then the current rotor angle is calculated.
Further, a phase-locked loop PLL is adopted to obtain the rotor position, and a sum and difference formula is adoptedWhere θ is the rotor actual position,/>For the estimated position of the rotor, when the estimated position coincides with the actual position, then/>And taking the angle difference as an error input of the PI controller, and finally locking the actual position when the position estimation approaches to the actual position and the error approaches to 0.
Further, weight coefficients by high frequency injection schemeThe weighting coefficients of the flux linkage observer scheme are 1-K.
The invention also protects a full-speed domain magnetic field directional control method of the permanent magnet synchronous motor, and rotor information estimation is carried out based on the full-speed domain rotor information estimation method of the permanent magnet synchronous motor.
Compared with the traditional estimation method without a position sensor, the estimation method for the rotor information of the full-speed domain of the permanent magnet synchronous motor provided by the invention combines a high-frequency injection scheme with a nonlinear flux linkage observer scheme by utilizing an average weighting scheme, so that the problem of overlarge rotating speed pulsation in the switching process of low speed, medium speed and high speed is avoided; in the medium-high speed range, three-phase current of the motor is monitored, current and voltage under a two-phase static coordinate system are obtained through transformation, a nonlinear flux linkage observer is constructed, rotor flux linkage based on the two-phase coordinate system is obtained to observe the position and the speed of the rotor, and speed closed-loop control can be directly realized in the medium-high speed range, so that the motor stably operates.
Drawings
FIG. 1 is a flow chart of a high frequency injection scheme disclosed in example 1;
FIG. 2 is a phase locked loop flow diagram;
FIG. 3 is a flow chart of a nonlinear flux linkage observer scheme disclosed in example 2;
FIG. 4 is a full-speed domain rotor information estimation flow chart disclosed in embodiment 4;
Fig. 5 is a schematic diagram of weight coefficients in full-speed domain rotor information estimation.
Detailed Description
The invention will be described in further detail with reference to the drawings and the detailed description. The embodiments of the invention have been presented for purposes of illustration and description, and are not intended to be exhaustive or limited to the invention in the form disclosed. Many modifications and variations will be apparent to those of ordinary skill in the art. The embodiments were chosen and described in order to best explain the principles of the invention and the practical application, and to enable others of ordinary skill in the art to understand the invention for various embodiments with various modifications as are suited to the particular use contemplated.
The present embodiment defines a motor speed lower than ω l at low speed and higher than ω h at medium and high speed, with a transition range therebetween, where in practice ω l is typically 300r/min and ω h is typically 400r/min.
The two-phase rotating coordinate system, also called dq coordinate system, is a commonly used motor coordinate system. In a two-phase rotating coordinate system, the d-axis and q-axis are two coordinate axes that are spatially fixed relative to each other, but rotate as the motor rotor rotates. The d-axis is along the direction of the rotor magnetic field, while the q-axis is perpendicular to the d-axis. Under a two-phase rotating coordinate system, the electric quantity such as voltage, current and flux linkage of the motor can be decomposed into two components of d axis and q axis. The excitation current and the torque current of the motor can be separated by decoupling the voltage, the current, the flux linkage and other electrical quantities of the motor in a two-phase rotation coordinate system, so that the independent control of the motor is realized.
A two-phase stationary coordinate system, also known as the αβ coordinate system, is a relatively stationary coordinate system, similar to a three-phase stationary coordinate system. In a two-phase stationary coordinate system, the α -axis and the β -axis are two mutually perpendicular coordinate axes, whose positions are fixed relative to the three-phase stationary coordinate system, but independent of the rotation of the motor rotor. Under a two-phase static coordinate system, the electric quantity such as voltage, current and magnetic linkage of the motor can be decomposed into two components of an alpha axis and a beta axis. The two-phase stationary coordinate system can in some cases provide a simpler motor model and control method than the three-phase stationary coordinate system.
Example 1
When the motor rotation speed omega is less than or equal to omega l, a high-frequency injection scheme is adopted to estimate the rotor positionAnd rotor speed/>
The voltage equation of the motor under the two-phase rotation coordinate system is thatWherein u d、uq is the voltage component of the d axis and the q axis under the two-phase rotation coordinate system, i d、iq is the current component of the d axis and the q axis under the two-phase rotation coordinate system, L d、Lq is the inductance component of the d axis and the q axis under the two-phase rotation coordinate system, R s is the stator resistance, omega e is the electrical angular velocity, p is the differential operator, and psi f is the permanent magnet flux linkage.
Because the motor is in a low-speed state at this time, the counter electromotive force can be ignored, and the impedance generated by the inductance of the motor is far greater than the resistance, the voltage equation can be simplified into
Converting the simplified voltage equation into a current-voltage relation under the dq coordinate system estimated by high-frequency injection by using Park conversion and iPak conversion
Wherein/>Is the average inductance in a two-phase rotating coordinate system,/>Is half difference inductance under two-phase rotation coordinate system,/>An error between the estimated position and the actual position of the rotor.
Injecting a high-frequency voltage signal u=ucos ω h t on the d-axis in the estimated two-phase rotation coordinate system, the current change caused by the high-frequency voltage signal isWhere ω h is the frequency of the injected high-frequency voltage signal and U is the amplitude of the injected high-frequency voltage signal.
As can be seen from the current transformation formula caused by the high-frequency voltage signal, under the estimated two-phase rotation coordinate system, the amplitudes of the high-frequency current components of the d axis and the q axis are both identical to the error of rotor position estimationRelated to the following.
Referring to fig. 1, the high-frequency injection scheme is to inject a high-frequency voltage signal into a d-axis in a two-phase rotating coordinate system, and generate sinusoidal current for motor rotation control after iPark transformation and SVPWM module; then sampling to obtain a current component i a、ib、ic under a three-phase stationary coordinate system of the motor, obtaining a current component i d、iq under a two-phase rotating coordinate system through Clark conversion and Park conversion, multiplying a current component i q of a q-axis by a modulation signal cos omega ht·sin2ωh t after bandwidth filtering (BPF) to obtain a high-frequency current componentObtaining the rotor speed/>, after Low Pass Filtering (LPF) and PID adjustmentControl/>Zero, so that the error of the estimated value and the actual value of the rotor position/>Gradually becomes 0, and the rotor position is obtained by a phase-locked loop (PLL)
The phase-locked loop flow chart is shown in FIG. 2, and the phase-locked loop obtains the rotor positionIs characterized by comprising the following working principle: according to the sum and difference formulaWhere θ is the rotor actual position,/>For the estimated position of the rotor, when the estimated position is consistent with the actual position, the sum and difference formula is utilized to obtain/>And taking the angle difference as an error input of the PI controller, and finally locking the actual electrical angle, namely the actual position when the position estimation approaches to the actual position and the error approaches to 0.
Example 2
When the motor rotation speed omega is more than omega h, a nonlinear flux linkage observer is adopted to observe the rotor positionAnd rotor speed
The control flow of the flux linkage observer is shown in figure 3, the collected current component i a、ib、ic under the three-phase static coordinate system is subjected to Clark transformation to obtain a current component i α、iβ under the two-phase static coordinate system, the current component u α、uβ under the two-phase static coordinate system output by the iPark transformation is taken as the input of the flux linkage observer, the state variable and the output variable of the observer are defined, the vector function value of the state variable is calculated, the difference value of the estimated flux linkage amplitude and the actual flux linkage amplitude is taken as the flux linkage component compensation item obtained by estimation, and the rotor position is obtained according to the flux linkageAnd speed/>In obtaining rotor position/>And speed/>After that, sinusoidal current for motor rotation control is generated through the iPark conversion and SVPWM module.
The voltage equation of the motor in the two-phase stationary coordinate system is as follows:
Equation analysis is carried out on the flux linkage of the motor under the two-phase rotating coordinate system, and the flux linkage equation under the two-phase rotating coordinate system can be converted into the two-phase static coordinate system according to the iPak conversion Wherein, psi α、ψβ is the component of alpha axis and beta axis under two-phase static coordinate system, L s is the stator inductance.
IPark conversion formula combined with currentThe flux linkage equation/>, under the two-phase static coordinate system, can be obtainedAfter determining the flux linkage equation in the two-phase stationary coordinate system, the state variables and output variables of the observer are defined.
Firstly, defining a flux linkage equation under a two-phase static coordinate system as a state variable, and carrying out observation, derivation and analysis on flux linkage of a motor to obtain
By comparing with the voltage equation of the motorTaking the derivative of the state variable as an output variable, taking the estimated state variable as a vector function, and constructing an observer to obtain a nonlinear flux linkage observer equationWhere gamma is the gain of the observer,Respectively, the estimated value of the state variable, namely the flux linkage component under the two-phase static coordinate system,/>Is the first derivative of the state variable estimate.
Discretizing the nonlinear flux linkage observer equation to obtain a program executable processing equation:
the flux linkage under a two-phase static coordinate system is calculated through the observation of the flux linkage by a nonlinear flux linkage observer, and the current rotor angle is calculated through arctangent
A phase-locked loop is also used to extract rotor position information, avoiding the influence of noise and high frequency jitter present in the flux linkage estimated by the flux linkage observer on the angle estimation. And taking the angle difference as an error input of the PI controller, and when the angle estimation approaches the real angle, the error continuously approaches 0, and finally locking the actual electrical angle. In order to reduce the calculation, PI is not divided by the input, but the rotor speed is obtained by controlling the closed-loop phase-locked loop by adjusting the parameters of PIAnd then integrating to obtain the rotor angle/>
Example 3
When the motor speed is omega l＜ω≤ωh, a high-frequency injection scheme is combined with a nonlinear flux linkage observer scheme by adopting an average weighting scheme, so that the rotor position and the rotor speed are obtained.
Setting the weight coefficient of the high-frequency injection scheme asThe weighting coefficient of the flux linkage observer scheme is (1-K), and the final rotor position and rotor speed value are obtained by fusion
Since the weight coefficient of the high-frequency injection scheme is set to be K, when the rotor position and the rotor speed are obtained by adopting the high-frequency injection scheme, the high-frequency voltage signal is injected to the d-axis in the two-phase rotation coordinate system as u=k·ucos ω h t. With the rise of the rotating speed of the rotor, the amplitude of the injected high-frequency voltage signal is reduced, the influence of the high-frequency voltage signal on the high-speed nonlinear flux linkage observer is reduced, the motor control precision is improved, and the motor noise is reduced.
Example 4
The embodiment provides a method for estimating full-speed domain rotor information of a permanent magnet synchronous motor, the flow is shown in fig. 4, the methods of embodiments 1-3 are combined, and the weight coefficient of a high-frequency injection scheme is adoptedObtain rotor position/>And rotor speed/>As shown in fig. 5.
When the motor rotation speed omega is less than or equal to omega l, the weight coefficient K=1 of the high-frequency injection scheme, the weight coefficients 1-K=0 of the flux linkage observer scheme, and the rotor position and the rotor speed are estimated by using the high-frequency injection scheme only;
when the motor rotating speed omega is larger than omega h, the weight coefficient K=1 of the high-frequency injection scheme and the weight coefficient 1-K=1 of the flux linkage observer scheme are used for estimating the rotor position and the rotor speed by using the nonlinear flux linkage observer;
When the rotating speed of the motor is omega l＜ω≤ωh, estimating a weight coefficient, and simultaneously, changing the amplitude of a high-frequency voltage signal injected at high frequency along with the weight coefficient, and estimating rotor information; after the estimation of the rotor position and the rotor speed is completed under different rotating speeds, the motor is controlled according to a magnetic field orientation control method, so that the motor can smoothly rotate in a full speed range.
It will be apparent that the described embodiments are only some, but not all, embodiments of the invention. All other embodiments, which can be made by those skilled in the art and which are included in the embodiments of the present invention without the inventive step, are intended to be within the scope of the present invention.
Claims (7)
1. A method for estimating the information of a full-speed domain rotor of a permanent magnet synchronous motor is characterized in that,
When the motor rotation speed omega is less than or equal to omega l, a high-frequency injection scheme is adopted, a high-frequency voltage signal is injected to the d axis in a two-phase rotation coordinate system, and sinusoidal current for motor rotation control is generated after iPak conversion and SVPWM module; sampling to obtain a current component i a、ib、ic under a three-phase static coordinate system, then obtaining a current component i d、iq under a two-phase rotating coordinate system through Clark conversion and Park conversion, and multiplying a current component i q of a q-axis with a modulation signal cos omega ht·sin2ωh t after bandwidth filtering to obtain a high-frequency current componentWherein ω h is the frequency of the injected high-frequency voltage signal; then the rotor speed/>, is obtained after low-pass filtering and PI regulationControl/>Zero, so that the error of the estimated value and the actual value of the rotor position/>Gradually becomes 0, and the rotor position/> is obtained through an integrator
When the motor rotation speed omega is more than omega h, a nonlinear flux linkage observer scheme is adopted, a current component i a、ib、ic under a three-phase static coordinate system is obtained by sampling, a current component i α、iβ under a two-phase static coordinate system obtained by Clark conversion and a voltage component u α、uβ under the two-phase static coordinate system output by iPark conversion are used as inputs of a flux linkage observer, state variables and output variables of the observer are defined, vector function values of the state variables are calculated, a difference value between an estimated flux linkage amplitude and an actual flux linkage amplitude is used as an estimated flux linkage component compensation term, and a rotor position is obtained according to the flux linkageAnd speed/>In obtaining rotor position/>And speed/>Generating sinusoidal current for motor rotation control through an iPak conversion and SVPWM module;
When the rotating speed of the motor is omega l＜ω≤ωh, an average weighting scheme is adopted, and a high-frequency injection scheme is combined with a nonlinear flux linkage observer scheme to obtain the position and speed information of the rotor so as to ensure the smooth rotation of the rotor in the transition stage.
2. The method of claim 1, wherein in the high frequency injection scheme, the d-axis injection of the high frequency voltage signal into the two-phase rotating coordinate system is u=k·ucos ω h t, where K is a weight coefficient of the high frequency injection scheme when the motor speed is ω l＜ω≤ωh, and U is an amplitude of the injected high frequency voltage signal.
3. The method for estimating full-speed domain rotor information of a permanent magnet synchronous motor according to claim 1, wherein in the scheme of a nonlinear flux linkage observer, a flux linkage equation under a two-phase stationary coordinate system is defined as a state variable, and observation derivative analysis is performed on flux linkage of the motor; taking the derivative of the state variable as an output variable, taking the estimated state variable as a vector function, constructing an observer to obtain a nonlinear flux linkage observer equation, and then performing discretization to obtain a processing equation executable by a program.
4. The method for estimating full-speed domain rotor information of permanent magnet synchronous motor according to claim 3, wherein the flux linkage under a two-phase stationary coordinate system is calculated by observing flux linkage by a nonlinear flux linkage observer, so as to calculate the current rotor angle.
5. The method for estimating full-speed domain rotor information of permanent magnet synchronous motor according to claim 1, wherein the rotor position is obtained by using a phase-locked loop PLL according to a sum-difference formulaWhere θ is the rotor actual position,/>For the estimated position of the rotor, when the estimated position coincides with the actual position, then/>And taking the angle difference as an error input of the PI controller, and finally locking the actual position when the position estimation approaches to the actual position and the error approaches to 0.
6. The method for estimating full-speed domain rotor information of a permanent magnet synchronous motor according to claim 1, wherein the weight coefficients are obtained by a high-frequency injection schemeThe weighting coefficients of the flux linkage observer scheme are 1-K.
7. A method for controlling the orientation of a full-speed domain magnetic field of a permanent magnet synchronous motor, characterized in that the rotor information is estimated based on the method for estimating the full-speed domain rotor information of the permanent magnet synchronous motor according to any one of claims 1 to 6.
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