CN113644654A - Active damping method based on LCL active filter network side current feedback - Google Patents

Active damping method based on LCL active filter network side current feedback Download PDF

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CN113644654A
CN113644654A CN202110504225.0A CN202110504225A CN113644654A CN 113644654 A CN113644654 A CN 113644654A CN 202110504225 A CN202110504225 A CN 202110504225A CN 113644654 A CN113644654 A CN 113644654A
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network side
resonance
active
lcl
gain
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CN113644654B (en
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陈冬冬
林燕强
许振明
吴正炀
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Minnan University of Science and Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/01—Arrangements for reducing harmonics or ripples
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H02J3/381—Dispersed generators
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/20—Active power filtering [APF]

Abstract

The invention discloses an active damping method based on LCL active filter network side current feedback, which relates to the field of power technology application and comprises the following steps: collecting a network side voltage signal and a network side current signal of the LCL active filter in each sampling period; introducing a transfer function into the acquired signal, and then calculating the component gain of the system near the resonant frequency through Clark coordinate transformation; the component gain near the resonance frequency is brought into a phase-shifting link, so that the damping control of the resonance peak can be realized. By adopting the technical scheme, the network side inductive current is sampled and is taken as a feedback quantity, and the active damping component is extracted and controlled, so that the function of resonance suppression is realized, the hardware cost can be saved, the problem of system reliability reduction possibly caused by a sensor and a peripheral circuit is avoided, the active damping function can be realized on the premise of not influencing the system performance, and the method has obvious significance and engineering practical value.

Description

Active damping method based on LCL active filter network side current feedback
Technical Field
The invention relates to the field of power technology application, in particular to an active damping method based on LCL active filter network side current feedback.
Background
The output of an Active Power Filter (APF) is usually filtered by a LCL filter, which naturally has a resonant peak. The LCL parameter design of the APF module checks the frequency point of the resonance peak, and generally makes the frequency point between the cut-off frequency of the pass band of the controller and 1/2 times of the switching frequency. In order to ensure reliable operation of the system, relevant measures must be taken to suppress resonance. Common methods for eliminating resonance have been disclosed in the literature and fall into two broad categories: passive damping methods and active damping methods.
The basic idea of the passive damping method is to attenuate the resonance peaks by connecting resistors in series or in parallel on the individual components of the LCL filter. The specific resistor placement is shown in fig. 1. Generally, the larger the damping resistance value, i.e. the larger the damping ratio, the greater the damping peak damping force, but the damping ratio should not be too large, otherwise the dynamic response speed of the system is affected. The passive damping structure is simple and easy to realize, and the practical application is limited due to the loss and serious heating of the damping resistor.
The basic idea of the active damping method is to suppress the resonance peak by introducing a control loop to make the controlled object generate an equivalent damping effect. Since active damping has no physical resistor device, no loss is generated, so the method is also called as a virtual damping method. Comparing the typical virtual damping based on filter capacitor voltage feedback with the virtual damping based on filter capacitor current feedback, the corresponding control block diagrams are shown in fig. 2 and 3. H1(s) and H2(s) are feedback transfer functions. The active damping method has the advantage of avoiding the problems of loss and heating caused by passive damping. The method has the disadvantages that sensor feedback control variables need to be added, and the method is sensitive to environmental parameters, so that the reliability and robustness of the system are reduced.
Considering that the controller samples the network side inductive current to complete closed-loop control, and if the current can be used as a feedback quantity to extract and control the active damping component, so as to realize the function of resonance suppression, the hardware cost can be saved, the problem of system reliability reduction possibly caused by a sensor and a peripheral circuit can be avoided, and the active damping function can be realized on the premise of not influencing the system performance, so that the invention has obvious significance and engineering practical value, and the invention can be used for researching the active damping function.
Disclosure of Invention
The invention aims to solve the technical problems that an active damping method based on LCL active filter network side current feedback is provided, and the passive damping method in the prior art is serious in resistance loss and heating and the system reliability and robustness of the active damping method are reduced.
In order to solve the technical problems, the technical scheme of the invention is as follows:
an active damping method based on LCL active filter network side current feedback comprises the following steps:
step one, acquiring a network side voltage signal and a network side current signal of an LCL active filter in each sampling period;
step two, introducing a transfer function according to the signal acquired in the step one, and then calculating the component gain of the system near the resonant frequency through Clark coordinate transformation;
and step three, bringing component gain near the resonance frequency obtained in the step two into a phase-shifting link, and realizing damping control on the resonance peak.
In the second step, the component gain of the system near the resonant frequency is calculated as specifically shown in formula (1):
Figure BDA0003057681150000021
in the second step, when calculating the component gain of the system near the resonant frequency, the tuning controller is also adjusted to extract the high frequency component without affecting the harmonic gain of the low frequency band.
Specifically, the following steps are performed when the alignment resonance controller is adjusted:
step 2.1, selecting omega according to the requirement of cut-off frequency bandwidthc；
Step 2.1, designing K according to the requirement of resonance peak value gaingTherefore, the steady-state performance and the anti-interference capability of the system can be optimized.
Specifically, the parameters of the quasi-resonant controller are calculated by the following formula (2):
Figure BDA0003057681150000022
wherein, KgThe resonant gain, omega, of a quasi-resonant controllercFor the resonance bandwidth, ωgIs the resonance bandwidth of the controlled object.
Specifically, in order to optimize both the steady-state performance and the anti-interference capability of the system, the resonance bandwidth ω is taken from the above formulac5rad/s, resonant gain Kg＝100。
Wherein a virtual impedance is connected in parallel to the filter capacitor, the virtual impedance being calculated by the following equation (3):
Figure BDA0003057681150000023
wherein Z isvIs a virtual impedance, L1Is an inverter-side filter inductor, L2For the grid-side filter inductance, GvIs a current regulator, KpwmThe gain of a bridge arm of the grid-connected inverter is obtained.
By adopting the technical scheme, the network side inductive current is sampled and is taken as a feedback quantity, and the active damping component is extracted and controlled, so that the function of resonance suppression is realized, the hardware cost can be saved, the problem of system reliability reduction possibly caused by a sensor and a peripheral circuit is avoided, the active damping function can be realized on the premise of not influencing the system performance, and the method has obvious significance and engineering practical value.
Drawings
FIG. 1 is a schematic diagram of the placement of a passive damping resistor in the prior art;
FIG. 2 is a block diagram of a virtual damping control based on filter capacitor voltage feedback in the prior art;
FIG. 3 is a block diagram of a virtual damping control based on filter capacitor current feedback in the prior art;
FIG. 4 is a schematic diagram of an active damping circuit based on high-frequency component feedback of network-side inductor current in the present application;
fig. 5 is an equivalent control block diagram of active damping based on high-frequency component feedback of network-side inductive current in the present application;
FIG. 6 shows the difference K in the present applicationgA quasi-resonant controller bode plot of the parameters;
FIG. 7 shows the difference ω in the present applicationcA quasi-resonant controller bode plot of the parameters;
fig. 8 is a control block diagram of the feedback control based on the network side inductor current quasi-resonance in the present application;
FIG. 9a is an equivalent parallel impedance circuit diagram of the high frequency component feedback active damping of the grid side inductor current in the present application;
FIG. 9b is an equivalent schematic diagram of the virtual impedance of the present application;
FIG. 10a is a waveform of the module output current without active damping in the present application;
FIG. 10b is a graph of a module grid current FFT analysis result when there is no active damping in the present application;
FIG. 11a is a waveform of the output current of the module after active damping is added in the present application; and
fig. 11b is a graph of a module grid current FFT analysis result after active damping is added in the present application.
Detailed Description
The following further describes embodiments of the present invention with reference to the drawings. It should be noted that the description of the embodiments is provided to help understanding of the present invention, but the present invention is not limited thereto. In addition, the technical features involved in the embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.
As a first embodiment of the present invention, an active damping method based on the current feedback on the network side of the LCL active filter is proposed, which is based on the circuit shown in fig. 4, and includes the following steps:
step one, acquiring a network side voltage signal and a network side current signal of an LCL active filter in each sampling period;
step two, introducing a transfer function of a feedback channel according to the signal acquired in the step one, and then calculating the component gain of the system near the resonant frequency through Clark coordinate transformation;
the feedback channel is shown in fig. 5, and in the second step, the component gain of the system around the resonant frequency is calculated as specifically shown in formula (1):
Figure BDA0003057681150000031
in the second step, when calculating the component gain of the system near the resonant frequency, the tuning controller is also adjusted to extract the high frequency component without affecting the harmonic gain of the low frequency band.
When the real part is negative, negative feedback control is realized, and the system can reliably and stably operate and along with GLCL(jωr)Gv(jωr) The peak value of the system at the resonance frequency will be reduced, and the better the resonance suppression effect. In order to ensure stability in the whole frequency range and weaken the influence of the feedback on low-frequency-band components, the damping of the resonance peak can be realized only by extracting the component gain of the system near the resonance frequency and introducing a phase-shifting link. Therefore, the high-frequency component feedback of the network side inductive current is introduced, so that the influence on the stability of the low frequency is weakened, and the resonance peak at the resonance frequency is well inhibited.
Specifically, the following steps are performed when the alignment resonance controller is adjusted:
step 2.1, selecting omega according to the requirement of cut-off frequency bandwidthc；
Step 2.1, according to the resonance peak valueGain requirement to design KgTherefore, the steady-state performance and the anti-interference capability of the system can be optimized.
And step three, bringing component gain near the resonance frequency obtained in the step two into a phase-shifting link, and realizing damping control on the resonance peak.
Considering that the gain of the traditional resonance controller at the non-fundamental wave integer frequency multiplication part is very small, when the frequency of a power grid deviates from a rated value, the controller cannot effectively track harmonic waves, the bandwidth of the quasi-resonance controller is large, the adaptability to frequency fluctuation is good, and the sufficient resonance gain can be ensured by adjusting parameters, so that the quasi-resonance controller is selected as a feedback function. It should be noted that a proportional element cannot be introduced into the feedback function to form a quasi-PR controller, because the introduction of the proportional element also has a gain effect on the middle and low frequency bands, which results in poor accuracy of low-frequency harmonic compensation. Therefore, the parameters of the quasi-resonant controller are calculated by the following formula (2):
Figure BDA0003057681150000041
wherein, KgThe resonant gain, omega, of a quasi-resonant controllercFor the resonance bandwidth, ωgIs the resonance bandwidth of the controlled object.
From the above formula (2), when s ═ j ωgWhen the resonant gain reaches a maximum value, ωgAccording to the characteristic determination of the actual controlled object, the quasi-resonance controller has 2 parameters to be designed: kgAnd ωc. In order to analyze the influence of each parameter on the control performance, one of the parameters is assumed to be unchanged, and then the influence of the other parameter on the control performance of the system is analyzed.
Wherein when ω iscWhen 1, and analysis KgWhen is changed, is different from KgA bode plot of the quasi-resonant controller transfer function under the parameters is shown in fig. 6. It can be seen that the gain of the resonance peak follows KgIncreased by increased parameters, however, the controller bandwidth is not limited by KgThe parameter impact.
When K isg10, andanalysis of omegacWhen the variation of (2) is changed, the Bode plot is shown in FIG. 7. From the graph, ω iscNot only does it affect the resonant peak gain of the controller, but it also affects the bandwidth at the cut-off frequency. With omegacThe resonant gain and bandwidth are increased. Substitution of S ═ JW into formula (2)
Figure BDA0003057681150000042
When in use
Figure BDA0003057681150000043
And calculating the difference between the two frequencies to obtain the bandwidth at the resonance frequency. Order to
Figure BDA0003057681150000044
From this the bandwidth of the quasi-resonant controller can be calculated as ωc/πHz。
The general steps for the design of a quasi-resonant controller thus obtained are: 1. selecting omega according to cut-off frequency bandwidth requirementc(ii) a 2. Designing K according to the requirement of resonance peak gaingTherefore, the steady-state performance and the anti-interference capability of the system can be optimized. Through the calculation, in order to optimize the steady-state performance and the anti-interference capability of the system, the resonance bandwidth omega in the formula (2) is obtainedc5tad/s, resonant gain K g100, i.e. the optimally selected quasi-resonant controller parameter is the resonance bandwidth ωc5rad/s, resonant gain Kg＝100。
The characteristic of a control object after the quasi-resonance control feedback is introduced is completely superposed with the curve before the quasi-resonance control feedback is introduced in the middle and low frequency band, the tracking compensation of the current inner loop on low-frequency harmonics is not influenced, and therefore OdB and the zero phase shift characteristic can be achieved in the low frequency band of the whole controller closed-loop system even if the original current inner loop design is maintained. And can provide an inverted gain at the resonant frequency for suppressing the resonant peak. In comparison, when the feedback function is the quasi-resonant controller, the resonance peak can be effectively suppressed, and the non-static tracking of the low-frequency harmonic current signal can not be influenced.
Therefore, the control block diagram based on the network-side inductor current quasi-resonant control feedback provided by the present application is finally expressed as shown in fig. 8, in which the feedback quantity of gv(s) in the diagram is moved forward to the output terminal of 1/sCf, and the feedback point is moved backward to the output terminal of gv(s), so that an equivalent active damping model can be obtained as shown in fig. 9 a. That is, a virtual impedance is connected in parallel to the filter capacitor, and the virtual impedance is calculated by the following equation (3) as shown in fig. 9 b:
Figure BDA0003057681150000051
wherein Z isvIs a virtual impedance, L1Is a filter inductor at the inverter side,/2For the grid-side filter inductance, GvIs a current regulator, KpwmThe gain of a bridge arm of the grid-connected inverter is obtained.
By adopting the technical scheme, the network side inductive current is sampled and is taken as a feedback quantity, and the active damping component is extracted and controlled, so that the function of resonance suppression is realized, the hardware cost can be saved, the problem of system reliability reduction possibly caused by a sensor and a peripheral circuit is avoided, the active damping function can be realized on the premise of not influencing the system performance, and the method has obvious significance and engineering practical value.
In order to verify the correctness and the effectiveness of the novel active damping method based on the high-frequency component feedback of the network side inductive current, the method is applied to a built APF parallel operation platform for experimental verification, and the parameters of the adopted quasi-resonant controller are as follows: omegac＝5rad/s，K g100. In order to simulate actual resonance, a resonance frequency component is added to each of the three-phase command currents as follows: i.e. ia_r＝50sinωgt，ib_r＝50sin(ωgt-2π/3)，ic_r＝50sin(ωgt +2 pi/3). Because the FFT analysis function of Wavestar software can only support 51 harmonics, the actual LCL resonant frequency omegagAt about 70 th harmonic frequency, the FFT analysis was performed by importing the waveform file into Matlab/Simulink.
Fig. 10a is a waveform of a module output current without active damping, and fig. 10b is a FFT analysis result of a module grid current without active damping. It can be seen from the figure that the resonance current component is amplified seriously before the active damping is not added, the output current waveform of the module has high-frequency oscillation, the current waveform of the power grid is seriously distorted, the THD is as high as 12.98%, and the component content near the resonance frequency is very high as can be seen from the spectrogram. Fig. 11a is a waveform of a module output current after active damping is added, and fig. 11b is a result of FFT analysis of a module grid current after active damping is added. Therefore, after the active damping is put into, the resonance peak value is well suppressed, the high-frequency burrs in the output current waveform basically disappear, the grid current THD is also remarkably reduced to 4.56%, the frequency spectrogram can show that the resonance frequency component is greatly attenuated, and meanwhile, the low-frequency harmonic content has no great difference compared with that before the active damping is added, which shows that the original compensation precision cannot be influenced after the active damping based on the quasi-resonance controller feedback is introduced.
The embodiments of the present invention have been described in detail with reference to the accompanying drawings, but the present invention is not limited to the described embodiments. It will be apparent to those skilled in the art that various changes, modifications, substitutions and alterations can be made in these embodiments without departing from the principles and spirit of the invention, and the scope of protection is still within the scope of the invention.

Claims (7)

1. An active damping method based on LCL active filter network side current feedback is characterized by comprising the following steps:
step one, acquiring a network side voltage signal and a network side current signal of an LCL active filter in each sampling period;
step two, introducing a transfer function according to the signal acquired in the step one, and then calculating the component gain of the system near the resonant frequency through Clark coordinate transformation;
and step three, bringing component gain near the resonance frequency obtained in the step two into a phase-shifting link, and realizing damping control on the resonance peak.
2. The active damping method based on LCL active filter network side current feedback according to claim 1, wherein in the second step, the component gain of the system around the resonance frequency is calculated specifically according to equation (1):
Figure FDA0003057681140000011
3. the active damping method based on LCL active filter network side current feedback according to claim 1, wherein in said step two, when calculating the component gain of the system near the resonant frequency, in order to extract the high frequency component without affecting the harmonic gain of the low frequency band, the adjustment is also made to the resonant controller.
4. The active damping method based on LCL active filter network side current feedback according to claim 3, characterized in that when adjusting the alignment resonance controller, the following steps are performed:
step 2.1, selecting omega according to the requirement of cut-off frequency bandwidthc；
Step 2.1, designing K according to the requirement of resonance peak value gaingTherefore, the steady-state performance and the anti-interference capability of the system can be optimized.
5. The LCL active filter grid-side current feedback-based active damping method according to claim 3, wherein the parameters of the quasi-resonant controller are calculated by the following formula (2):
Figure FDA0003057681140000012
wherein, KgThe resonant gain, omega, of a quasi-resonant controllercFor the resonance bandwidth, ωgIs the resonance bandwidth of the controlled object.
6. The LCL active filter grid-side current feedback-based active damping method of claim 5, wherein: in order to optimize the steady-state performance and the anti-interference capability of the system, the resonance bandwidth omega is taken from the above formulac5rad/s, resonant gain Kg＝100。
7. The active damping method based on LCL active filter network side current feedback according to claim 1, characterized in that: connecting a virtual impedance in parallel to the filter capacitor, the virtual impedance being calculated by the following equation (3):
Figure FDA0003057681140000013
wherein Z isvIs a virtual impedance, L1Is an inverter-side filter inductor, L2For the grid-side filter inductance, GvIs a current regulator, KpwmThe gain of a bridge arm of the grid-connected inverter is obtained.
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CN102522879A (en) * 2011-11-18 2012-06-27 东南大学 Active damping method of LCL grid-connected inverter system and circuit for realizing active damping method
CN108471124A (en) * 2018-04-24 2018-08-31 华北电力大学（保定） Photovoltaic DC-to-AC converter accesses the resonance suppressing method of alternating current-direct current mixing micro-capacitance sensor
CN109167361A (en) * 2018-10-17 2019-01-08 东北大学 A kind of new type of control method of LCL type parallel three phase Active Power Filter-APF

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