CN112701952B - PWM method and system for minimum effective value of current ripple of three-phase two-level inverter - Google Patents
PWM method and system for minimum effective value of current ripple of three-phase two-level inverter Download PDFInfo
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Abstract
According to the three-phase two-level inverter current ripple minimum effective value PWM method and system, a current ripple RMS prediction model is constructed, the relation between the current ripple RMS and PWM is established, and zero vector distribution is optimized in real time through a calculation result; finally, calculating a three-phase reference modulation signal by using the optimized zero vector distribution factor to generate an output control pulse; when the current ripple RMS of the inverter is reduced, the current THD and the switching frequency are reduced, the current quality is effectively improved, and the loss is reduced.
Description
Technical Field
The invention relates to the technical field of pulse width modulation in the power electronic technology, in particular to a PWM method and a PWM system for a minimum effective value of current ripples of a three-phase two-level inverter.
Background
With the popularization of industrial automation, the voltage source inverter is widely applied to the fields of industrial and household motors and control. The three-phase two-level voltage source inverter is most widely used due to low hardware cost, simple structure and mature control theory. The current PWM techniques for controlling inverters are mainly seven-segment SVPWM methods, five-segment SVPWM methods, and harmonic injection PWM techniques.
The simplified algorithm of the classical seven-segment SVPWM (also referred to as mean zero sequence voltage injection PWM) is most widely applied, and has the advantages of high voltage utilization rate, low harmonic distortion rate, low torque ripple and the like, but the advantages are not without space for improvement compared with SPWM. The theory does not analyze the effect of PWM on current ripple, nor does it explain the application of some degrees of freedom (e.g., vector selection, zero vector allocation, switching period and pulse position, etc.), making it a more sophisticated room. The harmonic injection PWM technique with third harmonic amplitude factor 1/4 proved to be the PWM technique that minimizes the current harmonic distortion rate, but also had the disadvantage of reduced voltage utilization (about 2%). Over-modulation occurs near the maximum linear modulation range (0.98-1), increasing the current THD. The five-segment SVPWM method significantly increases current ripple although the switching times are reduced.
The existing PWM technology only simply utilizes the degree of freedom in the modulation process, for example, zero vector action time of seven-segment SVPWM is always evenly distributed, for example, five-segment SVPWM obviously increases current ripple although the switching frequency is reduced, and for example, harmonic retention coefficient of harmonic injection PWM is unchanged. The existing PWM technology does not establish a model of PWM and current ripple and does not fully utilize the degree of freedom in the PWM process.
Disclosure of Invention
The invention provides a PWM method and a PWM system for a three-phase two-level inverter, which have the minimum effective value of current ripples, and are used for overcoming the technical defect that the existing PWM technology for controlling the inverter has larger current ripples or larger current THD.
In order to solve the technical problems, the technical scheme of the invention is as follows:
the PWM method for the minimum effective value of the current ripple of the three-phase two-level inverter comprises the following steps:
s1: carrying out scaling calculation and coordinate transformation processing on the voltage vector and the electrical angle of the modulated given quantity to obtain a three-phase expected phase voltage;
s2: calculating a zero vector distribution factor according to the three-phase expected phase voltage;
s3: calculating a three-phase modulation signal according to the zero vector distribution factor;
s4: and acquiring a switching period, and generating a final output pulse on the basis of the three-phase modulation signal.
In the above scheme, the calculation method in step S2 is based on a current ripple RMS prediction model, which establishes a given voltage vector relationship between the current ripple RMS and the PWM, and calculates an optimal zero vector allocation factor; finally, calculating a three-phase reference modulation signal by using the optimized zero vector distribution factor to generate an output control pulse; when the PWM current ripple RMS is reduced, the current THD and the switching frequency are reduced, the current quality is effectively improved, and the loss is reduced.
Wherein, the step S1 specifically includes the steps of:
s11: the input is the voltage U output by a d-q axis current controller under the motor vector control strategyd、UqAnd the electrical angle θ fed back by the encoder;
s12: using input voltage Ud、UqAnd calculating a three-phase expected phase voltage according to the electrical angle theta, specifically:
wherein, Ua、Ub、UcI.e. representing the desired phase voltages of the three phases.
Wherein, in the step S2, U is defined000Vector action time zero vector (U)000And U111) The proportion of the total acting time is k, namely a zero vector distribution factor, and k is more than or equal to 0 and less than or equal to 1; three phases of desired phase voltage Ua、Ub、UcInputting the calculated zero vector distribution factor k into a constructed zero vector distribution factor calculation module, and calculating an output zero vector distribution factor k specifically as follows:
s21: calculating the intermediate variable UmaxAnd Umin：
S22: calculating the intermediate variable T0：
T0＝2-Umax+Umin (3)
Wherein, T0The modulation degree is a variable related to the modulation degree in a direct proportion mode, a theoretical calculation result is 0 under the condition of a specific phase when the maximum linear modulation degree is stably operated, and 0 and a negative value can appear under the condition of overmodulation; to avoid computation holes with denominator zero, T is calculated0Performing amplitude limiting to ensure the stability of the amplitude limiting;
s23: if T0Making k equal to 0.5 and jumping out of the module, otherwise, executing the next step;
s24: calculating a zero vector distribution factor k, specifically:
s25: if k is less than 0, let k equal to 0; if k >1, let k equal 1.
Wherein, the step S3 specifically includes:
s31: calculating a common modulus U according to the zero vector distribution factoreThe method specifically comprises the following steps:
Ue＝k(1-Umax)+(1-k)(-1-Umin) (5)
s32: according to the common modulus UeCalculating a three-phase modulation signal, specifically:
wherein m isa、mb、mcRepresenting a three-phase modulated signal.
Wherein, the step S4 specifically includes:
by modulating the signal with three phases and the period TsComparing the isosceles triangle waves to generate the final output pulse;
or:
by switching period TsDirectly calculating the switching time t of the three-phase bridge arm switcha、tb、tc：
And generating final symmetrical output pulses on the basis of the switching time of the three-phase bridge arm switches.
The PWM system comprises a coordinate transformation module, a zero vector distribution calculation module, a modulation signal calculation module and a pulse generation module; wherein:
the coordinate transformation module performs scaling calculation and coordinate transformation processing on the input voltage vector and the input electrical angle to generate a three-phase expected phase voltage;
the zero vector distribution calculation module is based on a current ripple effective value prediction model; calculating to obtain a zero vector distribution factor by taking the minimum effective value of the current ripple as a target;
the modulation signal calculation module is used for calculating a three-phase modulation signal according to the zero vector distribution factor;
the pulse generating module is used for generating final output pulses on the basis of the three-phase modulation signals.
In the scheme, a pulse width modulation technology is formed by a coordinate transformation module, a zero vector distribution calculation module, a modulation signal calculation module and a pulse generation module. In order to reduce the current THD on the basis of the maximum voltage utilization rate and reduce the switching times of an inverter during high modulation ratio operation, a PWM system with the minimum effective value of the current ripple of a three-phase two-level inverter is provided.
Wherein, in the coordinate transformation module, the input is a voltage vector U output by a d-q axis current controller under a motor vector control strategyd、UqAnd the electrical angle θ fed back by the encoder; coordinate transformation module transforms voltage vector Ud、UqMultiplying by a modulation factor
Then, constant amplitude transformation is carried out by utilizing the back sum electrical angle theta, and the transformation from a two-phase rotating coordinate to a three-phase static coordinate system is realized; and finally, calculating three-phase expected phase voltage under a three-phase static coordinate system, and transmitting the result to the zero vector distribution calculation module.
Wherein, in the zero vector distribution calculation module, three-phase expected phase voltage U is calculateda、Ub、UcInputting the zero vector distribution factor k into a model, and calculating an output zero vector distribution factor k, wherein the zero vector distribution factor k specifically comprises the following steps:
calculating the intermediate variable UmaxAnd Umin：
Calculating the intermediate variable T0：
T0＝2-Umax+Umin
Wherein, T0The modulation degree is a variable related to the modulation degree in a direct proportion mode, a theoretical calculation result is 0 under the condition of a specific phase when the maximum linear modulation degree is stably operated, and 0 and a negative value can appear under the condition of overmodulation; to avoid computation holes with denominator zero, T is calculated0Clipping to ensure it is stable: if T0Not more than 0.001, and making k equal to 0.5; otherwise, the zero vector assignment factor k is calculated as:
to obtain maximum voltage utilization, k needs to be clipped: if k is less than 0, let k equal to 0; if k is greater than 1, making k equal to 1; and finally, transmitting the calculation result to the modulation signal calculation module.
Wherein, in the modulation signal calculation module, the common modulus U is calculated according to the zero vector distribution factoreAnd then calculating a three-phase modulation signal, specifically:
Ue＝k(1-Umax)+(1-k)(-1-Umin)
wherein, UeDenotes the common modulus, ma、mb、mcRepresenting a three-phase modulated signal.
Wherein, in the pulse generation module, the following calculation process is performed:
by modulating the signal with three phases and the period TsComparing the isosceles triangle waves to generate the final output pulse;
or:
by switching period TsDirectly calculating the switching time t of the three-phase bridge arm switcha、tb、tc：
And finally, generating the final symmetrical pulse on the basis of the switching time of the three-phase bridge arm switch.
In the scheme, the optimal zero vector distribution factor is calculated in the zero vector distribution calculation module, then the common-mode injection amount and the three-phase modulation signal are calculated, and finally the modulation signal is compared with the isosceles triangular carrier to generate the control pulse of the bridge arm. The technology is programmed by C language, simulation verification is carried out on MATLAB/simulink, and experimental verification is carried out on a TMS320F28377S chip.
Compared with the prior art, the technical scheme of the invention has the beneficial effects that:
the invention provides a PWM method and a PWM system for a current ripple minimum effective value of a three-phase two-level inverter, wherein a relation between the current ripple RMS and the PWM is established by constructing a current ripple RMS prediction model, and zero vector distribution is optimized in real time through a calculation result; finally, calculating a three-phase reference modulation signal by using the optimized zero vector distribution factor to generate an output control pulse; when the current ripple RMS of the inverter is reduced, the current THD and the switching frequency are reduced, the current quality is effectively improved, and the loss is reduced.
Drawings
FIG. 1 is a topology diagram of a three-phase two-level inverter;
FIG. 2 is a diagram of switching pulses and corresponding a-phase current ripples;
FIG. 3 is a schematic flow diagram of the process of the present invention;
fig. 4 is a schematic structural diagram of the system of the present invention.
Detailed Description
The drawings are for illustrative purposes only and are not to be construed as limiting the patent;
for the purpose of better illustrating the embodiments, certain features of the drawings may be omitted, enlarged or reduced, and do not represent the size of an actual product;
it will be understood by those skilled in the art that certain well-known structures in the drawings and descriptions thereof may be omitted.
The technical solution of the present invention is further described below with reference to the accompanying drawings and examples.
Example 1
To explain the technical solution of the present application in more detail, a current ripple RMS prediction model is explained in an actual three-phase two-level inverter system, specifically:
the topology of a three-phase two-level inverter is shown in fig. 1 below. The total number of the three-phase bridge arms is 6 semiconductor switches, each phase of bridge arm can only open up and down or open up and down, and the switching pulse corresponding to the phase is a high level (1) and a low level (0). When the upper bridge arm is switched on and the lower bridge arm is switched off, the bridge arm ends (points a, b and c) are connected with a positive pole Vdc/2 of the direct-current bus voltage, and conversely, the bridge arm ends are connected with a negative pole Vdc/2. The load end is replaced by a three-phase symmetrical voltage source, and current ripples are generated due to the instantaneous voltage difference between the two ends of the inductor.
The current ripple prediction model has two key parts: current ripple slope and voltage vector action time. The initial value and the final value of the current ripple in the switching period are both corresponding current fundamental waves, and the waveform of the current ripple is determined by the current slope and the time and sequence of each section of vector action. The three-phase switching pulse and a-phase current ripple is shown in fig. 2 below.
The current ripple slope can be obtained by a thevenin equivalent circuit under the action of a single voltage vector, and can be represented as the following table 1 after being substituted into the expected phase voltage:
TABLE 1 ripple slope table of circuit under voltage vector
On the premise that the output pulse is centered, the voltage vector U000A first section and a seventh section corresponding to the seven-section SVPWM current ripple; voltage vector U100、U010Or U001A second section and a sixth section corresponding to the current ripple; voltage vector U011、U101Or U110A third section and a fifth section corresponding to the current ripple; voltage vector U111Corresponding to the fourth segment of the current ripple.
Zero voltage vector U in the first, fourth and seventh sections of the current ripple000Sum voltage vector U111Corresponding current ripple slope k0Equal, zero vector action total time t0Comprises the following steps:
t0＝Ts*(2-Umax+Umin)/2
but U is000And U111The action time is respectively as follows:
t000＝k*t0
t111＝(1-k)*t0
in the current ripple correspondence formula (2): voltage vector U100Corresponding Umax＝UaThe case (1); voltage vector U010Corresponding Umax＝UbThe case (1); voltage vector U001Corresponding Umax＝UcThe case (1). The current ripple slope k corresponding to three conditions1The total action time is as follows:
t1＝Ts*(2Umax+Umin)/2
in the current ripple correspondence formula (2): voltage vector U011Corresponding Umin＝UaThe case (1); voltage vector U101Corresponding Umin＝UbThe case (1); voltage vector U110Corresponding Umin＝UcThe case (1). The current ripple slope k corresponding to three conditions2The total action time is as follows:
t2＝Ts*(-Umax-2Umin)/2
on the premise that the output pulse is centered and symmetrical, the duration of each section of current ripple is k × t0/2，t1/2，t2/2，(1-k)*t0，t2/2，t1(1-k) × t and/20/2. When U is turnedmax＝UaAnd U ismin＝UcThe phase a current ripple is shown in fig. 4.
The effective value of the a-phase current ripple can be obtained by combining the current ripple slope of table 1:
and further obtaining the effective value of the three-phase current ripple:
the effective value of the three-phase current ripple is easily obtained under the following conditions:
substituting the three-phase current ripple slope and the action time under different conditions to obtain a formula (4). It is noted that the current ripple RMS prediction does not need to be calculated in implementing the PWM.
In the specific implementation process, the result obtained by the scheme reduces the current ripple RMS of the inverter, simultaneously reduces the current THD and the switching frequency, effectively improves the current quality and reduces the loss.
Example 2
More specifically, on the basis of embodiment 1, as shown in fig. 3, the present invention provides a three-phase two-level inverter current ripple minimum effective value PWM method, including the following steps:
s1: carrying out scaling calculation and coordinate transformation processing on the voltage vector and the electrical angle of the modulated given quantity to obtain a three-phase expected phase voltage;
s2: calculating a zero vector distribution factor according to the three-phase expected phase voltage;
s3: calculating a three-phase modulation signal according to the zero vector distribution factor;
s4: and acquiring a switching period, and generating a final output pulse on the basis of the three-phase modulation signal.
In a specific implementation process, the calculation method in step S2 is based on a current ripple RMS prediction model, which establishes a given voltage vector relationship between the current ripple RMS and the PWM, and calculates an optimal zero vector allocation factor; finally, calculating a three-phase reference modulation signal by using the optimized zero vector distribution factor to generate an output control pulse; when the PWM current ripple RMS is reduced, the current THD and the switching frequency are reduced, the current quality is effectively improved, and the loss is reduced.
More specifically, the step S1 specifically includes the following steps:
s11: the input is the voltage U output by a d-q axis current controller under the motor vector control strategyd、UqAnd the electrical angle θ fed back by the encoder;
s12: using input voltage Ud、UqAnd calculating a three-phase expected phase voltage according to the electrical angle theta, specifically:
wherein, Ua、Ub、UcI.e. representing the desired phase voltages of the three phases.
In the above scheme, compared with the seven-segment SVPWM established under the two-phase stationary coordinate system, the invention is established under the two-phase rotating coordinate system, so that the intermediate calculation process (such as ipark transformation) can be reduced. The input quantity of the invention is the output U of a d-q axis current controller under a motor vector control strategyd、UqAnd the electrical angle theta fed back by the encoder. For voltage vector Ud、UqThe transformation range of the coordinate transformation method is-1, the constant amplitude value is carried out with the electrical angle theta after the factor 2/sqrt (3) is multiplied, and the coordinate transformation from a two-phase rotating coordinate to a three-phase static coordinate system is realized.
More specifically, wherein in said step S2, U is defined000Vector action time zero vector (U)000And U111) The proportion of the total acting time is k, namely a zero vector distribution factor, and k is more than or equal to 0 and less than or equal to 1; three phases of desired phase voltage Ua、Ub、UcInputting the calculated zero vector distribution factor k into a constructed zero vector distribution factor calculation module, and calculating an output zero vector distribution factor k specifically as follows:
defining intermediate variables UmaxAnd Umin：
Calculating the intermediate variable T0：
T0＝2-Umax+Umin (3)
Wherein, T0The modulation degree is a variable related to the modulation degree in a direct proportion mode, a theoretical calculation result is 0 under the condition of a specific phase when the maximum linear modulation degree is stably operated, and 0 and a negative value can appear under the condition of overmodulation; to avoid computation holes with denominator zero, T is calculated0Clipping to ensure it is stable: if T0Not more than 0.001, and making k equal to 0.5; otherwise, calculating the zero vector distribution factor k according to the formula (4):
to obtain maximum voltage utilization, k needs to be clipped: if k is less than 0, let k equal to 0; if k >1, let k equal 1.
More specifically, step S3 specifically includes:
s31: calculating a common modulus U according to the zero vector distribution factoreThe method specifically comprises the following steps:
Ue＝k(1-Umax)+(1-k)(-1-Umin) (5)
s32: according to the common modulus UeCalculating a three-phase modulation signal, specifically:
wherein m isa、mb、mcRepresenting a three-phase modulated signal.
More specifically, step S4 specifically includes:
by modulating the signal with three phases and the period TsComparing the isosceles triangle waves to generate the final output pulse;
or:
by switching period TsDirect-computing three-phase bridge arm switchSwitching time ta、tb、tc：
And generating final symmetrical output pulses on the basis of the switching time of the three-phase bridge arm switches.
In the specific implementation process, compared with a seven-segment SVPWM and an equivalent algorithm thereof, the invention has lower current harmonic distortion rate in a full linear modulation range, and the reduction of the current harmonic distortion rate can improve the electric energy quality and reduce the resistance loss of a system; when the maximum linear modulation degree (m is 1), the hybrid use mode of the seven-segment type and the five-segment type is adopted, so that not only is the current harmonic distortion rate reduced, but also the switching frequency is reduced, and the switching loss can be effectively reduced by reducing the switching frequency; compared with a seven-segment SVPWM established under a two-phase static coordinate system, the method is directly established under a d-q coordinate system, and the calculation amount of coordinate transformation is reduced.
Example 3
More specifically, on the basis of embodiment 2, as shown in fig. 4, a three-phase two-level inverter current ripple minimum effective value PWM system is provided, which includes a coordinate transformation module, a zero vector distribution calculation module, a modulation signal calculation module, and a pulse generation module; wherein:
the coordinate transformation module performs scaling calculation and coordinate transformation processing on the input voltage vector and the input electrical angle to generate a three-phase expected phase voltage;
the zero vector distribution calculation module is based on a current ripple effective value prediction model; calculating to obtain a zero vector distribution factor by taking the minimum effective value of the current ripple as a target;
the modulation signal calculation module is used for calculating a three-phase modulation signal according to the zero vector distribution factor;
the pulse generating module is used for generating final output pulses on the basis of the three-phase modulation signals.
In the specific implementation process, a pulse width modulation technology is formed by a coordinate transformation module, a zero vector distribution calculation module, a modulation signal calculation module and a pulse generation module. In order to reduce the current THD on the basis of the maximum voltage utilization rate and reduce the switching times of an inverter during high modulation ratio operation, a PWM system with the minimum effective value of the current ripple of a three-phase two-level inverter is provided.
More specifically, in the coordinate transformation module, the input is a voltage vector U output by a d-q axis current controller under a motor vector control strategyd、UqAnd the electrical angle θ fed back by the encoder; coordinate transformation module transforms voltage vector Ud、UqMultiplying by a modulation factor
Then, constant amplitude transformation is carried out by utilizing the back sum electrical angle theta, and the transformation from a two-phase rotating coordinate to a three-phase static coordinate system is realized; and finally, calculating three-phase expected phase voltage under a three-phase static coordinate system, and transmitting the result to the zero vector distribution calculation module.
More specifically, in the zero vector distribution calculation module, three phases of desired phase voltages U are calculateda、Ub、UcInputting the zero vector distribution factor k into a model, and calculating an output zero vector distribution factor k, wherein the zero vector distribution factor k specifically comprises the following steps:
calculating the intermediate variable UmaxAnd Umin：
Calculating the intermediate variable T0：
T0＝2-Umax+Umin
Wherein, T0The modulation degree is a variable related to the modulation degree in a direct proportion mode, a theoretical calculation result is 0 under the condition of a specific phase when the maximum linear modulation degree is stably operated, and 0 and a negative value can appear under the condition of overmodulation; to avoid computation holes with denominator zero, T is calculated0Clipping to ensure it is stable: if T0Not more than 0.001, and making k equal to 0.5; otherwise, the zero vector assignment factor k is calculated as:
to obtain maximum voltage utilization, k needs to be clipped: if k is less than 0, let k equal to 0; if k is greater than 1, making k equal to 1; and finally, transmitting the calculation result to the modulation signal calculation module.
More specifically, in the modulation signal calculation module, the common modulus U is first calculated based on the zero vector partition factoreAnd then calculating a three-phase modulation signal, specifically:
Ue＝k(1-Umax)+(1-k)(-1-Umin)
wherein, UeDenotes the common modulus, ma、mb、mcRepresenting a three-phase modulated signal.
More specifically, in the pulse generation module, the following calculation process is performed:
by modulating the signal with three phases and the period TsComparing the isosceles triangle waves to generate the final output pulse;
or:
by switching period TsDirectly calculating the switching time t of the three-phase bridge arm switcha、tb、tc：
And finally, generating the final symmetrical output pulse on the basis of the switching time of the three-phase bridge arm switch.
In the specific implementation process, the optimal zero vector distribution factor is calculated in the zero vector distribution calculation module, then the common-mode injection amount and the three-phase modulation signal are calculated, and finally the modulation signal is compared with an isosceles triangle carrier to generate the control pulse of the bridge arm. The technology is programmed by C language, simulation verification is carried out on MATLAB/simulink, and experimental verification is carried out on a TMS320F28377S chip.
It should be understood that the above-described embodiments of the present invention are merely examples for clearly illustrating the present invention, and are not intended to limit the embodiments of the present invention. Other variations and modifications will be apparent to persons skilled in the art in light of the above description. And are neither required nor exhaustive of all embodiments. Any modification, equivalent replacement, and improvement made within the spirit and principle of the present invention should be included in the protection scope of the claims of the present invention.
Claims (6)
1. The PWM method for the minimum effective value of the current ripple of the three-phase two-level inverter is characterized by comprising the following steps of:
s1: carrying out scaling calculation and coordinate transformation processing on the voltage vector and the electrical angle of the modulated given quantity to obtain a three-phase expected phase voltage;
s2: calculating a zero vector distribution factor according to the three-phase expected phase voltage;
s3: calculating a three-phase modulation signal according to the zero vector distribution factor;
s4: acquiring a switching period, and generating a final output pulse on the basis of the three-phase modulation signal;
the step S1 specifically includes the steps of:
s11: the input is the voltage U output by a d-q axis current controller under the motor vector control strategyd、UqAnd the electrical angle θ fed back by the encoder;
s12: using input voltage Ud、UqAnd calculating a three-phase expected phase voltage according to the electrical angle theta, specifically:
wherein, Ua、Ub、UcI.e. to indicate three-phase expectationA phase voltage;
in said step S2, U is defined000The proportion of the vector action time to the total zero vector action time is k, namely a zero vector distribution factor; three phases of desired phase voltage Ua、Ub、UcInputting the constructed zero vector distribution factor calculation module, and calculating the output k, specifically:
s21: calculating the intermediate variable UmaxAnd Umin：
S22: calculating the intermediate variable T0：
T0＝2-Umax+Umin (3)
Wherein, T0The modulation degree is a variable related to the modulation degree in a direct proportion mode, a theoretical calculation result is 0 under the condition of a specific phase when the maximum linear modulation degree is stably operated, and 0 and a negative value can appear under the condition of overmodulation; to avoid computation holes with denominator zero, T is calculated0Performing amplitude limiting to ensure the stability of the amplitude limiting;
s23: if T0Making k equal to 0.5 and jumping out of the module, otherwise, executing the next step;
s24: calculating a zero vector distribution factor k, specifically:
s25: if k is less than 0, let k equal to 0; if k >1, let k equal 1.
2. The three-phase two-level inverter current ripple minimum effective value PWM method according to claim 1, wherein the step S3 specifically comprises:
s31: calculating a common modulus U according to the zero vector distribution factoreThe method specifically comprises the following steps:
Ue＝k(1-Umax)+(1-k)(-1-Umin) (5)
s32: according to the common modulus UeCalculating a three-phase modulation signal, specifically:
wherein m isa、mb、mcI.e. representing a three-phase modulated signal.
3. The three-phase two-level inverter current ripple minimum effective value PWM method according to claim 2, wherein the step S4 specifically comprises:
by modulating the signal with three phases and the period TsComparing the isosceles triangle waves to generate the final output pulse;
or:
by switching period TsDirectly calculating the switching time t of the three-phase bridge arm switcha、tb、tc：
And generating the final symmetrical pulse on the basis of the switching time of the three-phase bridge arm switch.
4. The PWM system is characterized by comprising a coordinate transformation module, a zero vector distribution calculation module, a modulation signal calculation module and a pulse generation module; wherein:
the coordinate transformation module performs scaling calculation and coordinate transformation processing on the input voltage vector and the input electrical angle to generate a three-phase expected phase voltage;
the zero vector distribution calculation module is based on a current ripple effective value prediction model; calculating to obtain a zero vector distribution factor by taking the minimum effective value of the current ripple as a target;
the modulation signal calculation module is used for calculating a three-phase modulation signal according to the zero vector distribution factor;
the pulse generation module is used for generating final output pulses on the basis of the three-phase modulation signals;
wherein, in the coordinate transformation module, the input is the voltage U output by a d-q axis current controller under a motor vector control strategyd、UqAnd the electrical angle θ fed back by the encoder; coordinate transformation module transforms voltage vector Ud、UqMultiplying by a modulation factor
Then, constant amplitude transformation is carried out by utilizing the back sum electrical angle theta, and the transformation from a two-phase rotating coordinate to a three-phase static coordinate system is realized; finally, calculating three-phase expected phase voltage under a three-phase static coordinate system, and transmitting the result to the zero vector distribution calculation module;
in the zero vector distribution calculation module, three-phase expected phase voltage U is calculateda、Ub、UcInputting the zero vector distribution factor k into a model, and calculating an output zero vector distribution factor k, wherein the zero vector distribution factor k specifically comprises the following steps:
calculating the intermediate variable UmaxAnd Umin：
Calculating the intermediate variable T0：
T0＝2-Umax+Umin
Wherein, T0The modulation degree is a variable related to the modulation degree in a direct proportion mode, a theoretical calculation result is 0 under the condition of a specific phase when the maximum linear modulation degree is stably operated, and 0 and a negative value can appear under the condition of overmodulation; to avoid computation holes with denominator zero, T is calculated0Clipping to ensure it is stable: if T0Not more than 0.001, and making k equal to 0.5; otherwise, the zero vector assignment factor k is calculated as:
to obtain maximum voltage utilization, k needs to be clipped: if k is less than 0, let k equal to 0; if k is greater than 1, making k equal to 1; and finally, transmitting the calculation result to the modulation signal calculation module.
5. The three-phase two-level inverter current ripple minimum effective value PWM system according to claim 4, wherein in the modulation signal calculation module, the common modulus U is first calculated according to a zero vector distribution factoreAnd then calculating a three-phase modulation signal, specifically:
Ue＝k(1-Umax)+(1-k)(-1-Umin)
wherein, UeDenotes the common modulus, ma、mb、mcRepresenting a three-phase modulated signal.
6. The three-phase two-level inverter current ripple minimum effective value PWM system according to claim 5, characterized in that in the pulse generation module, the following calculation process is performed:
by modulating the signal with three phases and the period TsComparing the isosceles triangle waves to generate the final output pulse;
or:
by switching period TsDirectly calculating the switching time t of the three-phase bridge arm switcha、tb、tc：
And finally, generating the final symmetrical pulse on the basis of the switching time of the three-phase bridge arm switch.
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