CN101854120B - High-efficiency multifunctional flyback converter - Google Patents

Abstract

The invention relates to a lossless resonant absorption circuit and integrates the operating principle of the lossless resonant absorption circuit into the operation of the flyback power converter to form a high-performance high-efficiency converter structure. In the operation of the converter, the absorption circuit can absorb leakage inductance of a transformer for energy storage at no loss to effectively suppress the over-voltage peak of a power switch. Moreover, energy stored by an absorption capacitor and the inductance of the transformer are utilized to form resonance to create zero voltage condition for the power switch to realize zero voltage switching operation. A unique self-driven synchronizing rectifier circuit in a secondary loop can further improve the efficiency of the converter.

Description

A kind of high-efficiency multifunctional flyback converter

Background introduction

Field that the present invention belongs to

This patent is the invention belonging in electrical power conversion field, more specifically says a kind of flyback power transformer configuration and mode of operation thereof of uniqueness.This kind of converter can realize high efficiency DC-DC voltage transitions or the adjustment of single stage type power factor adds complex function of DC-DC voltage transformation etc.For electric power conversion apparatus provides a kind of low cost, high performance design concept.

The description of association area

Along with the demand people of day by day urgent environmental protection require to use green energy resource in every field more and more widely.This certainly will require power consumption equipment and device for converting electric energy further to raise the efficiency and use few material of trying one's best in electric energy use field.Another imperative requirement is to exchanging the raising of the power factor of power device, to reduce the loss of electric energy.In our daily life, numerous semiconductor electronic devices all needs to have one AC energy is converted to the electric power conversion apparatus of direct current energy so that obtain the required direct current energy of operation from electric main network.In this case, if can design and manufacture a high efficiency, low cost, low material consumption also has the AC-DC supply convertor of power factor regulation, and it is apparent that the environment and economy widely that it brings people is worth.

Traditional A.C.-D.C. converter adopts two kinds of general ways conventionally in the market.A kind of is first with diode rectifier bridge, alternating voltage to be converted to pulsating direct current, with capacitor filtering pulsation ripple, then use DC-to-DC (referred to as DC-DC) converter to become needed magnitude of voltage to export and provide direct current output and the secured electrical insulation exchanging between input filtered DC voltage conversion.The typical circuit structure of this way as shown in Figure 1.In Fig. 1, rectifier bridge BG1 converts pulsating direct current to again by capacitor C1 filtering pulsation ripple alternating voltage.Electronic power switch Q1, transformer TX1, secondary commutation diode D2 and secondary filter capacitor C2 form a basic inverse-excitation type DC-DC translation circuit.R3 and C3 be a typical absorbing circuit in order to absorb the energy storage of primary leakage inductance in switching process, thereby suppress the switching voltage spike that it causes on electronic switch.Here it may be noted that except inverse-excitation type translation circuit to also have other circuit structure as positive activation type, push-pull type, half-bridge circuit, full-bridge circuits etc. all can complete DC-DC mapping function.Reverse excitation circuit is used element minimum conventionally, and cost is also minimum.Sort circuit structure is generally used for lower-powered occasion and power factor is also lower.

When electric power is larger, generally all need to add power factor regulation (abbreviation is referred to as PFC) circuit to reach higher power factor at input.Typically with the A.C.-D.C. converter of power factor regulation circuit as shown in Figure 2.Compare with Fig. 1 circuit, electronic switch Q11, inductance L 11, diode D11 and capacitor C 11 have formed power factor regulation circuit.The function of power factor regulation circuit be mainly the switching manipulation by electronic switch Q11 make the envelope of the current waveform of inductance L 11 maintain sinusoidal waveform after full-wave rectification and with rectification after pulsation sinusoidal voltage homophase.Like this from rectifier bridge BG1 ac input end AC1, the alternating current that AC2 inputs naturally also maintain sinusoidal waveform and with input voltage homophase, thereby can make power factor reach the perfect condition close to 1.Electronic power switch in Fig. 1 and Fig. 2 is the most conventional with MOSFET.The switching device of other type is as also alternative in bipolar transistor or IGBT etc.

Circuit working efficiency shown in Fig. 1 and Fig. 2 is generally all lower.Electronic power switch Q1 shown in figure, Q11 is all operated in hard switching operating state.Particularly when they are converted to conducting state from turn-offing due to drain electrode and source electrode between potential difference from high pressure, suddenly change to close to zero, between source-drain electrode, the high-voltage energy storage of parasitic capacitance is in extremely short moment by the forced discharge of switch own, and its energy all consumes and is converted into heat energy in switching tube inside.Both lowered efficiency like this, increased again pipe heating, also produced stronger electromagnetic radiation simultaneously.Another loss factor is the leakage inductance energy of transformer.When Q1 conducting, electromagnetic energy is along with the increase of transformer TX1 armature winding PN1 electric current is set up gradually.The electromagnetic energy being stored in when Q1 turn-offs in coupling inductance is coupled to secondary winding SN2 by magnetic circuit, and rectifier diode D2 conducting is powered to output.The energy being stored in leakage inductance have to be maintained circulation and by charging, energy be transferred to source drain capacitance and get on by electric capacity between the source-drain electrode of Q1 owing to cannot being coupled to secondary going.The drain electrode of Q1 electricity, for may be pushed away very highly, even causes Q1 over-voltage breakdown in this case.In order to suppress this voltage overshoot phenomenon, people conventionally have to adopt absorbing circuit to absorb and consume this part leakage inductance energy.R3 in Fig. 1 and Fig. 2, C3 network is a kind of absorbing circuit.Also has in actual applications multiple different absorbing circuits design.These circuit are generally as everyone knows, therefore will not describe in detail here.In addition, the loss of rectifier diode D2 is also very important.Particularly when being output as low-voltage, high-current, the conduction voltage drop of D2 can account for output voltage 5% to 10% or even higher, thereby be directed at proportional power loss therewith.Due to the imperfect reverse recovery characteristic of diode, its caused switching loss is also considerable on the other hand.

Summary of the present invention

Object of the present invention is exactly the weak point that will overcome above-mentioned traditional translation circuit, the converter scheme that provides a kind of high-performance, low-power consumption, circuit to try one's best simple and with low cost.The present invention adopts a kind of loss-free active absorbing resonant circuit to suppress switch overshoot voltage.And utilize LC resonance principle make power switch work in soft switch or approach soft on off state.The present invention also adopts this high efficiency translation circuit to realize the single stage type AC-DC voltage conversion function with power factor regulation.In conjunction with the minimizing of advantage and the power transfer link of lossless active absorption circuit, the multi-functional conversion circuit of this single-stage can be realized by lower product cost and reach higher conversion efficiency.Meanwhile, invention has also adopted a kind of simple circuit of synchronous rectification to substitute diode rectifier, thereby further improves the efficiency of doing of transducer.

Accompanying drawing explanation

Fig. 1 is depicted as a typical classical inverse excitation type converter circuit structure.Fig. 2 is depicted as a traditional AC-DC transformer configuration with power factor regulation.Fig. 3 has described a kind of high efficiency converter circuit structure and principle thereof with lossless active absorbing resonant circuit of the present invention.Fig. 4 has described another kind of the present invention with high efficiency change device circuit structure and the principle thereof of lossless active absorbing resonant circuit.Fig. 5 has described a kind of synchronous commutating control circuit of the present invention.Fig. 6 has described another kind of synchronous commutating control circuit of the present invention.

The detailed description of invention

As previously mentioned, the energy storage of traditional inverse excitation type converter leakage inductance and the deflection type absorbing circuit adopting for the due to voltage spikes that suppresses to cause in its switching process, and the switching loss of power switch pipe is loss factor main in the anti exciting converter course of work.Key concept of the present invention is to utilize the resonance characteristic of controlled inductance, electric capacity to form a kind of loss-free absorbing circuit.This circuit can can't harm on the one hand and effectively when power switch pipe turn-offs, absorb the energy of transformer leakage inductance and suppress switching voltage spike.Can before power switch pipe conducting, utilize energy storage and the resonance between transformation inductance of absorbing circuit to carry out to produce for power switch the condition of no-voltage, thereby realize the soft switching manipulation of no-voltage type simultaneously.One of its concept circuit is as shown in Fig. 3 (A).In Fig. 3 (A), N-type metal-oxide-semiconductor field effect t (being designated hereinafter simply as MOSFET) Q1 is master power switch.The armature winding end of oppisite phase (need to illustrate that end of oppisite phase means the one end of not being with phase point mark herein, the called after relative concept of anti-phase and homophase, elementary end of oppisite phase and secondary end of oppisite phase are homophase, vice versa) of transformer TX1 is received in the drain electrode of Q1.The source electrode of Q1 is received the negative terminal of the direct voltage after rectifier bridge BG1 rectification by current sense resistor R1, be also direct current hold PGND.The armature winding in-phase end of TX1 is connected to the direct current anode VDC+ after BG1 rectification.The series loop being comprised of capacitor C3 and N-type MOSFET Q3 that is connected across TX1 armature winding two ends is lossless absorbing circuit.Square type square CU1 representative in Fig. 3 is controlled and drive circuit.This control drive circuit has two drive output signal DR1 and DR2.DR1 is used for driving Q1, and DR2 is used for driving Q3.CU1 accepts two feedback input signals simultaneously.Input signal ISN1 accepts the Q1 source current signal being detected by R1.Input signal VS1 is the voltage VQ1D of the detected Q1 drain electrode in the voltage detecting networking that is comprised of C4, R2, R4 Q1D.The circuit that C4, R2, R4 form is the differential detection network with dividing potential drop effect.The voltage change ratio of the VQ1D transmitting by C4 feeds back to CU1 after by R2, R4 dividing potential drop and controls drive circuit CU1.The needed direct current output of the last generation of the current rectifying and wave filtering circuit Vo that the secondary output of transformer TX1 is comprised of D2 and C2.

The groundwork signal waveform of Fig. 3 (A) circuit is as shown in Fig. 3 (B).When Q1 is when to starts conducting, the electric current in primary winding PN1 is by being set up gradually to the flow cycle of PGND through PN1, Q1 and R1 by VDC+ in the course of the work.When Q1 is when t1 turn-offs, the energy that is stored in PN1 coupling inductance part is coupled to the secondary winding SN2 of TX1 and is made D2 conducting by magnetic circuit, to C2 and be connected on Vo and GND between load supplying.And be stored in energy in PN1 leakage inductance owing to cannot being coupled to secondary going, keeping the next meeting of the rule source-drain electrode parasitic capacitance charging to MOSFET Q1 naturally of its current continuity, while also charges to C3 by the parasitic diode of Q3, and causes the voltage on these electric capacity to continue to rise until leakage inductance electric current decays to zero constantly at t2.At this moment be stored in that electromagnetic energy in leakage inductance is all transferred to the source-drain electrode parasitic capacitance of Q1 and capacitor C 3 gets on and make the voltage on these electric capacity reach its peak value.Voltage on C3 be upper negative under just, if Q3 conducting during this period of time, C3 charging completes by Q3.The size of the crest voltage on electric capacity and the size of leakage inductance energy and electric capacity is directly relevant.Under identical leakage inductance energy, capacitance is larger, and crest voltage when leakage inductance energy shifts end on electric capacity is less.Because what adopt here is lossless absorbing circuit, therefore C3 can use larger capacitance, thereby can effectively reduce its voltage peak, thereby the drain voltage VQ1D that also makes Q1 by Pliers in lower level.Simultaneously because the capacitance of C3 and Q1 source-drain electrode parasitic capacitance is proportional with the energy that they absorb.C3 is larger, and the energy proportion that it absorbs is also larger, and the energy that Q1 parasitic capacitance absorbs is less.The control signal waveform of Q3 is as shown in Fig. 3 (B).As shown in the figure, from Q1, at t1, turn-off Q3 during this period of time that C3 voltage reaches peak value at t2 in conducting state, the process that leakage inductance energy is transferred to C3 completes by Q3.So both reduced the caused loss of Q3 parasitic diode conduction voltage drop, and the more important thing is and avoided the caused vibration of parasitic diode reverse recovery characteristic.When C3 voltage reaches peak value, Q3 turn-offs, and its parasitic diode is in reverse blocking state, and the path between C3 and PN1 is cut off, and C3 voltage is next a period of time maintains its peak value.

T1 to the energy that is stored in primary winding coupled inductance during t3 transfer to constantly secondary by D2 to C2 and load supplying.In t3 this part energy storage constantly, exhaust, secondary current interrupts, and the source of Q1 is leaked parasitic capacitance and started to the electric discharge of primary winding, and the voltage of Q1 drain electrode Q1D starts to fall after rise.At this moment Q3 starts again conducting.The energy being stored in C3 starts to shift and form resonance to the armature winding PN1 of transformer.1/4 cycle of first of resonance is that t3 to t4 is interval, and the energy of C3 shifts to PN1.T4 constantly C3 voltage drop be zero and the electric current of PN1 reaches maximum.Second 1/4 cycle of resonance, since moment t4, charges to C3 when the electric current of PN1 starts, and the voltage on C3 is along with charging is progressively set up, and polarity is upper just lower negative.Along with the rising of C3 voltage, the current potential of Q1D correspondingly reduces gradually.If the energy of resonance is enough large, the voltage of C3 can reach with VDC+ and equate before t5 or t5, the at this moment parasitic diode conducting of Q1, and the drain potential of Q1 is close to zero, for condition has been created in the zero voltage switch operation of Q1.At t5 constantly, Q3 turn-offs, and Q1 is conducting subsequently, thereby has completed the soft switching manipulation of Q1, and starts the operating process of repetition next cycle.

Here it should be noted that not to be can create by resonance the zero voltage condition of Q1 in all cases.When resonant energy hour, the voltage of C3 does not reach the value of VDC+ when second 1/4 end cycle, thereby the drain voltage of Q1 does not reach zero yet.In this case, the switching over of Q1 and Q3 point is selected in the minimum point of VQ1D voltage, is also resonance valley point.Although do not reach like this zero voltage switch operation, switching loss still can be significantly less than traditional mode of operation.In addition, in classical inverse exciting converter design, how the leakage inductance of transformer is done to such an extent that to try one's best little be a very challenging problem, transformer leakage inductance is larger, and switching loss is larger.The in the situation that of this programme, the major part of the leakage inductance energy of transformer does not consume, but by and Absorption Capacitance C3 between resonance be that power switch Q1 creates no-voltage or minimum voltage Switching Condition.Therefore, not harsh like that to the requirement of transformer leakage inductance, manufacturing and designing of transformer is easier comparatively speaking.

Here be noted that for the switch of Q3 simultaneously and control, the drain voltage of monitoring Q1 with the Sampling network that C4, R2, R4 form is a conceptual example.The voltage detecting circuit of other kind also can be used for reaching same object.Except detecting the drain voltage of Q1, by the source current signal of monitoring Q3, also can be with the switching manipulation of helping control Q3.Because the source electrode of Q3 floats, it is more convenient that its current signal detects the general Current Mutual Inductance transformer that uses.In order to keep illustrative simplicity understandable, in Fig. 3 (A), do not draw this part circuit.When utilizing voltage signal that VSN1 obtains to control, the switching over point of Q3 is selected in the zero crossing of voltage change ratio.Because the zero crossing of voltage change ratio is also the zero crossing of electric current.So the switching over point of Q3 is selected in the zero crossing of electric current during by electric current signal controlling.Output for converter at Vo end, both can do voltage-type, also can do current mode output and control.In the time need to controlling output voltage, Vo end output voltage is fed back to and controls the switching manipulation that drive circuit CU1 controls Q1.In the time need to controlling output current, the current signal of Vo or GND end is fed back to and controls the switching manipulation that drive circuit CU1 controls Q1.The implementation method of above-mentioned these concepts is dealer to be familiar with, therefore do not mark in Fig. 3 (A).In addition, in Fig. 3 (A) circuit, switching device Q1, Q3 used also can combine to replace with ambipolar NPN transistor and anti-paralleled diode.Its port corresponding relation is as shown in Fig. 3 (C).

Circuit shown in Fig. 3 (A) both can have been realized does not have the AC-DC of PFC function mapping function, can realize the single stage type AC-DC converter with PFC function yet.When not needing PFC function, the capacitance of C1 can be selected greatlyr, approaches pure direct current like this through the filtered VDC+ of C1.When needs PFC function, C1 capacitance is selected smallerly, and the HF switch that its capacitance needs only enough filtering Q1 operates the ripple producing, and to exchanging the low frequency sinuous pulsation voltage of input after BG1 rectification, substantially there is no filter action.This VDC+ keeps the sinuous pulsation waveform of simple alternating current after full-wave rectification substantially.At this moment the current envelope curve of Q1 and PN1 controls that to take the sinuous pulsation waveform of VDC+ be reference signal, finally by the switch of Q1 control make from the electric current of ac input end AC1 and AC2 input be sinusoidal waveform and and input voltage homophase reach power factor close to 1 effect.The electric current of noting transformer secondary output in this case also with the low frequency sinuous pulsation waveform of the elementary the same VDC+ of following, so C2 need to carry out with the electric capacity of larger capacity these low frequency pulsating compositions of filtering.If the ripple coefficient of output voltage is required strictly, to it is also conceivable that after C2 and add one-level LC filter circuit again.Here require emphasis, if there is no the lossless resonance absorbing circuit in Fig. 3 (A), the switching loss being caused by the leakage inductance of transformer TX1 will lower the efficiency of the AC-DC converter of this single stage type PFC comprehensive function widely.Employing due to lossless resonance absorbing circuit in Fig. 3 (A), the energy storage of transformer leakage inductance in switching process not only do not consume, be utilized on the contrary the switching loss of attenuating power switch, so whole efficiency can be widely higher than the traditional two-stage type translation circuit shown in Fig. 2, thereby provide a kind of efficiency high, use material is few, the green AC-to DC converter scheme that product cost is low.

Fig. 4 (A) is depicted as the converter circuit that adopts another kind of lossless resonance absorbing circuit.To in figure and identical circuit part shown in Fig. 3 (A) repeat no more here.The place different with Fig. 3 (A) is mainly that C3 and Q3 have become and are connected across between Q1 drain electrode and elementary direct current power ground PGND from being connected across PN1 two ends.Q3 has also changed P type MOSFET into by N-type MOSFET, and between Q3 source electrode and PGND, has been connected in series a current sense resistor R3, the current signal of Q3 is fed back to the feedback input end ISN2 that controls drive circuit CU1.Described in [0012] joint, the current feedback signal of Q3 also can be with the switching manipulation of helping control Q3, and the current signal on R3 be take PGND as reference point in Fig. 4 (A), so feedback circuit is very simple and convenient.By the reason of P type MOSFET, be that and the polarity of N-type MOSFET parasitic diode makes, in Fig. 4 (A) circuit, C3 cannot be controlled to the discharge process of PN1 because the polarity of its parasitic diode and N-type MOSFET is contrary.Q3 is not so good as to come conveniently with N-type MOSFET with P type MOSFET in general, but take PGND as reference point because of its source electrode, and gate electrode drive signals does not need to float, and drive circuit is comparatively simple.

In Fig. 4 (A), the switch control law of Q1 and Q3 is the same with circuit in Fig. 3 (A).Its main signal waveform is as shown in Fig. 4 (B).In the interval Q1 conducting of moment to t1, primary winding current under the effect of VDC+, flow through PN1, Q1, R1 linear growth.The conducting pulse duration of to t1 is determined by the PWM modulation circuit that requires process to control drive circuit CU1 to the adjusting of Vo end output voltage or electric current.In t1 Q1 shutoff constantly, Q3 conducting is by its parasitic diode bypass.The leakage inductance energy of PN1 is charged until this part energy is all transferred to C3 to C3.T2 constantly C3 voltage reach peak value, and the electric current that flows through Q3 drops to zero point.At this moment Q3 turn-offs, and C3 voltage maintains peak state.In the meantime, the energy being stored in PN1 coupling inductance continues to be coupled to secondary to C2 and Vo end load supplying.When this part energy is when t3 exhausts constantly, transformer secondary output winding current decays to zero.The voltage of just going up negative voltage and Q1 drain electrode Q1D under armature winding PN1 two ends starts to fall after rise.At this moment Q3 again conducting make the inductance of C3 and PN1 carry out energy exchange by resonance.Interval at first 1/4 cycle of resonance t3～t4, the stored energy of C3 progressively shifts the inductive current into PN1, until VQ1D equals VDC+ at t4, at this moment the electric current of PN1 reaches maximum.At the inductive current of second 1/4 cycle t4～t5 PN1 of resonance through VDC+, C1, R3 and Q3 to C3 reverse charging.Inductance energy in moment t5 PN1 exhausts, and Q3 electric current drops to zero, and Q1 drain voltage VQ1D drops to the lowest point.If resonant energy is enough large, C3 voltage and Q1 drain voltage can be pushed to zero, for the zero voltage switch operation of Q1 creates conditions.When Q1 drain voltage is pushed to zero or the lowest point, Q3 turn-offs, and Q1 gate electrode drive signals makes Q1 conducting, thereby has realized no-voltage or minimum voltage switching manipulation.

Described at [0012] joint, because the resonance frequency of circuit is determined by L, C and the R value of participating in the element of resonance substantially, so inductance parameters when transformer, the capacitance of C3, after the source leakage parasitic capacitance value of Q1 and the resistive impedance in resonant tank are determined, its resonance frequency is also substantially fixing.Based on this electricity, for the Q3 ON time from t1 to t2 and the Q3 ON time from t3 to t5, also can carry out approximate set time control according to the resonant frequency characteristic of circuit.That is to say, VQ1D starts to rise to from t1 the time that t2 reaches peak value constantly constantly, and from t3, constantly start to drop to time that t5 arrives valley point constantly and all can be similar to by the resonance frequency of circuit and determine and substantially constant, so Q3 also can set according to this time in the ON time of t1 to t2 and t3 to t5.In this situation, control drive circuit and only need to rather than by the drain voltage VQ1D of Real-Time Monitoring Q1 or the source current of Q3, control the ON time of Q3 by timing circuit.Timing circuit starts with the gate pole Continuity signal of Q3 at t1 and t3 constantly simultaneously, at t2 and t5, constantly reaches the time of setting and turn-offs Q3.Here should be noted that the inductance composition that participates in resonance at the resonant process in t1 to t2 interval is the leakage inductance of TX1 primary coil PN1, and the self-inductance that is PN1 at the interval inductance composition that participates in resonance of t3 to t5, so these two interval resonance frequencys are different.

Same as Fig. 3 (A), in Fig. 4 (A), do not draw from output end vo to the feedback signal loop of controlling drive circuit CU1 yet.In concrete application, if need regulating and controlling to as if output voltage, get Vo voltage signal as feedback.If need regulating and controlling to as if output current, get the output current signal of Vo end or GND end as feedback.This class feedback circuit is familiar with by this area personage, therefore repeat no more here.Equally also as Fig. 3 (A) circuit, circuit shown in Fig. 4 (A) both can have been realized does not have the DC-DC of PFC function or AC-DC converter, also can control and make the envelope of primary winding current follow the pulsation sinusoidal waveform of input voltage after full-wave rectification by the switching manipulation of Q1, thereby realize the AC-DC converter of single stage type band PFC function.P type MOSFET Q3 in Fig. 4 (A) also can combine to replace with ambipolar PNP transistor and anti-paralleled diode in addition.As shown in Fig. 4 (C).

Except use lossless resonance absorbing circuit in primary loop, the present invention can also replace conventional diode rectification further to raise the efficiency with circuit of synchronous rectification in secondary loop.The converter of 5V output of take is example, and while conventionally adopting Schottky diode rectification, its forward voltage drop is about 0.3V to 0.5V left and right, and 5V exports and compare, and the caused loss of diode forward pressure drop accounts for 6% to 10% of power output.Synchronous rectification replaces rectifier diode with MOSFET conventionally, and its conduction voltage drop can be controlled at 0.1V between 0.15V, is also that efficiency can improve 4% to 7% left and right.The more existing implementation methods of synchronous rectification at present.Wherein typical way is to add an assistive drive winding at transformer secondary output, utilizes the voltage of assistive drive winding to drive synchronous rectifier.This way control circuit is relatively simple, but transformer is necessary for each synchronous rectifier, adds a winding.This can increase widely complexity and the manufacture difficulty of transformer when multichannel is exported.Another kind of way is to utilize an assistive drive transformer or photoelectric coupled device, and synchronous rectification control signal is coupled to secondary and is equipped with corresponding drive circuit and drives synchronous rectifier from primary control drive circuit.This way cost can be higher, and number of elements used is also many.

Circuit of synchronous rectification of the present invention as shown in Figure 5.The generation of synchronous rectification control signal neither needs assistive drive winding, does not also need to use auxiliary transformer or photoelectric device from just grade coupled, but according to the voltage of secondary loop, curent change, directly processes generation with a unique circuit.As shown in Fig. 5 (A), rectifier diode D2 is replaced by N-type MOSFET Q2, and by output plus terminal, has been moved on to the loop on output ground.This transposition is mainly for the source electrode of Q2 is connected with GND, the relatively good processing of gate electrode drive signals like this.Q5 and Q6 are PNP transistor, a differential circuit of composition, and it is common current source resistance that R9 is connected between the emitter of Q5, Q6 and Vo.The collector electrode output of Q5 removes to drive the gate pole of Q2 through resistance R 11 and R12 dividing potential drop.By R5 and R6, dividing potential drop between Vo and the drain electrode of Q2 drives the base stage of Q5.And Q6 base stage by R7 and R8 at Vo and GND, be also that between the source electrode of Q2, dividing potential drop drives.The parameter selection of R5, R6, R7 and R8 makes the voltage ratio of R7 slightly be greater than the voltage ratio of R5.Voltage when R5, R6 loop and R7, R8 loop both end voltage equate on R7 is greater than voltage on R5 slightly, but at R5, R6 loop both end voltage during higher than about tens millivolts of R7, R8 loop, the pressure drop of R5 is higher than the voltage drop of R7.Like this when the source-drain electrode voltage of Q2 reduces to zero, the preferential conducting of Q6, Q5 cut-off, Q2 is also in off state.In circuit working, when Q1 closes, have no progeny, the voltage of transformer secondary output winding is upper just lower negative.This voltage need to rise to the forward conduction voltage drop that exceeds a Q2 parasitic diode than C2 both end voltage (being also output voltage) just can be so that secondary current forms flow cycle.The voltage that minute hydraulic circuit that R5, R6 form is in this case accepted is higher than the voltage of the acceptance in R7, R8 loop.And the voltage ratio of above-mentioned two bleeder circuits design makes voltage drop on R5 in this case higher than the voltage drop of R7, thereby make Q5 conducting, Q6 cut-off, the gate pole of Q2 obtains positive driving voltage by Q2 from Vo like this, Q2 conducting, secondary winding electric current by Q2 to C2 and load supplying.When secondary current exhausts, the conduction voltage drop of Q2 is zero, now Q6 conducting, and Q5 cut-off, Q2 turn-offs, and prevents that the voltage of C2 from discharging by secondary winding SN2 and Q2.When Q1 is during in next switching manipulation cycle conducting, the voltage of winding SN2 becomes lower just upper negative, and Q5 deepens to end, Q6 is still in conducting state, and Q2 still ends, and continues to stop C2 electric discharge, until Q1 turn-offs, circuit repeats the synchronous rectification operation that foregoing process is carried out another cycle.

Operating principle based on same, circuit of synchronous rectification of the present invention also can be transformed into another kind of structure as shown in Fig. 5 (B).Compare with Fig. 5 (A), Fig. 5 (B) is essentially its dual circuit.Q2 becomes P type metal-oxide-semiconductor from N-type metal-oxide-semiconductor, and changes to the positive end loop of output, and Q5, Q6 also become NPN transistor from PNP transistor.Its operation principle is the same with Fig. 5 (A), therefore repeat no more here.What need prompting is that Q5, Q6 also can adopt small-signal MOSFET or other active electronic device to realize same function.If the differential circuit being comprised of Q5, Q6 here drives gain not, can between Q5 collector electrode and Q2 gate pole, insert one-level in-phase amplification circuit increases driving force again.Q4 shown in Fig. 6 (A) and figure (B) is an example.Another kind of possible way is through one-level inverting amplifier, to drive the gate pole of Q2 by the output of Q6 collector electrode again.Apparent because of its reason, also repeat no more here.

Here also it may be noted that above description and relevant indicators are mainly to set forth as an example principle of the present invention and concept simultaneously.In practical application, following same concept can realize by different forms.Therefore being applied in the situation of its basic conception of this patent is not limited to implementation method described herein.Same right described in ensuing rights statement, also for from principle concept, can realize with different circuit without prejudice to its basic principle in the situation that.

Claims (8)

1. a converter circuit, it is characterized in that: a rectifier bridge is direct current exchanging input rectifying, an electric capacity is connected across two DC output end mouths of this rectifier bridge, a flyback transformer, the first port of its armature winding is received the positive voltage output end of rectifier bridge, the second port of armature winding is received the positive voltage port of a main electronic switch, the negative voltage port of this main electronic switch is received the negative voltage output of rectifier bridge by a current measuring element, a lossless absorbing circuit is formed by an Absorption Capacitance and an auxiliary electron switch serial connection, the positive voltage port of one end of Absorption Capacitance and auxiliary electron switch is in series, the other end of Absorption Capacitance is connected with the positive voltage output end of rectifier bridge, the negative voltage port of auxiliary electron switch and the positive voltage port of main electronic switch are connected, one by a Detection capacitance and two detection resistance voltage detecting networks in series, this voltage detecting network is connected across between the positive voltage port of main electronic switch and the negative voltage output of rectifier bridge, one end of Detection capacitance is received the positive voltage port of main electronic switch, the other end of Detection capacitance is connected with two detection resistance potential-divider networks in series, the intermediate node of these two detection resistance is received a feedback input end of a control drive circuit, this is controlled drive circuit and accepts from above-mentioned voltage detecting network, the electric current of main electronic switch current measuring element and secondary output end or voltage signal are feedback, and generation drives signal to remove to drive above-mentioned main electronic switch and auxiliary electron switch, a rectifier diode, its anode is received the secondary winding of flyback transformer and the anti-phase port of the first port of armature winding, the negative electrode of this rectifier diode is received the anode of an output filter capacitor and as the positive output end of converter, the other end of the negative terminal of this output filter capacitor and the secondary winding of flyback transformer is connected and as the negative output terminal of converter, due to voltage spikes when this converter circuit utilizes lossless absorbing circuit to absorb main electronic switch shutoff in operating process, and to utilize the formed resonance of inductance of this lossless absorbing circuit and flyback transformer be the operating characteristic that main electronic switch is realized zero voltage switch or minimum voltage switch, when main electronic switch turn-offs, auxiliary electron switch conduction, then when reaching maximum, main electronic switch both end voltage turn-offs, make the leakage inductance energy of flyback transformer be stored on Absorption Capacitance losslessly, when the coupling electromagnetic energy of flyback transformer exhausts, the conducting utilize Absorption Capacitance and the resonance characteristic of flyback transformer primary inductance declines main electronic switch both end voltage again of auxiliary electron switch, when reaching zero or minimum point, main electronic switch both end voltage turn-offs auxiliary electron switch, and make main electronic switch conducting, thereby realize no-voltage or the operation of minimum voltage switch conduction.

2. converter circuit according to claim 1, it is characterized in that: this lossless absorbing circuit being composed in series by Absorption Capacitance and auxiliary electron switch is connected across between the positive voltage port and rectifier bridge negative voltage output port of main electronic switch, one end of Absorption Capacitance is received the positive voltage port of main electronic switch, the negative voltage port of the other end of Absorption Capacitance and auxiliary electron switch is connected, and the positive voltage port of auxiliary electron switch is received the negative voltage output port of rectifier bridge.

3. converter circuit according to claim 1, it is characterized in that: in the operating process of circuit, utilize the monitoring rate of change of main electronic switch voltage or the electric current of auxiliary electron switch to control the operation of auxiliary electron switch, when closing, main electronic switch has no progeny, make auxiliary electron switch conduction, voltage at main electronic switch rises to peak value, its voltage change ratio is for the first time during zero passage, or during the current over-zero of auxiliary electron switch, turn-off auxiliary electron switch, before main electronic switch conducting, the conducting again when the couple current of flyback transformer decays to zero of auxiliary electron switch, resonance by Absorption Capacitance and flyback transformer primary inductance makes the voltage of main electronic switch start to decline, when the voltage drop of main electronic switch is to minimum, its voltage change ratio is again during zero passage, or when the voltage of main electronic switch is zero, or during the current over-zero of auxiliary electron switch, turn-off auxiliary electron switch, then make main electronic switch conducting.

4. converter circuit according to claim 2, it is characterized in that: in the operating process of circuit, utilize the monitoring rate of change of main electronic switch voltage or the electric current of auxiliary electron switch to control the operation of auxiliary electron switch, when closing, main electronic switch has no progeny, make auxiliary electron switch conduction, voltage at main electronic switch rises to peak value, its voltage change ratio is for the first time during zero passage, or during the current over-zero of auxiliary electron switch, turn-off auxiliary electron switch, before main electronic switch conducting, the conducting again when the couple current of flyback transformer decays to zero of auxiliary electron switch, resonance by Absorption Capacitance and flyback transformer primary inductance makes the voltage of main electronic switch start to decline, when the voltage drop of main electronic switch is to minimum, its voltage change ratio is again during zero passage, or when the voltage of main electronic switch is zero, or during the current over-zero of auxiliary electron switch, turn-off auxiliary electron switch, then make main electronic switch conducting.

5. converter circuit according to claim 1, it is characterized in that: the ON time of auxiliary electron switch is carried out to timing controlled according to the natural resonance cycle of circuit, at main electronic switch, close and have no progeny, make auxiliary electron switch conduction, and after 1/4th harmonic periods of circuit state, auxiliary electron switch is turn-offed maintaining conducting this moment, when the couple current in flyback transformer winding decays to zero, make the conducting again of auxiliary electron switch, and after 1/2nd harmonic periods of circuit state, auxiliary electron switch is turn-offed maintaining conducting this moment, then make main electronic switch conducting.

6. converter circuit according to claim 2, it is characterized in that: the ON time of auxiliary electron switch is carried out to timing controlled according to the natural resonance cycle of circuit, at main electronic switch, close and have no progeny, make auxiliary electron switch conduction, and after 1/4th harmonic periods of circuit state, auxiliary electron switch is turn-offed maintaining conducting this moment, when the couple current in flyback transformer winding decays to zero, make the conducting again of auxiliary electron switch, and after 1/2nd harmonic periods of circuit state, auxiliary electron switch is turn-offed maintaining conducting this moment, then make main electronic switch conducting.

7. converter circuit according to claim 1, it is characterized in that: keeping the voltage after rectification circuit rectification is all-wave sinuous pulsation waveform, and the operation by main electronic switch controls and make the input current envelope of ac input end follow the sinusoidal waveform with input supply voltage homophase, reach with this converter circuit and realize the high efficiency complex transformation function with power factor regulation and DC-dc conversion.

8. converter circuit according to claim 2, it is characterized in that: keeping the voltage after rectification circuit rectification is all-wave sinuous pulsation waveform, and the operation by main electronic switch controls and make the input current envelope of ac input end follow the sinusoidal waveform with input supply voltage homophase, reach with this converter circuit and realize the high efficiency complex transformation function with power factor regulation and DC-dc conversion.