CH712011B1 - Converter for electrically isolated transmission of electrical energy. - Google Patents

Abstract

A converter according to the invention for the electrically isolated transmission of electrical energy between a primary system and a secondary system has: an uncontrolled resonant primary isolation stage (1), which is connected for energy exchange between the primary system and at least one intermediate circuit, and a secondary converter converter stage (2), which for energy exchange between the at least one intermediate circuit and the secondary system (2). The potential separation stage (1) is set up to transform one or more primary voltages specified by the primary system into an, in particular lower, voltage level of the at least one intermediate circuit and thus to generate an intermediate circuit voltage in the form of a sine voltage or an absolute value voltage at the at least one intermediate circuit , and • the regulating converter stage (2) is set up to convert an intermediate circuit voltage into a secondary voltage, the secondary voltage being a direct voltage or sinusoidal alternating voltage. This can be ensured by switching frequency and duty cycle variation smooth switching of the transistors of the converter modules (11) of the electrical isolation stage (1), and a symmetrical distribution of the primary voltage between the converter modules (11) can be achieved automatically.

Description

For low-network potential isolated supply of low voltage consumers from the medium-voltage AC network, e.g. the single-phase AC rail network, multi-cell converter circuits are proposed in the literature, see the dissertation by M. Steiner mentioned below, where each converter cell has a rectifier stage on the input side with a step-up converter function for sinusoidal control of the input current and for generating a regulated, well above the peak value of the input AC voltage of the converter cell lying DC link voltage and at the output of each converter cell a regulated or unregulated DC / DC converter is arranged with a medium or high frequency transformer for electrical isolation and a rough adaptation of the intermediate circuit voltage to the output voltage level serving number ratio. If a DC voltage consumer is to be fed (in the sense of a simple explanation, only this case is considered in more detail below in connection with the prior art), the output voltage of the DC / DC converter in the case of a regulated converter can be directly applied to the via a corresponding duty cycle of the power transistors required load voltage is set or the load is arranged directly at the output of the DC / DC converter. In order to be able to control the high voltage of the medium-voltage network with power semiconductors with comparatively low blocking voltage resistance, the cells are generally on the input side. connected in series and on the output side in parallel (on the input side it is then advantageous to arrange only one boost converter inductance for all cells). This advantageously achieves a distribution of the mains voltage between the input stages and the load current, which is typically relatively high due to the lower output voltage, between the output stages of the converter cells. Furthermore, the series parallel connection, in addition to the number of turns ratio, takes a further step in the direction of adapting the mains and load voltage levels.

Such systems, however, have a relatively high complexity of measurement and control. In addition to the output voltage and the input current of the overall system, i.A. the intermediate circuit voltage and the output current of each cell are measured and regulated in order to ensure a symmetrical distribution of the power flow between the individual cells and to define the voltage load on the medium-voltage side semiconductors and the current load on the low-voltage side semiconductors. Furthermore, the timing of the rectifier stages and output stages advantageously has the same frequency and is out of phase, i.e. coordinated to take place in order to achieve a minimal ripple of the input current and the output voltage of the overall system.

A simplification of the concept outlined above is only possible if there is no direct control of the output voltage of the overall system. Unregulated series resonance DC / DC converters operated at the resonance point are then used as output stages of the cells, which ideally have a fixed (load-independent) voltage transformation ratio equal to the number of turns of the transformer. The output voltage of the converters, which is the same due to the parallel connection of the converter cells on the output side, is then mapped into corresponding intermediate circuit voltages on the medium voltage side, which are the same for all converter cells, so that no regulation of the intermediate circuit voltages is required and the balancing of the power flow of the cells can be achieved simply by equal control of the rectifier input stages of the cells . The level of the output voltage of the overall system results from the balance of power exited on the network side and output on the load side, i.e. for a given load resistance, the load voltage is set such that there is a load-side power consumption in the amount of the mains input power; this immediately shows the possibility of regulating the load voltage. Such a system is e.g. described in the mentioned dissertation by M. Steiner.

However, to transmit the amplitude of the input current (the line-side power consumption) determining the output signal of the output voltage regulator to the input current control, communication between the primary and secondary side is still to be provided, with a high insulation voltage being guaranteed here; Furthermore, the input current control must control the rectifier stages of the cells with sinusoidal, phase-shifted pulse width modulation, which is made more difficult by the line frequency and switching frequency potential differences occurring between the cells. In addition, the high number of power components on the medium-voltage side and the relatively high volume of the medium-voltage-side parts of the converter cells (there is an intermediate circuit capacitor with a capacity in each converter cell such that, despite the input-side power flow pulsating at twice the mains frequency, an approximately constant intermediate circuit voltage results) disadvantageous for the practical implementation of such a system, since in connection with the insulation distances to be maintained a high overall volume results.

Another general disadvantage is the increase in the input voltage by the step-up function of the rectifier stage; the sum of the intermediate circuit voltages of the cells is therefore typically 20% above the amplitude of the mains voltage, which means that a higher number of cells (or generally a higher total reverse voltage of the series-active power semiconductors) is required to control a given voltage of the medium-voltage network.

An alternative converter concept which is advantageous with regard to voltage loading and complexity can be achieved by interchanging the partial functions of the system described above, i.e. By arranging an unregulated potential isolation stage on the input side and a subsequent secondary or low-voltage control converter stage, which on the one hand draws a current from the potential isolation stage in such a way that a sinusoidal input current results on the network side and on the other hand keeps the system output voltage at a constant value regardless of the load. All control functions and a large part of the modules of the power section are thus advantageously arranged on the low-voltage side, or the size of the medium-voltage side part is minimized, since only the primary side of the electrical isolation stage remains there.

Such a system was presented in the basic form in the publication by H. Weiss mentioned below. The input AC voltage (mains voltage) is rectified there by a diode bridge on the input side without smoothing. The input stage of the potential-isolating converter generates a switching-frequency symmetrical square-wave AC voltage with full duty cycle (no regulation) and an amplitude that varies according to the absolute value voltage, which is applied to the primary winding of a transformer, from the resulting absolute value voltage (hereinafter referred to as «| AC | voltage») is and occurs according to the number of turns ratio scaled occurs on the secondary winding. On the secondary side, the voltage pulses are generated by an output stage, i.e. another bridge rectifier without smoothing, rectified and thus converted back into a continuous absolute value voltage. The potential-isolating converter therefore actually has the function of an | AC | / | AC | converter; In the following, this designation is therefore used for the potential-isolating converter (but not previously used in the literature). A | AC | / | | AC | converter now extracts a | AC | current which is in phase with the | AC | voltage and which is due to a control converter stage, which is designed as an | AC | / DC step-up converter stage the missing smoothing elements via the transformer of the | AC | / | AC | converter directly to the input of the input-side diode bridge and ultimately results in an advantageously voltage-proportional profile of the mains current. On the other hand, the | AC | / DC step-up converter delivers power to the output of the system and thus keeps the DC voltage of the output capacitor at a constant value, i.e. the mean power flow to the output is always set by selecting the amplitude of the | AC | current in such a way that the power requirement of the load is exactly covered at a desired voltage.

In the below-mentioned publication by P. Dràbek et al. The concept described above is expanded with the aim of using it at medium voltage to a primary-side series connection of several converter stages, all primary windings being arranged on a common magnetic core which also carries the secondary winding common to all primary windings. A direct matrix converter circuit is used for the primary side of the potential isolation stages, which converts the line-frequency AC voltage on the input side without prior rectification into an | AC | voltage into a switching-frequency square-wave AC voltage with an amplitude that varies according to an | AC | curve. (With reference to this implementation, the implementation described above (literature (Weiss)) can be seen as an indirect matrix converter, for which a separation of the rectifier function (conversion of the AC voltage into a | AC | voltage) and the subsequent conversion of the | AC Voltage into a switching-frequency square-wave AC voltage.) On the secondary side, the amplitude-modulated square-wave AC voltage is led to the input of an active full-bridge rectifier circuit (formed from two bridge branches with transistors and to these antiparallel diodes), which integrates the function of the rectification and the step-up converter and, on the one hand, the secondary winding a voltage proportional, ie takes switching frequency amplitude-modulated square wave current and on the other hand generates a defined DC voltage supported by an output capacitor at its output for supplying the load. Alternatively, as described in the publication by B. Hafez et al. described, again the functions are separated into a rectifier and a step-up converter stage, the circuit structure and signal waveforms being the same on the secondary side compared to the publication by H. Weiss (see above; the potential isolation stage is then AC / | AC | - and the Step-up converter stage to be referred to as | AC | / DC converter).

Regardless of the specific implementation (according to Dràbek or Hafez et al.), The arrangement of the primary windings on the same iron core is ideal (assuming ideal coupling) for all windings the same induced voltage and thus advantageously a symmetrical distribution of the mains voltage to the primary converter stages ensured. However, this symmetry becomes real due to the non-ideal coupling of the primary windings and also due to the leakage inductances present between the primary windings and the secondary winding due to the high insulation requirement, which generally depends on the geometric arrangement of the windings. have different values. Another disadvantage is the lack of full modularity of the system due to the common magnetic core, which complicates the structural design and has significant disadvantages in terms of redundancy, since a fault in a converter stage due to the magnetic coupling of the windings of all stages directly impairs the function of the overall system leads or a simple exchange of a module is not possible compared to a fully modular system. In addition, the suitability of the concept for high switching frequencies is also in question, since no measures are described to minimize the switching losses of the power semiconductors of the potential isolation stage.

State of the art literature

M. Steiner, "Series-connected DC link converter in traction applications on AC contact wire", PhD dissertation, ETH Zurich, Switzerland, 2000.

H. Weiss, "Elimination of the 162/3 Hz 15 kV main transformer on electrical traction vehicles", in Proc. 1st Europ. Power Electronics and Applications Conf. (EPE), Brussels, Belgium, Nov. 1985.

P. Dràbek, Z. Peroutka, M. Pittermann, and M. Cédl, "New configuration of traction converter with medium-frequency transformer using matrix converters", IEEE Trans. Ind. Electron., Vol. 58, no. 11, pp. 5041-5048, Nov. 2011.

B. Hafez, H.S. Krishnamoorthy, P. Enjeti, S. Ahmed, and I. J. Pitel, "Medium voltage power distribution architecture with medium frequency isolation transformer for data centers", in Proc. 29th Annu. IEEE Applied Power Electronics Conf. and Expo. (APEC), Fort Worth, TX, USA, Mar. 2014.

The object of the present invention is therefore to create a floating converter system of reduced complexity. This can have one or preferably several of the following properties: extensive modularity, medium-voltage input and low complexity of the power section on the medium-voltage side, i.e. a small or minimal number of power semiconductors and filter elements, small or minimal number of converter cells, automatic symmetrization of the distribution of the input voltage to the inputs of the converter cells, curve of the input or mains current proportional to the mains voltage, i.e. ohmic network behavior, regulated output voltage of the entire system and a smooth switching of all converter cells of the potential isolation stage.

In order to ensure broad applicability, the basic concept of the overall system is intended both for the conversion of a single-phase and a three-phase AC mains voltage into a floating, regulated DC output voltage of the overall system (this also includes the case frequency = 0, ie the Conversion of a DC mains voltage into a potential-separated regulated DC load voltage) or a single-phase AC mains voltage as well as a three-phase AC mains voltage into a potential-separated regulated single-phase AC or three-phase AC load voltage.

[0016] The object is achieved by a converter according to the patent claims.

In particular, there is a converter for electrically isolated transmission of electrical energy between a primary system and a secondary system, comprising a primary-side potential isolation stage, which is connected for energy exchange between the primary system and the at least one intermediate circuit, and a secondary-side control converter stage, which for energy exchange between the at least one intermediate circuit and the secondary system are connected, <tb> • <SEP> where the potential isolation stage is set up to transform one or more primary voltages specified by the primary system into a, in particular lower, voltage level of the at least one intermediate circuit and thus an intermediate circuit voltage in the form of a sinusoidal voltage or a Generate absolute value voltage, and <tb> • <SEP> the regulating converter stage is set up to convert an intermediate circuit voltage into a secondary voltage, the secondary voltage being a direct voltage sinusoidal alternating voltage.

This can be ensured by switching frequency and duty cycle variation, a smooth switching of the transistors of the converter modules of the electrical isolation stage, and a symmetrical distribution of the primary voltage can be achieved automatically on the converter modules.

In general, the potential isolation stage can be implemented by converter modules connected in series on the primary side in the form of series resonance converters with potential separation operated at the resonance point, which isolate the AC voltage present on the input side via a ballast or filter inductance - the filter inductance is used to filter switching frequency processes negligible impedance at mains frequency, the mains voltage can therefore be seen as being directly connected to the series connection of the converter modules - with a transformation ratio defined directly by the number of turns ratio (load-independent) into a smooth | AC | secondary voltage or smooth AC secondary voltage, i.e. transform have the function of a floating AC / | AC | or AC / AC converter. The outputs of the converter modules, which are advantageously phase-shifted to minimize the high-frequency spectral components of the resulting mains current caused by the clocking, are then directly connected in parallel in the simplest case (see FIG. 1 (a)) and the low-voltage | AC | - or AC- DC link by means of a control converter stage on average over a switching period proportional to the | AC | or AC DC link voltage, ie Magnitude sine current (| AC | current) or sinusoidal current (AC current), taken, the amplitude of the current being selected such that the power requirement of a load connected to the output of the regulating converter stage is exactly covered or a defined one at the load DC or AC voltage occurs. Alternatively, as shown in Fig. 1 (b), the control converter stage can also be arranged separately for each converter module, which means that an AC or | AC | intermediate circuit voltage is present at the output of each converter module, which current flows through the downstream control converter stage is taken in such a way that each control converter stage makes an equal contribution to covering the power requirement of the load or to impressing the load voltage on the one hand, and on the other hand ensures a stable and equal distribution of the mains voltage to the inputs of the converter modules of the potential isolation stage, which advantageously achieves both can be that the control converter stages emulate a replacement input resistor that is the same for all converter modules.

If the electrical isolation stage has at least two converter modules and these are clocked out of phase, it is not necessary to equip the converter modules with individual filter elements, that is to say each associated with a converter module. Such filter elements, which are no longer necessary, are capacitors and inductors for smoothing switching-frequency (i.e. high-frequency) currents to the primary system.

The series-resonant electrical isolation stage can be operated unregulated and advantageously with a high duty cycle, which means that ideal symmetrical square-wave voltages with the line voltage divided by the number of modules or rectified line voltage occur as envelopes on the primary transformer winding. In principle, no control-technical coupling of the potential isolation stage and the subsequent regulating converter stage is advantageously required. In the simplest case, the electrical isolation stage works autonomously, which results in a low level of control complexity and high robustness of the overall system (self-sufficiency of the sub-functions). In order to minimize the current load on the low-voltage side DC link capacitance (Fig. 1 (a)) or DC link capacitance (Fig. 1 (b)), however, synchronization and phase-shifted timing of the converter modules of the potential isolation stage or the individual control converter stages (Fig. 1 ( b)) advantageous with respect to each other and also the electrical isolation stage compared to the subsequent control converter stage or control converter stages. In the circuit according to FIG. 1 (b), the current load of the capacitor supporting the output voltage of the overall system or the high-frequency ripple of the output voltage occurring as a result of the clocking is also minimized.

In general, it should be pointed out that the concept according to the invention, when the potential isolation stage and control converter stage are designed appropriately, both for a power transfer from the medium-voltage to the low-voltage as well as for the inverse direction of power flow, i.e. a transfer of power from the low voltage to the medium voltage side is suitable. In the case of an AC / DC converter function of the overall system, both rectifier and inverter operation are possible.

Finally, with a view to a practical implementation, it should be emphasized as a further advantage of the concept that a buffer capacitor (buffering the load voltage) of significant capacitance can only be arranged at the output of the overall system, i.e. In contrast to a conventional version at the converter module level, there is no need to buffer the power flow in an intermediate circuit on the primary side (see section on the prior art). The buffer capacitor is thus also directly available for buffering load jumps or a pulsed load current profile, which results in a corresponding current load on the output capacitance (which, for example, is only available for electrolytic capacitors for correspondingly high capacitance values). If there is a smooth load current curve and a supply of a three-phase medium-voltage network system explained below, the output capacitor can theoretically even be omitted entirely or can be implemented practically with only a small capacitance value due to the then constant instantaneous power curve.

With regard to the operating behavior, the operation of the series resonance converter of the converter modules of the potential isolation stage at the resonance point represents a main advantage of the system according to the invention. The resonant operation compensates for the leakage inductance of the transformers, which means that there is ideally no series impedance between the primary and secondary side, i.e. there is a very direct coupling of both sides with regard to current and voltage translation or dynamics of the power transfer. For an embodiment according to FIG. 1 (a), the input voltages of the potential isolation stages are all directly coupled to the same secondary voltage and thus ideally have the same value (same number of turns ratio of the transformers of all modules) in the amount of the input voltage divided by the number of modules, it is This means that there is ideal symmetrization with regard to the input voltage distribution to the inputs of the converter modules. The same also applies to the concept according to FIG. 1 (b), assuming the above-mentioned emulation of the same input resistance at the input of the control converter stages, since these resistors are then translated to the primary side and again ensure a symmetrical distribution of the input voltage. Furthermore, the current or power flow is distributed symmetrically to the modules, since the input current flows through all converter modules in series and is directly reflected in the module's secondary currents, which form the same power with the same voltages for the converter modules.

Furthermore, due to the series capacities of the series resonance converter, a parasitic, e.g. DC component of the transformer primary or secondary voltage caused by different switching times of the converter switching stages is absorbed and thus reliably prevents saturation of the transformer. The same applies mutatis mutandis to one embodiment, described below (cf. FIG. 4), of executing one or both sides of the series resonance converter in a half-bridge circuit, with two resonance capacitors, each connected as a capacitive voltage divider, forming the second bridge branch.

Finally, operation at resonance frequency offers the advantage of a primary or secondary current curve composed of half-sine oscillations, with which only the relatively small transformer magnetization current can be switched actively. De-energized or soft switching can then be ensured relatively simply by switching just before the point at which the half-sine wave reaches zero again (or returns to the magnetizing current). As an alternative, a transition to discontinuous operation is also possible, where a waiting time with the primary voltage still present is inserted after each sine current half-oscillation. (Note: For rectifier operation, when the primary voltage is applied, the diodes of the output stage of a converter module prevent the direction of the secondary current from being reversed).

The converter can be realized with only a single, common diode bridge for all converter modules (11) of the electrical isolation stage. This single diode bridge can be implemented as an active rectifier that is only clocked at the mains frequency, e.g. consisting of high-voltage SiC-FETs as switches.

In the following, the subject matter of the invention is explained in more detail with the aid of preferred exemplary embodiments which are illustrated in the accompanying drawings. These show: <tb> Fig. 1: <SEP> embodiments of a converter for unidirectional or bidirectional single-phase AC / DC conversion; (a) Series-resonant modular AC / | AC | potential isolating stage 1 with input-side series connection and output-side parallel connection of converter modules 11 and output-side | AC | / DC control converter stage 2 with a common control converter 21; (b) as (a), but with a separate control converter 21 for each converter module 11. <tb> Fig. 2: <SEP> embodiment of a floating single-phase AC / DC converter system with regulated DC output voltage; In the interest of clarity, only 2 AC / | AC | converter modules 11 are shown for the potential isolation stage, the outputs of which are connected in parallel as shown in FIG. 1 (a) and with a subsequent | AC | / DC control converter stage 2 with a common step-up converter are connected. <tb> Fig. 3: <SEP> Characteristic voltage and current profiles of the system according to FIG. 2. <tb> Fig. 4: <SEP> Detailed structure of an embodiment for a single-phase AC / AC conversion. In the sense of a clear representation, only 2 modules are shown for the potential isolation stage (each module has the function of a high-frequency potential-isolated series resonance AC / AC matrix converter working at the resonance point). Furthermore, for the sake of clarity, an embodiment according to Fig. 1 (a), i.e. selected with only one AC / AC control converter 21 (step-up converter) in AC / AC control converter stage 2. <tb> Fig. 5: <SEP> Example for the emulation of a conventional three-phase transformer using the example of a Scott transformer; (a) switching the conventional Scott transformer; (b) embodiment as a circuit with series connection of the first and second output voltage, i.e. of the outputs of the RX- | AC | / DC and ST- | AC | / DC control converter stage. <tb> Fig. 6: <SEP> Example of a three-phase AC / AC conversion with phase branches of the potential isolation stages connected on the primary side in a triangle and secondary side in an asterisk and a downstream three-phase AC / AC boost converter (three-phase control converter) for regulating the output voltage. <tb> Fig. 7: <SEP> Characteristic voltage and current profiles of a module of the potential isolation stage for partially hard switching (a), soft, i.e. de-energized switching (b) and (c) and low-loss soft switching (d); (e): Envelope of the pulse-shaped symmetrical pulse-frequency square-wave voltage v_T, mv, e and the secondary current pulses i_r, lv, e applied to the transformer with a magnitude sine curve corresponding to the network-frequency sinusoidal voltage applied to the network.

A possible circuit implementation of the concept for a floating single-phase AC / DC conversion is with reference to Fig. 1 (a), i.e. with a rule converter 21 common to all converter modules, shown in FIG. 2. The associated characteristic voltage and current profiles are given in FIG. 3, the magnetization current mentioned above not being shown explicitly for the sake of clarity. A series connection of the converter modules 11 forms the potential isolation stage 1, which can be connected via a network-side filter inductance LF to a network which supplies a network voltage vg and a network current ig.

Each converter module 11 has on the input side a switching stage in the form of an indirect matrix converter 111, which is formed by an input diode bridge without smoothing on the output side and a subsequent transistor-side transistor full-bridge circuit, the output of which is connected to the transformer primary winding via a primary-side series resonance capacitor. The output voltage of the input diode bridge, which is to be seen as the primary DC link voltage, shows the course of the rectified AC input voltage of the converter module and is converted by the transistor full bridge circuit into a symmetrical square wave AC voltage vT, mv, which has the primary side | AC | DC link voltage as an envelope (see Fig. 3). The switching-frequency pulsating input current in the transistor full-bridge circuit (which is equal to the output current of the input diode bridge) follows on average a line-frequency-varying input current. It is converted by the transistor full-bridge circuit into a switching current iT, mv which varies in switching frequency.

The transformer is in Fig. 2 in the form of an equivalent circuit diagram, i.e. with an explicit leakage inductance Lσ (which can be increased by a series inductance) and an ideal transformer which effects the voltage transformation approximately according to the number of turns ratio. The transformer secondary winding is connected to the input of a secondary full diode bridge circuit without smoothing on the output side via a further secondary resonance capacitor Cr, lv. If the primary-side transistor full-bridge circuit is now operated with the resonance frequency of the series branch formed from leakage inductance Lσ and the primary-side resonance capacitor Cr, mv and the secondary-side resonance capacitor Cr, lv, the series branch is to be regarded as a short circuit, the output voltage vT, mv of the transistor switching stage is therefore directly connected to the Primary winding of the ideal transformer and is translated into a voltage that is the same over the transformer secondary winding, but scaled with the number of turns ratio. At the output of the full-diode full-bridge circuit on the secondary side, a smooth voltage curve vC, lv then occurs again in the form of an absolute value voltage (| AC | voltage), which is used as a step-up converter as a secondary-side intermediate circuit voltage by the downstream, in the present case known for single-phase pulse rectifier circuits with power factor correction The inductance LB of the control converter 21 of the control converter stage 2 is converted into a constant output voltage Vout of the overall system supported by a capacitor, the input current iLb of the control converter 21 having a profile which is proportional to the secondary-side intermediate circuit voltage vC, lv (see FIG. 3).

The power transfer between the primary and secondary side takes place by resonant current half-oscillations (series resonance of the leakage inductance of the transformer and the primary and secondary capacitors), for which an amplitude is set such that the current drawn by the control converter stage from the secondary DC link voltage iLb is delivered on average over a cycle period; In order to accomplish the subsequent delivery with a minimal current amplitude, pauses between the current pulses are therefore advantageously avoided and the operating frequency ideally chosen to be equal to the resonance frequency; in real terms, a pause time must be observed between the current pulses in such a way that voltage-free switching of the switching elements of the transistor full-bridge circuit is ensured, i.e. the discharge of the parasitic output capacitances of a switching element is waited for before switching on. If the operating frequency is ideally assumed to be the same as the resonance frequency, the series connection of resonance capacitance and leakage inductance is operated in resonance and thus has a negligible impedance, or only a low voltage occurs across the series connection (ideal, ie if losses are neglected, no voltage occurs on). The voltage present at the output of the transistor switching stage is thus transformed directly into a voltage occurring at the terminals of the secondary diode full bridge circuit in accordance with the number of turns ratio.

It should be noted that for practical implementation in the primary and secondary-side intermediate circuit, capacitors with a small capacitance can advantageously be arranged (shown in broken lines) in order to ensure a low-impedance path for the current commutation of the primary-side transistor full-bridge circuit, or switching-frequency fluctuations in the input current Filter converter stage 2 (of the step-up converter). The | AC | voltage curve is fundamentally not influenced by these capacitances; However, the current required for the charge reversal of the capacitor in the secondary-side and in the primary-side intermediate circuit can advantageously be taken into account when specifying the current impressed by the control converter stage 2 such that a phase shift in the ultimately resulting mains current ig and the mains voltage vg is largely minimized.

Furthermore, it should be pointed out that the circuit shown in FIG. 2 represents only one of numerous possible embodiments of the concept according to the invention.

In particular, the primary-side as well as the secondary-side diode bridge circuit can be replaced by a full transistor bridge circuit (which would also initially enable active rectification and thus advantageously a reduction in the conduction losses) and a transistor arranged antiparallel to the step-up converter diode and a diode antiparallel to the step-up converter transistor and such a bidirectional power transfer, ie also inverter operation. The primary-side full-bridge circuit would then only be clocked at the mains frequency; on the secondary side, in order to reliably avoid saturation of the transformer, a capacitance would have to be connected in series with the secondary winding. Furthermore, in addition to the full-bridge circuit, half-bridge circuits or midpoint circuits (including the center point of the primary or secondary winding of the transformer in the circuit function) could also be used.

Finally, the primary-side indirect matrix converter shown in FIG. 2 can also be replaced by a direct matrix converter with four-quadrant switches (for example realized by series connection of power transistors with an anti-parallel freewheeling diode or counter-parallel connection of reverse-blocking power semiconductors), which directly converts the low-frequency AC input voltage into a converter module the symmetrical square-wave AC voltage described above and shown in FIG. 3 - which is applied to the transformer primary winding via the series capacitance - is transformed (see FIG. 4).

In addition, instead of a simple step-up converter for implementing the regulating converter stage, any other converter circuit could also be used, which is able to impress a current profile proportional to the secondary-side intermediate circuit voltage over a clock period and e.g. can also have a step-up and step-down converter function. Examples include SEPIC converters, Cuk converters or inverse converters. Furthermore, multilevel converter circuits (e.g. three-level step-up converter) or parallel circuits of several similar converter stages with continuous or discontinuous or triangular current routing (TCM or resonant transition mode) can be used.

A modification of the circuit shown in FIG. 2, which is particularly interesting in terms of implementation complexity, represents the omission of the primary-side diode bridges at module level and replacement by a diode bridge with diodes corresponding to all converter modules, with a correspondingly high blocking voltage resistance. This can be advantageous with regard to a high overall efficiency of the system common diode bridge also replaced by a transistor full-bridge circuit and thus be designed as an active rectifier, the transistors only having to be clocked at the mains frequency. If the transistors are expanded to four-quadrant switches (e.g. realized by the series connection of power transistors with an anti-parallel free-wheeling diode), there is also bidirectional energy flow, e.g. in the case of an AC / DC application, both rectifier and inverter operation possible.

Finally, it should be pointed out that the circuit shown in FIG. 2 also allows operation with a DC instead of an AC input voltage (DC voltage as a special case of an AC voltage with frequency = 0). This also applies to all circuits described below. Due to the clocking of the primary-side indirect or direct matrix converter stage, the potential separation is independent of the mains frequency.

[0040] The concept according to the invention can also advantageously be used for generating a regulated single-phase AC output voltage instead of a DC output voltage. A realization that is advantageous in this regard is shown in FIG. 4 with reference to FIG. 1 (a) (only one control converter 21 for all converter modules 11 of the potential isolation stage connected in parallel on the secondary side). Alternatively, according to FIG. 1 (b), a separate AC / AC control converter 21 could also be arranged for each of the series-resonant AC / AC converter modules 11.

The primary-side and secondary-side switching stage of the converter modules is exemplified as a direct AC / AC matrix converter in half-bridge circuit with a bridge arm with four-quadrant switches and a second bridge arm in the form of a capacitive voltage divider, the transformer primary and the transformer secondary winding each between the output of the associated Bridge branch and the center of the associated capacitive voltage divider lie and the capacitors Cmv, 1, Cmv, 2, Clv, 1, Clv, 2 of the voltage divider also take over the function of the primary and secondary resonance capacitor. The output voltage of the secondary matrix converter stage, i.e. the secondary-side AC intermediate circuit voltage is converted by the AC / AC control converter 21 in the form of an AC / AC step-up or step-down converter implemented with four-quadrant switches (or another circuit structure, such as an AC / AC Cuk converter) into the actual regulated output voltage of the entire system, again supported by a capacitance.

Another advantageous embodiment of the concept according to the invention is the converter system shown in FIG. 5 (b) with a three-phase AC input via input filter inductances LF, R, LF, S, LF, T and a regulated DC output. The converter modules of the potential isolation stage are only shown schematically, a practical implementation of the individual converter modules 11 could, for example 2 take place.

Basically, the transformation of a three-phase AC voltage into a DC voltage for each phase can be seen as an AC / DC conversion, and thus a single-phase AC / DC converter has to be arranged for each phase. In the simplest case, the phase systems can then be connected in a star or triangle on the primary side, on the secondary side a direct parallel connection of the module outputs of a phase according to FIG. 1 (a) or a connection according to FIG. 1 (b) is possible. The subordinate control converters 21 can only be brought together at the terminals of the output voltage of the overall system, or a one-sided connection of the outputs of the converter modules of the phases, e.g. negative rail of the total output voltage of the system are formed and the arrangement of the control converter 21 (step-up converter functions) is based on this voltage rail. The outputs of the control converters 21 assigned to the phases (here step-up converter) are then brought together at the positive terminal of the output voltage of the overall system. Due to the constant instantaneous power of a symmetrically loaded symmetrical three-phase system, it is advantageous that no capacitive buffering of the output voltage of the overall system is ideally required, or an output capacitor with only a small capacitance can actually be used.

Compared to that described above, the same for all phases, i.e. Implementation of the concept according to the invention comprising three potential isolating stages and three regulating converter stages has the distinct advantages shown in FIG. 5 (b). The basic idea is to simulate the function of a Scott transformer (see FIG. 5 (a)), which converts a three-phase voltage system into two orthogonal voltages of the same amplitude. If the orthogonal output voltages are loaded equally, a symmetrical load of the three-phase network results on the primary side. It should be noted that other transformer circuits are also known for such a voltage conversion; the Leblanc circuit may be mentioned as an example, which, however, has a relatively complicated winding configuration. In principle, the implementation of the basic concept described below with reference to the Scott circuit can also be used for these circuits.

Instead of each (partial) winding of the conventional Scott transformer, a potential isolation stage 1XT, 1SX, 1RX is arranged, the outputs of the converter modules of the two between the network phases S and T arranged and having a common circuit point X potential isolation stages (SX- Isolation stage 1SX and XT isolation stage 1XT) are either all directly connected, so that only a subsequent ST control converter stage 2ST is required for both isolation stages to form a first DC output voltage Vdc, 2 (as in FIG. 5 (b) shown), or (not shown) the outputs of the converter modules of the SX potential isolation stage can be merged and provided with a subsequent SX control converter stage and the outputs of the converter modules of the XT potential isolation stage can also be connected and with their own subsequent XT control stage Converter stage and then the outputs of the SX and XT control converter stages are connected in order to form the first DC output voltage, an identical input resistance then having to be emulated at the input of the SX and XT control converter stages in order to symmetrically distribute the mains voltage present between phases S and T between the inputs of the SX - and XT potential isolation stage.

Between phase R and the common node X, an RX potential isolation stage 1RX with subsequent RX control converter stage 2RX is placed, which forms a second DC output voltage Vdc, 1. The input current setpoints of the ST control converter stage and the RX control converter stage must be specified such that both stages consume the same average power over a network period. If an SX and an XT control converter stage are provided, then half of the power to be provided for the ST control converter stage 2ST is to be supplied by both stages. Furthermore, in order to simulate the magnitude relationships of the voltages between the switching points R and X, S and X as well as X and T for the conventional Scott transformer circuit, half the number of converter modules used for the RX potential isolating stage is required for the SX and XT isolating stages and the ratio of the turns of the transformers of the SX and XT potential isolators compared to the transformers of the RX potential isolators by a factor of 2 / sqrt (3) higher (higher number of primary turns), so that at the input of the RX and ST control converter stages 2RX , 2ST voltages with the same amplitude. The power semiconductors of the primary sides of the converter modules of the SX and XT potential isolation stage 1SX, 1XT are then compared to the RX potential isolation stage 1RX by a factor of 2 / sqrt (3) = 1.15, i.e. 15% higher than the primary-side power semiconductors of the RX potential isolation stage 1RX. In order to avoid this, the number of modules and the ratio of turns could be adjusted, whereby in any case it must be ensured that the switching point X lies in the middle between S and T in terms of voltage.

In addition to the symmetrical division of the voltage between S and T, the circuit described above also implements the flux balance for the winding of the conventional Scott transformer lying between the phases S and T with the associated secondary winding, since the input current of the ST -Rule converter stage 2ST will be divided among the converter modules of the SX and XT potential isolation stages in such a way that the flow equilibrium is given in the branches SX and XT. A major advantage of the described replica of the Scott transformer is that in the simplest case only two and not three control converter stages are required, as in the three-phase case with three secondary DC link voltages. Since the RX and ST control converter stages are supplied potential-free, the first and second DC output voltage Vdc, 2, Vdc, 1 can in principle be connected in parallel or in series. The series connection is e.g. can be used advantageously for supplying the partial input voltages of a three-point pulse inverter or for generating an earth-symmetrical DC voltage, in which case the connection point of the first and second output voltages must be connected to earth or ground.

It should be noted that the replacement of a conventional transformer by a circuit structure described above using the example of the Scott transformer is basically the same - i.e. with consideration of the simulation of the voltage transformation as well as the flow equilibrium of the associated winding parts - also for other transformer circuits with any combination of a primary-side delta, star, or zigzag circuit and a secondary-side triangle, star, or zigzag circuit.

A further advantageous embodiment of the concept described for the implementation of a potential-isolated converter system with three-phase AC input and regulated three-phase AC output is shown in FIG. 6. The converter modules of the electrical isolation stage are only shown schematically, a practical implementation of the converter modules could e.g. 4 take place.

Each phase of the network is assigned a single-phase AC / AC potential isolation stage 1R, 1S, 1T, the output voltages of the potential isolation stages 1R, 1S, 1T, i.e. the assigned secondary-side AC intermediate circuit voltages are connected in a star, leaving three phase outputs u, v, w in addition to the star point, to which advantageously, as an alternative to single-phase AC / AC control converter stages arranged between these phase terminals and the star point, an implementation effort that is advantageous Three-phase control converter stage 2 is placed, which is shown in Fig. 6 as a step-up converter. Three series inductances LB, U, LB, V, LB, W are then connected to the three phase outputs u, v, w on the secondary side, and other star ends of transistors with antiparallel diodes are connected to the winding ends facing away from the phase outputs u, v, w. Furthermore, diodes with anti-parallel transistors are switched from these winding ends to the three phase output terminals U, V, W of the overall system, the voltages of which, e.g. can be buffered by a star connection of output capacitors.

In addition to the primary-side delta connection and secondary-side star connection shown in FIG. 6, any other three-phase combinations of the primary and secondary-side phase branches, such as e.g. a triangle-triangle or star-star connection, can be used.

In conclusion, it should be pointed out that, just as for the three-phase AC / AC step-up converter shown in FIG. 6, also for the implementation of a three-phase AC / AC step-down converter or other DC / DC converter known in the basic function. Basic circuits corresponding circuits (AC / AC-SEPIC converter or AC / AC inversion converter) only unidirectionally controllable switches, ie Transistors with anti-parallel freewheeling diode are required. All of these circuits can be used with the potential isolation stage described above.

Fig. 7 illustrates different variants a, b, c, d for soft switching of resonance converters used in particular as described above over the entire network period and within the entire load range by changing the switching frequency and the duty cycle of the square wave voltage considered symmetrical within a clock period for the potential separation in the converter modules 11. Pulse periods of the high-frequency switched bridge currents are designated in the different variants with T_s, a T_s, b etc., the pulse period of a primary-side AC voltage with T_g.

As mentioned at the outset, the resonant operation of the potential isolation stage offers the advantage of a current profile set on the primary and secondary side from sine half-oscillations (starting from a primary-side voltage impression, v_T, mv, which is considered here by way of example without restricting generality, through a transistor switching stage, the current is still on the primary side the magnetizing current of the transformer is superimposed), which leads to low values of the current to be switched by the transistors, since only the relatively small transformer magnetizing current i_M, mv is to be actively switched. On the one hand, voltage-free or soft switching can be achieved by deliberately reaching the value zero just before the point at which the sine current half-oscillation (secondary-side current) i_r, lv (with the envelope i_r, lv, e proportional to the mains current) . goes back to the magnetizing current) is switched to a sufficiently high current for the recharging of parasitic capacitances of the transistor switching stage, ie in particular for discharging the parasitic output capacitance of the next transistor to be switched on (hereinafter referred to as transistor output capacitance). Alternatively, according to the invention, FIG. 7 (b-d) is also possible, where switching only takes place after the sine current half-oscillation (secondary-side current i_r, lv) has been switched and thus only the relatively small magnetizing current of the transformer i_M, mv has to be interrupted on the primary side. Due to the small value of the current then available for the recharging of the parasitic capacitances, however, a sufficiently long waiting time t_d, b must be inserted after switching off a transistor until the next transistor is switched on (FIG. 7 (b)) in order to complete the process Ensure discharge of the transistor output capacitance and avoid a partially hard switching (see Fig. 7 (a)), which would lead to switching losses again. Note that due to the same duty cycle of the power transistor in Fig. 7 (b), the magnetizing current rises to the same value iDach_M, b as the magnetizing current iDach_M, a in Fig. 7 (a).

Another possibility is, as shown in Fig. 7 (c), to leave the voltage v_T, mv applied to the transformer after the current has become zero, whereby the magnetizing current reaches a higher value and thus the charge after switching off can be completed more quickly, ie within the previously insufficient waiting time t_d, a, or, as shown in FIG. 7 (c), a shorter waiting time t_d, c. However, this increases the core losses of the transformer due to the higher magnetic modulation and also increases the winding losses due to the higher effective value of the primary current. On the other hand, there is an overall shorter duration of the pulse interval T_s, c, with which a smaller amplitude of the secondary current pulse is sufficient to deliver a defined charge to the output within the pulse interval. This means lower conduction losses of the power semiconductors and lower transformer winding losses.

A mixed form of the method according to FIGS. 7 (b) and 7 (c), i.e. the controller shown in Fig. 7 (d) are used, for which on the one hand the magnetizing current is slightly increased by the remaining voltage on the transformer after the secondary current has become zero and on the other hand the waiting time t_d, d is slightly extended. The exact switching times can then be determined via an optimization with the aim of minimal total losses of the potential isolation stage, the resulting parameters relating to the course of a mains voltage half-oscillation, which as envelope v_T, mv, e of the transformer primary voltage in accordance with. Fig. 7 (e) appears to vary (typically a lowering of the switching frequency or an increase in the pulse interval near the zero crossings of the mains voltage is required, at the same time the waiting time has to be adjusted with regard to the lowest possible total losses), i.e. generally have a dependency on the current intermediate circuit voltage and the resulting parameters can also vary depending on the transmitted power. In this case, it may also be advantageous to record the point in time of the complete discharge of the transistor output capacitance using a sensor circuit in order to minimize the duration of the waiting time and to avoid model uncertainties compared to a pure control.

Claims (9)

1.Converter for electrically isolated transmission of electrical energy between a primary system and a secondary system, comprising a primary-side potential isolation stage (1), which is connected for energy exchange between the primary system and at least one intermediate circuit, and a secondary-side converter converter stage (2), which for energy exchange between the at least one intermediate circuit and the secondary system are connected, • wherein the potential isolation stage (1) is an unregulated resonant potential isolation stage (1) and is set up to transform one or more primary voltages, which are predetermined by the primary system, to an, in particular lower, voltage level of the at least one intermediate circuit and thus an intermediate circuit voltage at the at least one intermediate circuit (vC, lv) in the form of a sine voltage or a magnitude sine voltage, and • The control converter stage (2) is set up to convert an intermediate circuit voltage into a secondary voltage, the secondary voltage being a direct voltage or sinusoidal alternating voltage. 2. Converter according to claim 1, wherein the potential isolation stage (1) has at least two converter modules (11) connected in series on the primary side, each of which has a transformer and can be operated as a series resonance converter operated at least near the resonance point, in order thereby to generate a primary voltage or a portion of one To transform primary voltage with a transmission ratio defined by a number of turns of the transformer into a smooth intermediate circuit voltage (vC, lv), and convert these modules (11) • are either connected in parallel on the DC link side and feed a common DC link, to which a single control converter (21) of the control converter stage (2) is connected, and this control converter (21) is set up to supply the DC link on average via a Switching period to take a current proportional to the DC link voltage, i.e. sinusoidal or sinusoidal, the amplitude of this current can be regulated in such a way that a power exchange with the secondary system is covered, or a predetermined DC or AC voltage (Vout) occurs at the connection of the secondary system ; • or on the intermediate circuit side each feed its own, assigned intermediate circuit, to each of which a dedicated control converter (21) of the control converter stage (2) is connected, wherein these control converters (21) can be connected in parallel on the secondary side and in this If they are set up to draw a current from the assigned intermediate circuit in such a way that each control converter (21) on the one hand makes an equal contribution to the exchange of power with the secondary system or to the maintenance of a predetermined DC or AC voltage (Vout) at the connection of the secondary system , and on the other hand realizes an equal distribution of the primary voltage to the inputs of the converter modules (11) of the potential isolation stage (1). 3. Converter according to claim 2, wherein the potential isolation stage (1) is set up to clock the at least two converter modules (11) out of phase with one another. 4. Converter according to one of the claims 2 to 3, wherein the converter modules (11) are set up for soft switching over an entire network period and regardless of load to generate a swept symmetrical square wave voltage on the respective transformer such that the frequency of the resonance frequency of the each converter module (11) operated as a series resonance converter is different and, in particular, deeper, and so to form a magnetizing current of the transformer which promotes soft switching and to switch on a transistor only when the voltage across the transistor has become zero. 5. Converter according to claim 1, wherein the potential isolation stage (1) is set up to be connected to a three-phase network forming primary system, and the control converter stage (2) is set up to a first and a second DC voltage connection of the secondary system to be connected, the converter having the following structure: • on the primary side, between a first phase connection point (R) and a connection point (X), a first series circuit of first converter modules (11) is connected, which are connected in parallel on the intermediate circuit side and feed a common first intermediate circuit, on which a first control converter (21) Control converter stage (2) is connected and is set up for connection to the first DC voltage connection of the secondary system; A second series connection of second converter modules (11) is connected on the primary side between a second phase connection point (S) and the connection point (X), On the primary side, between a third phase connection point (T) and the connection point (X), a third series connection of third converter modules (11) is connected in a number equal to that of the second series connection, • The second and third converter modules (11) of the second and third series circuit are connected in parallel on the intermediate circuit side and feed a common second intermediate circuit, to which a second regulating converter (21) of the regulating converter stage (2) is connected and for connection to the second DC voltage connection of the secondary system is set up, wherein the first, second and third converter modules (11) each have a transformer and can be operated as a series resonance converter operated at least near the resonance point. 6. Converter according to claim 5, in which the number of turns ratio of the transformers of the first series circuit compared to the second and third series circuit is such that voltages of the same amplitude occur in the first and second intermediate circuit. 7. Converter according to one of the claims 2 to 6, with only a single, common diode bridge for all converter modules (11) of the potential isolation stage (1). 8. Converter according to one of the preceding claims, wherein the potential isolation stage (1) is set up to generate the intermediate circuit voltage from one or more AC voltages predetermined by the primary system. 9. Converter according to one of the preceding claims, wherein the control converter stage (2) is set up to transform the intermediate circuit voltage into a DC voltage of the secondary system.